I) EQUATION 1: SHUNT-CONTROLLED

Transcription

1 Swich Mode Power Supply (SMPS) Topologies (Par I) Auhor: Mohammad Kamil Microchip Technology Inc. EQUATION 1: SHUNT-CONTROLLED REGULATOR POWER LOSS INTRODUCTION The indusry drive oward smaller, ligher and more efficien elecronics has led o he developmen of he Swich Mode Power Supply (SMPS). There are several opologies commonly used o implemen SMPS. This applicaion noe, which is he firs of a wo-par series, explains he basics of differen SMPS opologies. Applicaions of differen opologies and heir pros and cons are also discussed in deail. This applicaion noe will guide he user o selec an appropriae opology for a given applicaion, while providing useful informaion regarding selecion of elecrical and elecronic componens for a given SMPS design. WHY SMPS? The main idea behind a swich mode power supply can easily be undersood from he concepual explanaion of a DC-o-DC converer, as shown in Figure 1. The load, RL, needs o be supplied wih a consan volage, VOUT, which is derived from a primary volage source, VIN. As shown in Figure 1, he oupu volage VOUT can be regulaed by varying he series resisor (RS) or he shun curren (IS). When VOUT is conrolled by varying IS and keeping RS consan, power loss inside he converer occurs. This ype of converer is known as shun-conrolled regulaor. The power loss inside he converer is given by Equaion 1. Please noe ha he power loss canno be eliminaed even if IS becomes zero. FIGURE 1: VIN DC-DC CONVERTER R S IS IOUT R L V OUT However, if we conrol he oupu volage VOUT by varying RS and keeping IS zero, he ideal power loss inside he converer can be calculaed as shown in Equaion 2. EQUATION 2: P LOSS = V OUT I S + ( I OUT + I S ) 2 R S SERIES-CONTROLLED REGULATOR POWER LOSS 2 P LOSS = ( + ) 2 This ype of converer is known as a series-conrolled regulaor. The ideal power loss in his converer depends on he value of he series resisance, RS, which is required o conrol he oupu volage, VOUT, and he load curren, IOUT. If he value of RS is eiher zero or infinie, he ideal power loss inside he converer should be zero. This feaure of a series-conrolled regulaor becomes he seed idea of SMPS, where he conversion loss can be minimized, which resuls in maximized efficiency. In SMPS, he series elemen, RS, is replaced by a semiconducor swich, which offers very low resisance a he ON sae (minimizing conducion loss), and very high resisance a he OFF sae (blocking he conducion). A low-pass filer using non-dissipaive passive componens such as inducors and capaciors is placed afer he semiconducor swich, o provide consan DC oupu volage. The semiconducor swiches used o implemen swich mode power supplies are coninuously swiched on and off a high frequencies (50 khz o several MHz), o ransfer elecrical energy from he inpu o he oupu hrough he passive componens. The oupu volage is conrolled by varying he duy cycle, frequency or phase of he semiconducor devices ransiion periods. As he size of he passive componens is inversely proporional o he swiching frequency, a high swiching frequency resuls in smaller sizes for magneics and capaciors. While he high frequency swiching offers he designer a huge advanage for increasing he power densiy, i adds power losses inside he converer and inroduces addiional elecrical noise. R S R S R L 2007 Microchip Technology Inc. DS01114A -page 1

2 SELECTION OF SMPS TOPOLOGIES There are several opologies commonly used o implemen SMPS. Any opology can be made o work for any specificaion; however, each opology has is own unique feaures, which make i bes suied for a cerain applicaion. To selec he bes opology for a given specificaion, i is essenial o know he basic operaion, advanages, drawbacks, complexiy and he area of usage of a paricular opology. The following facors help while selecing an appropriae opology: a) Is he oupu volage higher or lower han he whole range of he inpu volage? b) How many oupus are required? c) Is inpu o oupu dielecric isolaion required? d) Is he inpu/oupu volage very high? e) Is he inpu/oupu curren very high? f) Wha is he maximum volage applied across he ransformer primary and wha is he maximum duy cycle? Facor (a) deermines wheher he power supply opology should be buck, boos or buck-boos ype. Facors (b) and (c) deermine wheher or no he power supply opology should have a ransformer. Reliabiliy of he power supply depends on he selecion of a proper opology on he basis of facors (d), (e) and (f). Buck Converer A buck converer, as is name implies, can only produce lower average oupu volage han he inpu volage. The basic schemaic wih he swiching waveforms of a buck converer is shown in Figure 2. In a buck converer, a swich (Q1) is placed in series wih he inpu volage source VIN. The inpu source VIN feeds he oupu hrough he swich and a low-pass filer, implemened wih an inducor and a capacior. In a seady sae of operaion, when he swich is ON for a period of TON, he inpu provides energy o he oupu as well as o he inducor (L). During he TON period, he inducor curren flows hrough he swich and he difference of volages beween VIN and VOUT is applied o he inducor in he forward direcion, as shown in Figure 2 (C). Therefore, he inducor curren IL rises linearly from is presen value IL1 o IL2, as shown in Figure 2 (E). During he TOFF period, when he swich is OFF, he inducor curren coninues o flow in he same direcion, as he sored energy wihin he inducor coninues o supply he load curren. The diode D1 complees he inducor curren pah during he Q1 OFF period (TOFF); hus, i is called a freewheeling diode. During his TOFF period, he oupu volage VOUT is applied across he inducor in he reverse direcion, as shown in Figure 2 (C). Therefore, he inducor curren decreases from is presen value IL2 o IL1, as shown in Figure 2 (E). DS01114A -page Microchip Technology Inc.

3 FIGURE 2: BUCK CONVERTER I IN Q 1 (A) VIN L + - IL I OUT D 1 V OUT (B) Q 1GATE (C) V L - V OUT -V OUT ( - V OUT )/L (D) I IN (E) I L I L2 I L1 (A) = Buck converer (B) = Gae pulse of MOSFET Q 1 (C) = Volage across he Inducor L (D) = Inpu curren I IN (E) = Inducor curren I L -V OUT /L CONTINUOUS CONDUCTION MODE The inducor curren is coninuous and never reaches zero during one swiching period (TS); herefore, his mode of operaion is known as Coninuous Conducion mode. In Coninuous Conducion mode, he relaion beween he oupu and inpu volage is given by Equaion 3, where D is known as he duy cycle, which is given by Equaion 4. EQUATION 4: DUTY CYCLE T ON T S D = where: T ON = ON Period T S = Swiching Period EQUATION 3: BUCK CONVERTER VOUT/VIN RELATIONSHIP V OUT = D If he oupu o inpu volage raio is less han 0.1, i is always advisable o go for a wo-sage buck converer, which means o sep down he volage in wo buck operaions. Alhough he buck converer can be eiher coninuous or disconinuous, is inpu curren is always disconinuous, as shown in Figure 2 (D). This resuls in a larger elecromagneic inerference (EMI) filer han he oher opologies Microchip Technology Inc. DS01114A -page 3

4 CURRENT MODE CONTROL While designing a buck converer, here is always a rade-off beween he inducor and he capacior size selecion. A larger inducor value means numerous urns o he magneic core, bu less ripple curren (<10% of full load curren) is seen by he oupu capacior; herefore, he loss in he inducor increases. Also, less ripple curren makes curren mode conrol almos impossible o implemen (refer o Mehod of Conrol for deails on curren mode conrol echniques). Therefore, poor load ransien response can be observed in he converer. A smaller inducor value increases ripple curren. This makes implemenaion of curren mode conrol easier, and as a resul, he load ransien response of he converer improves. However, high ripple curren needs a low Equivalen Series Resisor (ESR) oupu capacior o mee he peak-o-peak oupu volage ripple requiremen. Generally, o implemen he curren mode conrol, he ripple curren a he inducor should be a leas 30% of he full load curren. FEED-FORWARD CONTROL In a buck converer, he effec of inpu volage variaion on he oupu volage can be minimized by implemening inpu volage feed-forward conrol. I is easy o implemen feed-forward conrol when using a digial conroller wih inpu volage sense, compared o using an analog conrol mehod. In he feed-forward conrol mehod, he digial conroller sars aking he appropriae adapive acion as soon as any change is deeced in he inpu volage, before he change in inpu can acually affec he oupu parameers. SYNCHRONOUS BUCK CONVERTER When he oupu curren requiremen is high, he excessive power loss inside he freewheeling diode D1, limis he minimum oupu volage ha can be achieved. To reduce he loss a high curren and o achieve lower oupu volage, he freewheeling diode is replaced by a MOSFET wih a very low ON sae resisance RDSON. This MOSFET is urned on and off synchronously wih he buck MOSFET. Therefore, his opology is known as a synchronous buck converer. A gae drive signal, which is he complemen of he buck swich gae drive signal, is required for his synchronous MOSFET. A MOSFET can conduc in eiher direcion; which means he synchronous MOSFET should be urned off immediaely if he curren in he inducor reaches zero because of a ligh load. Oherwise, he direcion of he inducor curren will reverse (afer reaching zero) because of he oupu LC resonance. In such a scenario, he synchronous MOSFET acs as a load o he oupu capacior, and dissipaes energy in he RDSON (ON sae resisance) of he MOSFET, resuling in an increase in power loss during disconinuous mode of operaion (inducor curren reaches zero in one swiching cycle). This may happen if he buck converer inducor is designed for a medium load, bu needs o operae a no load and/or a ligh load. In his case, he oupu volage may fall below he regulaion limi, if he synchronous MOSFET is no swiched off immediaely afer he inducor reaches zero. MULTIPHASE SYNCHRONOUS BUCK CONVERTER I is almos impracical o design a single synchronous buck converer o deliver more han 35 amps load curren a a low oupu volage. If he load curren requiremen is more han amps, more han one converer is conneced in parallel o deliver he load. To opimize he inpu and oupu capaciors, all he parallel converers operae on he same ime base and each converer sars swiching afer a fixed ime/phase from he previous one. This ype of converer is called a muliphase synchronous buck converer. Figure 3 shows he muliphase synchronous buck converer wih a gae pulse iming relaion of each leg and he inpu curren drawn by he converer. The fixed ime/phase is given by Time period/n or 300/n, where n is he number of he converer conneced in parallel. The design of inpu and oupu capaciors is based on he swiching frequency of each converer muliplied by he number of parallel converers. The ripple curren seen by he oupu capacior reduces by n imes. As shown in Figure 3 (E), he inpu curren drawn by a muliphase synchronous buck converer is coninuous wih less ripple curren as compared o a single converer shown in Figure 2 (D). Therefore, a smaller inpu capacior mees he design requiremen in case of a muliphase synchronous buck converer. DS01114A -page Microchip Technology Inc.

5 FIGURE 3: MULTIPHASE SYNCHRONOUS BUCK CONVERTER + IQ 1 IQ 3 IQ 5 Q 1 Q 3 Q 5 L 3 I L3 V OUT (A) C IN L 2 I L2 L 1 I L1 C O Q 2 Q 4 Q 6 - I L1 (B) Q 1PWM I L2 (C) Q 3PWM I L3 (D) Q 5PWM IQ 5 +IQ 1 IQ 1 +IQ 3 IQ 3 +IQ 5 IQ 5 +IQ 1 (E) I IN IQ 1 IQ 3 IQ 5 (A) = Muliphase Synchronous Buck converer (B) = Gae pulse of Q 1, inducor curren I L1 (C) = Gae pulse of Q 3, Inducor curren I L2 (D) = Gae pulse of Q 5, Inducor curren I L3 (E) = Inpu curren I IN 2007 Microchip Technology Inc. DS01114A -page 5

6 Boos Converer A boos converer, as is name implies, can only produce a higher oupu average volage han he inpu volage. The basic schemaic wih he swiching waveform of a boos converer is shown in Figure 4. In a boos converer, an inducor (L) is placed in series wih he inpu volage source VIN. The inpu source feeds he oupu hrough he inducor and he diode D1. In he seady sae of operaion, when he swich Q1 is ON for a period of TON, he inpu provides energy o he inducor. During he TON period, inducor curren (IL) flows hrough he swich and he inpu volage VIN is applied o he inducor in he forward direcion, as shown in Figure 4 (C). Therefore, he inducor curren rises linearly from is presen value IL1 o IL2, as shown in Figure 4 (D). During his TON period, he oupu load curren IOUT is supplied from he oupu capacior CO. The oupu capacior value should be large enough o supply he load curren for he ime period TON wih he minimum specified droop in he oupu volage. During he TOFF period when he swich is OFF, he inducor curren coninues o flow in he same direcion as he sored energy wih he inducor, and he inpu source VIN supplies energy o he load. The diode D1 complees he inducor curren pah hrough he oupu capacior during he Q1 OFF period (TOFF). During his TOFF period, he inducor curren flows hrough he diode and he difference of volages beween VIN and VOUT is applied o he inducor in he reverse direcion, as shown in Figure 4 (C). Therefore, he inducor curren decreases from he presen value IL2 o IL1, as shown in Figure 4 (D). CONTINUOUS CONDUCTION MODE As shown in Figure 4 (D), he inducor curren is coninuous and never reaches zero during one swiching cycle (TS); herefore, his mehod is known as Coninuous Conducion mode, which is he relaion beween oupu and inpu volage, as shown in Equaion 5. FIGURE 4: BOOST CONVERTER + I L D 1 + V L - I D1 I OUT + (A) Q 1 C O V OUT - - (B) Q 1PWM (C) V L V OUT - I L2 IQ 1 I D1 (D) I L1 (E) V DS (A) = Boos converer (B) = Gae pulse of MOSFET Q 1 (C) = Volage across he inducor L (D) = Curren hrough he MOSFET Q 1 and diode D 1 (E) = Volage across he MOSFET Q 1 V OUT DS01114A -page Microchip Technology Inc.

7 EQUATION 5: VOUT/VIN RELATIONSHIP The roo mean square (RMS) ripple curren in he oupu capacior is given by Equaion 6. I is calculaed by considering he waveform shown in Figure 4 (D). During he TOFF period, he pulsaing curren ID1, flows ino he oupu capacior and he consan load curren (IOUT) flows ou of he oupu capacior. EQUATION 6: CAPACITOR RIPPLE RMS CURRENT Based on Equaion 5, he VOUT/VIN raio can be very large when he duy cycle approaches uniy, which is ideal. However, unlike he ideal characerisic, VOUT/VIN declines as he duy raio approaches uniy, as shown in Figure 5. Because of very poor uilizaion of he swich, parasiic elemens occur in he componens and losses associaed wih he inducor capacior and semiconducors. FIGURE 5: V OUT = ( 1 D) I RIPPLERMS = ( I D1 ) 2 ( I OUT ) 2 where: I D1RMS = RMS value of I D1 I RIPPLERMS = Ripple RMS curren of capacior I OUT = Oupu DC curren V OUT / VOUT/VIN AND DUTY CYCLE IN BOOST CONVERTER Ideal Duy Cycle = D POWER FACTOR CORRECTION Pracical When he boos converer operaes in Coninuous Conducion mode, he curren drawn from he inpu volage source is always coninuous and smooh, as shown in Figure 4 (D). This feaure makes he boos converer an ideal choice for he Power Facor Correcion (PFC) applicaion. Power Facor (PF) is given by he produc of he Toal Curren Harmonics Disorion Facor (THD) and he Displacemen Facor (DF). Therefore, in PFC, he inpu curren drawn by he converer should be coninuous and smooh enough o mee he THD of he inpu curren so ha i is close o uniy. In addiion, inpu curren should follow he inpu sinusoidal volage waveform o mee he displacemen facor so ha i is close o uniy. Forward Converer A forward converer is a ransformer-isolaed converer based on he basic buck converer opology. The basic schemaic and swiching waveforms are shown in Figure 6. In a forward converer, a swich (Q1) is conneced in series wih he ransformer (T1) primary. The swich creaes a pulsaing volage a he ransformer primary winding. The ransformer is used o sep down he primary volage, and provide isolaion beween he inpu volage source VIN and he oupu volage VOUT. In he seady sae of operaion, when he swich is ON for a period of TON, he do end of he winding becomes posiive wih respec o he non-do end. Therefore, he diode D1 becomes forward-biased and he diodes D2 and D3 become reverse-biased. As he inpu volage VIN is applied across he ransformer primary, he magneizing curren IM increases linearly from is iniial zero value o a final value wih a slope of VIN/LM, where LM is he magneizing inducance of he primary winding, as shown in Figure 6(D). The oal curren ha flows hrough he primary winding is his magneizing curren plus he inducor curren (IL) refleced on he primary side. This oal curren flows hrough he MOSFET during he TON period. The volage across he diode D2 is equal o he inpu volage muliplied by he ransformer urns raio (NS/NP). In he case of a forward converer, he volage applied across he inducor L in he forward direcion during he TON period, is given by Equaion 7, neglecing he ransformer losses and he diode forward volage drop. EQUATION 7: DISSIPATING ENERGY FORWARD VOLTAGE ACROSS INDUCTOR N V L V S ΔI L = IN V OUT = L Δ N P A he end of he ON period, when he swich is urned OFF, here is no curren pah o dissipae he sored energy in he magneic core. There are many ways o dissipae his energy. One such mehod is shown in Figure 6. In his mehod, he flux sored inside he magneic core induces a negaive volage a he do end of he NR winding, which forward biases he diode D3 and reses he magneizing energy sored in he core. Therefore, he NR winding is called he rese winding. Reseing he magneizing curren during he OFF period is imporan o avoid sauraion. During he TOFF period when he swich is OFF, he inducor curren (IL) coninues o flow in he same direcion, while he sored energy wihin he inducor coninues o supply he load curren IOUT Microchip Technology Inc. DS01114A -page 7

8 FIGURE 6: FORWARD CONVERTER + I 3 + V P N P T 1 N S D 1 + I L + V L - D 2 V OUT (A) - I SW D N R - D 3 - Q 1 G S (B) Q 1PWM (C) V P (1+N P /N R ) I M I M (D) I IN IP I M I P I M T ON I 3 T M T OFF I 3 T S (E) V DS (1+N P /N R ) (1+N P /N R ) I L IOUT (F) I L ΔI L (A) = Forward Converer power circui diagram. (B) = Gae pulse of MOSFET Q 1 (C) = Volage across he ransformer primary winding N P (D) = Curren hrough N P and N R (E) = Volage across he MOSFET Q 1 (F) = Oupu Inducor curren I L DS01114A -page Microchip Technology Inc.

9 The diode D2, called a freewheeling diode, complees he inducor curren pah during he Q1 off period (TOFF). During his TOFF period, he oupu volage VOUT is applied across he inducor in he reverse direcion. In a coninuous conducion mode of operaion, he relaion beween he oupu volage and inpu volage is given by Equaion 8, where D is he duy cycle. EQUATION 8: FORWARD CONVERTER VOUT/VIN RELATIONSHIP N V S IN V N OUT T = V T ON OUT OFF P CONTROLLING MAGNETIZATION When he swich is urned OFF, he diode D1 becomes reverse-biased, and IM canno flow in he secondary side. Therefore, he magneizing curren is aken away by he rese winding of he ransformer, as shown in Figure 6(A and D). The refleced magneizing curren I3 flows hrough he rese winding NR and he diode D3 ino he inpu supply. During he inerval TM when I3 is flowing, he volage across he ransformer primary as well as LM is given by Equaion 9. EQUATION 9: N S V OUT = D REFLECTED VOLTAGE AT PRIMARY N P N R Time aken by he ransformer o complee he demagneizaion can be obained by recognizing ha he ime inegral of volage across he LM mus be zero over one ime period. The maximum value of TM, as shown in Figure 6, is he ime i akes he ransformer o compleely demagneize before he nex cycle begins and is equal o TOFF. Therefore, he maximum duy cycle and he maximum drain-o-source blocking volage (VDS) seen by he swich (Q1) in a forward converer having number of primary and number of rese winding urns as NP and NR, is given by Equaion 10. N P V IN EQUATION 10: MAXIMUM DUTY CYCLE AND VDS The maximum value of TM/TS o compleely demagneize before he nex cycle begins is equal o (1-D), so he maximum duy raio for he forward converer is given by Equaion 10. From Equaion 10, i is undersood ha when he number of primary winding urns, NP, is equal o he number of he rese winding urns, NR, he swich can have a maximum 50% duy cycle and he blocking volage of he swich will be equal o wice he inpu volage. The pracical limi of maximum duy cycle should be 45%, and maximum blocking volage seen by he swich will be more han wice he inpu volage due o he nonlineariy of componens and he leakage inducance of he ransformer. EQUATION 11: N R ( 1 D MAX ) = N P D MAX D 1 MAX = N R MAGNETIZING STORED ENERGY IN FLYBACK TRANSFORMER If NR is chosen o be less han NP, he maximum duy cycle DMAX can be more han 50%; however, he maximum blocking volage sress of he swich becomes more han 2 VIN he value of DMAX and VDS, as shown in Equaion 10. If NR is chosen o be larger han NP, DMAX will be less han 50%, bu he maximum blocking volage sress of he swich is now less han 2 VIN, he value of DMAX and VDS, as shown in Equaion 10. Since large volage isolaion is no required beween he rese and he primary windings, hese wo windings can be wound bifilar o minimize leakage inducance. The rese winding carries only he magneizing curren, which means i requires a smaller size of wire as compared o he primary winding. N P N P V DS = E P = -- ( I 2 PK ) 2 L M ( V I IN T ON ) PK = L where: M E P = Joules I PK = Amps L M = Henries N R 2007 Microchip Technology Inc. DS01114A -page 9

10 To demagneize he ransformer core, a Zener diode or RC snubber circui can also be used across he ransformer insead of he ransformer rese winding. The incomplee uilizaion of he magneics, he maximum duy cycle limi and he high volage sress of he swich, make a forward converer feasible for he oupu power (up o 150 was) of an off-line low-cos power supply. Is non-pulsaing oupu inducor curren makes he forward converer well suied for he applicaion involving a very high load curren (>15A). The presence of he oupu inducor limis he use of a forward converer in a high oupu volage (>30V) applicaion, which requires a bulky inducor o oppose he high oupu volage. INCREASING EFFICIENCY The efficiency of a forward converer is low compared o oher opologies wih he same oupu power, due o he presence of four major loss elemens: he swich, ransformer, oupu diode recifiers and oupu inducor. To increase efficiency, a synchronous MOSFET can be used in place of he oupu diode recifier. The MOSFET can be self-driven hrough he exra or he same windings in he ransformer secondary, as shown in Figure 7. FIGURE 7: D Q 1 G Q 2 G SYNCHRONOUS RECTIFIER S D S Improving he load ransien response and implemening curren mode conrol requires reducing he oupu inducor value and he use of a beer oupu capacior o mee he oupu volage ripple requiremen, as discussed in he Buck Converer secion. A muliple oupu, forward converer coupled inducor is used o ge beer cross-load regulaion requiremens. Two-Swich Forward Converer The maximum volage sress of he swich in a forward converer can be limied o a value equal o he inpu volage, by placing one more swich (Q2) in series wih he ransformer primary winding, as shown in Figure 8. The resuling converer is called a wo-swich forward converer. The basic schemaic and swiching waveforms of he wo-swich forward converer are shown in Figure 8. The swiches Q1 and Q2 are conrolled by he same gae drive signal, as shown in Figure 8 (B and C). In he seady sae of operaion, when he swiches Q1 and Q2 are ON for a TON period, he inpu volage VIN is applied o he ransformer primary. During he TON period, he magneizing curren plus he refleced oupu inducor curren flows hrough he ransformer primary and he swiches Q1 and Q2. A he end of he ON period, when he swiches are urned OFF, he flux sored inside he magneic core induces a volage in he reverse direcion o he ransformer primary winding, which forward-biases he diodes D1 and D2, and provides a pah o he magneizing curren o rese he core. The volage VIN is applied across he ransformer primary winding in he reverse direcion, as shown in Figure 8 (D). If here is no leakage inducance in he ransformer T1, he volage across NP would be equal o VIN, and he maximum blocking volage across he swich is VIN. When he magneizing curren reaches zero, diodes D1 and D2 become reverse-biased and remain zero for he res of he swiching period. The secondary side operaion of he wo-swich forward converer is he same as he operaion of he forward converer explained earlier. APPLICATION CONSIDERATIONS Reducion in he blocking volage of he swich allows he designer o selec a beer low-volage MOSFET for he design. Therefore, he wo-swich forward converer can be used up o he oupu power level of 350 was. If peak curren is greaer han 350 was, losses across he MOSFET become impracical o handle, and incomplee uilizaion of magneic makes he ransformer bulky (see Figure 9). Therefore, he wo-swich forward converer is bes suied for applicaions wih an oupu power level range of 150 o 350 was. DS01114A -page Microchip Technology Inc.

11 FIGURE 8: TWO-SWITCH FORWARD CONVERTER + D 1 Q 2 D D 3 I L (A) V P N P N S + V - + L D 4 V OUT - D 2 Q 1 D - T S (B) Q 1PWM T OFF (C) Q 2PWM (D) V P VIN I N (E) V P I P (F) (A) = Two-swich forward converer power circui (B) = Gae pulse for MOSFET Q 1 (C) = Gae pulse for MOSFET Q 2 (D) = Volage across he primary winding N P (E) = Curren hrough he primary winding N P (F) = Volage across he MOSFET Q 1 and Q Microchip Technology Inc. DS01114A -page 11

12 FIGURE 9: TRANSFORMER BH CURVE OF SINGLE SWITCH CONVERTER B SAT B EQUATION 12: FLYBACK CONVERTER VOUT/VIN RELATIONSHIP V OUT N S D = V IN N P ( 1 D) where: D = he duy cycle of he flyback swich Flyback Converer (FBT) A flyback converer (FBT) is a ransformer-isolaed converer based on he basic buck boos opology. The basic schemaic and swiching waveforms are shown in Figure 10. In a flyback converer, a swich (Q1) is conneced in series wih he ransformer (T1) primary. The ransformer is used o sore he energy during he ON period of he swich, and provides isolaion beween he inpu volage source VIN and he oupu volage VOUT. In a seady sae of operaion, when he swich is ON for a period of TON, he do end of he winding becomes posiive wih respec o he non-do end. During he TON period, he diode D1 becomes reverse-biased and he ransformer behaves as an inducor. The value of his inducor is equal o he ransformer primary magneizing inducance LM, and he sored magneizing energy (see Equaion 11) from he inpu volage source VIN. Therefore, he curren in he primary ransformer (magneizing curren IM) rises linearly from is iniial value I1 o IPK, as shown in Figure 10 (D). As he diode D1 becomes reverse-biased, he load curren (IOUT) is supplied from he oupu capacior (CO). The oupu capacior value should be large enough o supply he load curren for he ime period TON, wih he maximum specified droop in he oupu volage. ΔB H A he end of he TON period, when he swich is urned OFF, he ransformer magneizing curren coninues o flow in he same direcion. The magneizing curren induces negaive volage in he do end of he ransformer winding wih respec o non-do end. The diode D1 becomes forward-biased and clamps he ransformer secondary volage equal o he oupu volage. The energy sored in he primary of he flyback ransformer ransfers o secondary hrough he flyback acion. This sored energy provides energy o he load, and charges he oupu capacior. Since he magneizing curren in he ransformer canno change insananeously a he insan he swich is urned OFF, he primary curren ransfers o he secondary, and he ampliude of he secondary curren will be he produc of he primary curren and he ransformer urns raio, NP/NS. DISSIPATING STORED LEAKAGE ENERGY A he end of he ON period, when he swich is urned OFF, here is no curren pah o dissipae he sored leakage energy in he magneic core of he flyback ransformer. There are many ways o dissipae his leakage energy. One such mehod is shown in Figure 10 as a snubber circui consising of D2, RS and CS. In his mehod, he leakage flux sored inside he magneic core induces a posiive volage a he non-do end primary winding, which forward-biases he diode D2 and provides he pah o he leakage energy sored in he core, and clamps he primary winding volage o a safe value. During his process, CS is charged o a volage slighly more han he refleced secondary flyback volage, which is known as flyback overshoo. The spare flyback energy is dissipaed in resisor RS. In a seady sae, and if all oher condiions remain consan, he clamp volage is direcly proporional o RS. The flyback overshoo provides addiional forcing vols o drive curren ino he secondary leakage inducance during he flyback acion. This resuls in a faser increase in he ransformer secondary curren, which improves he efficiency of he flyback ransformer. CONTINUOUS CONDUCTION MODE The waveform shown in Figure 10 (D) represens Coninuous Conducion mode operaion of a flyback converer. Coninuous Conducion mode corresponds o he incomplee demagneizaion of he flyback ransformer core. The core flux increases linearly from DS01114A -page Microchip Technology Inc.

13 he iniial value flux (0) o flux (PK) during he ON period, TON. In a seady sae, he change in core flux during he TON period should be equal o he change in flux during he TOFF period. This is imporan o avoid sauraion. The relaion beween he inpu and oupu volage in a seady sae and coninuous mode of operaion is given by Equaion 12. FIGURE 10: FLYBACK CONVERTER + R S C S D 1 I OUT (A) D2 V P N S NP I D1 V OUT I SW Q 1 D - (B) Q 1PWM T ON T OFF T S (C) V P V CLAMP I PK (D) I SW I 1 (E) I D1 N P N S I PK (F) + V CLAMP (A) = Flyback converer power circui (B) = Gae pulse for he MOSFET Q 1 (C) = Volage across he primary winding (D) = Curren hrough MOSFET Q 1 (E) = Curren hrough he diode D 1 (F) = Volage across he MOSFET Q Microchip Technology Inc. DS01114A -page 13

14 During Coninuous Conducion mode of operaion, he duy cycle is independen of he load drawn from he converer, and is a consan for he DC inpu volage. However, in a pracical siuaion he load increases he loss inside he ransformer and he oupu diode D2 loss is also increased. To mainain consan oupu volage, he duy cycle varies slighly in Coninuous Conducion mode a a consan DC inpu volage. Because of he presence of he secondary refleced volage on he primary winding and he leakage sored energy in he ransformer core, he maximum volage sress VDS of he swich is given by Equaion 13. If he flyback converer is used for universal inpu of he off-line power supply, he swich volage raing should be 700V, considering he secondary refleced volage of 180V and 20% vols of leakage spike due o leakage energy sorage in he ransformer. residual flux densiy, BR, as shown in Figure 11. Therefore, he air gap increases he working range of dela BH o increase he hroughpu of he flyback ransformer. FIGURE 11: ΔBAC B BSAT BH CURVE WITH AIR GAP FOR THE FLYBACK TRANSFORMER EQUATION 13: MAXIMUM VDS IN FLYBACK CONVERTER V DS = + V CLAMP + V LEAKAGE where: V CLAMP = Volage across he snubber circui (D 2, R 2, and C 2 ) V LEAKAGE = Leakage spike volage due o leakage energy ΔH wihou air gap ΔH (air gap) H SELECTING A CAPACITOR The pulsaing curren ID1, as shown in Figure 10(E), flows in, and he DC load curren flows ou of he oupu capacior, which causes he oupu capacior of he flyback converer o be highly sressed. In he flyback converer, he selecion of he oupu capacior is based on he maximum ripple RMS curren seen by he capacior given by Equaion 6, and he maximum peak-o-peak oupu volage ripple requiremens. The oupu volage peak-o-peak ripple depends on he ripple curren seen in he capacior and is Equivalen Series Resisor (ESR). The ESR of he capacior and he ripple curren cause heaing inside he capacior, which affecs is predicive life. Therefore, selecion of he capacior depends highly on he ripple curren raing and he ESR value so as o mee he emperaure rise and oupu volage ripple requiremen. If he oupu ripple curren is high, i is advisable o have more han one capacior in parallel in place of a single, large capacior. These capaciors should be placed a an equal disance from he diode cahode erminal, so ha each capacior shares equal curren. AIR GAP To increase he hroughpu capabiliy and reduce he chances of magneic sauraion in he flyback ransformer core, an air gap is insered in he limb of he ransformer core. This air gap doesn' change he sauraion flux densiy (BSAT) value of he core maerial; however, i increases he magneic field inensiy, H, o reach sauraion and reduces he ADVANTAGES OF FLYBACK TOPOLOGY Flyback opology is widely used for he oupu power from a maximum of a 5 o150 wa low-cos power supply. Flyback opology doesn use an oupu inducor, hus saving cos and volume as well as losses inside he flyback converer. I is bes suied for delivering a high oupu volage up o 400V a a low oupu power up o was. The absence of he oupu inducor and he freewheeling diode (used in he forward converer) makes he flyback converer opology bes suied for high oupu volage applicaions. In a flyback converer, when more han one oupu is presen, he oupu volages rack one anoher wih he inpu volage and he load changes, far beer han hey do in he forward converer. This is because of he absence of he oupu inducor, so he oupu capacior connecs direcly o he secondary of he ransformer and acs as a volage source during he urned off period (TOFF) of he swich. APPLICATION CONSIDERATIONS For he same oupu power level, and if he oupu curren requiremen is more han amps, he RMS peak-o-peak ripple curren seen by he oupu capacior is very large, and becomes impracical o handle. Therefore, i is beer o use he forward converer opology han he flyback opology for an applicaion where he oupu curren requiremen is high. DS01114A -page Microchip Technology Inc.

15 Push-Pull Converer A push-pull converer is a ransformer-isolaed converer based on he basic forward opology. The basic schemaic and swiching waveforms are shown in Figure 12. The high-volage DC is swiched hrough he cener-apped primary of he ransformer by wo swiches, Q1 and Q2, during alernae half cycles. These swiches creae pulsaing volage a he ransformer primary winding. The ransformer is used o sep down he primary volage and o provide isolaion beween he inpu volage source VIN and he oupu volage VOUT. The ransformer used in a push-pull converer consiss of a cener-apped primary and a cener-apped secondary. The swiches Q1 and Q2 are driven by he conrol circui, such ha boh swiches should creae equal and opposie flux in he ransformer core Microchip Technology Inc. DS01114A -page 15

16 In he seady sae of operaion, when Q1 is ON for he period of TON, he do end of he windings become posiive wih respec o he non-do end. The diode D5 becomes reverse-biased and he diode D6 becomes forward-biased. Thus, he diode D6 provides he pah o he oupu inducor curren IL hrough he ransformer secondary NS2. As he inpu volage VIN is applied o he ransformer primary winding NP1, a refleced primary volage appears in he ransformer secondary. The difference of volages beween he ransformer secondary and oupu volage VOUT is applied o he inducor L in he forward direcion. Therefore, he inducor curren IL rises linearly from is iniial value of IL1 o IL2, as shown in Figure 12(E). During his TON period while he inpu volage is applied across he ransformer primary NP1, he value of he magneic flux densiy in he core is changed from is iniial value of B1 o B2, as shown in Figure 13. FIGURE 12: PUSH-PULL CONVERTER D 6 I L + L - I OUT V OUT N P2 N S2 + (A) N P1 N S1 D 5 Q 2 D Q 1 D - Q 1PWM T ON T OFF (B) T S /2 Ts Q 2PWM V DS1 (C) I IN IQ 1 IQ 2 IQ 1 IQ 2 (D) V DS2 (E) I L I L2 I L1 (A) = Push-pull converer (B) = Gae pulse of MOSFET Q 1 (C) = Drain-o-source volage Vds of MOSFET Q 1 (D) = Curren hrough he MOSFET Q 1 and Q 2 (E) = Oupu inducor curren DS01114A -page Microchip Technology Inc.

17 A he end of he TON period, he swich Q1 is urned OFF, and remains off for he res of he swiching period TS. The swich Q2 will be urned ON afer half of he swiching period TS/2, as shown in Figure 12. Thus, during he TOFF period, boh of he swiches (Q1 and Q2) are OFF. When swich Q1 is urned OFF, he body diode of he swich provides he pah for he leakage energy sored in he ransformer primary, and he oupu recifier diode D5 becomes forward-biased. As he diode D5 becomes forward-biased, i carries half of he inducor curren hrough he ransformer secondary NS1, and half of he inducor curren is carried by he diode D6 hrough he ransformer secondary NS2. This resuls in equal and opposie volages applied o he ransformer secondaries, assuming boh secondary windings NS1 and NS2 have an equal number of urns. Therefore, he ne volage applied across he secondary during he TOFF period is zero, which keeps he flux densiy in he ransformer core consan o is final value B2. The oupu volage VOUT is applied o he inducor L in he reverse direcion when boh swiches are OFF. Thus, he inducor curren IL decreases linearly from is iniial value of IL2 o IL1, as shown in Figure 12 (E). AVOIDING MAGNETIC SATURATION Afer he ime period TS/2, when he swich Q2 urns ON, he diode D6 become reverse-biased, and he complee inducor curren sars flowing hrough he diode D5 and ransformer secondary NS1. During his TON period, when he swich Q2 is urned ON, he inpu volage VIN is applied o he ransformer primary NP2 in he reverse direcion, which makes he do end negaive wih respec o he non-do end. As he inpu volage applies across he ransformer primary NP2, he value of he magneic flux densiy in he core is changed from is iniial value of B2 o B1, as shown in Figure 13. Assuming he number of primary urns NP1 is equal o NP2, and he number of secondary winding urns NS1 is equal o NS2, he TON period of boh swiches should be he same o avoid magneic sauraion in he ransformer core. Afer he TON period, Q2 urns OFF and remains off for he res of he period TS, as shown in Figure 12. FIGURE 13: BH CURVE FOR PUSH-PULL TRANSFORMERVOLTAGE VOLTAGE RATING OF SWITCH During he TON period of any swich, he volage VIN is applied o half of he ransformer primary and induces equal volage o he oher half of he ransformer primary winding. This resuls in wice he inpu volage applied o he off swich. Therefore, he swiches used for he push-pull converer mus be raed a leas wice he maximum inpu volage. For pracical purposes, he volage raing of he swich should be 20% more han he heoreical calculaion due o leakage spike and ransiens. For he universal inpu volage, he raing of he swich used should be: = 895, which means a 900 vol swich is required. VOUT/VIN RELATIONSHIP In he seady sae and Coninuous Conducion mode of operaion, he relaion beween he inpu and oupu volage is given by Equaion 14, where D is he duy cycle of he swich. EQUATION 14: B SAT B 2 B B SAT B 1 ΔB PUSH-PULL CONVERTER VOUT/VIN RELATIONSHIP H N S V OUT = D N P T ON T S D = Microchip Technology Inc. DS01114A -page 17

18 REDUCING MAGNETIC IMBALANCE If he flux creaed by boh primary windings is no equal, a DC flux is added a every swiching cycle and will quickly saircase o sauraion. This magneic imbalance can be caused by an unequal TON period for boh swiches, an unequal number of urns of he primary NP1 and NP2 and he secondary NS1 and NS2, and an unequal forward volage drop of he oupu diodes D5 and D6. This imbalance can be reduced by careful selecion of he gae pulse drive circuiry, using a swiching device ha has a posiive emperaure co-efficien (PTC) for he ON sae resisance, adding air gap o he ransformer core, and using peak curren mode conrol echniques o decide he TON period of he swiches Q1 and Q2. Figure 14 explains how o deermine he saus of magneics imbalance in he core during he seady sae of operaion by looking a curren waveforms of he wo swiches Q1 and Q2. If he curren wave shape of boh swiches is symmerical and equal in magniude, as shown in Figure 14 (A), he flux excursion in he core is well balanced and he ransformer is operaing in a safe region. However, if he curren wave shape of boh swiches is no symmerical and he peak magniude curren is no equal, as shown in Figure 14 (B), here is an imbalance in he flux excursion inside he core; however, i is sill operaing a he safe operaing region of he BH loop. If he curren wave shape of one of he swiches has upward concaviy, as shown in Figure 14 (C), his means here is a large inequaliy in he flux excursion inside he magneic core, and magneic BH loop is close o sauraion. A small increase in he magneic field inensiy H will cause a decrease in magneizing inducance, whereas a significan increase in magneizing curren can desroy he swich and he ransformer. FLUX DOUBLING AND VOLT-SECOND CLAMPING When such a sysem is firs swiched ON or during he load ransien, he flux densiy will sar from zero raher han B1 or B2, and consequenly, he available flux excursion a his insan will be half ha normally available under he seady sae condiion. This is called flux doubling. The drive and conrol circuiry mus recognize his condiion and proec he applicaion from wide drive pulses unil he normal working condiion of he core is resored. This is known as vol-second clamping. COPPER UTILIZATION A push-pull ransformer requires a cener apped primary, and each winding is acive only for alernae power pulses, which means only 50% uilizaion of primary copper. The unused copper occupies space in he bobbin and increases he primary leakage inducance. A cener-apped primary would normally be bifilar wound, bu his will cause a large AC volage beween he adjacen urns. APPLICATION CONSIDERATIONS The high volage (2 VIN) sress on he swich, and 50% uilizaion of he ransformer primary makes using he push-pull opology undesirable when he inpu volage is European, Asian, he universal range (90 VAC-230 VAC), or when PFC is used as he fron end recifier. The reason for his is incomplee uilizaion of magneic core, which is due o only one swich conducing during each swiching cycle and full inpu volage is applied across he ransformer primary. The push-pull opology is mos favorable for low-volage applicaions such as US regulaion 110 VAC inpu direc off-line SMPS, or low inpu volage DC-DC isolaed converer for he power raing of up o 500 was. FIGURE 14: (A) PUSH-PULL CONVERTER SWITCH CURRENT IQ 1 IQ2 IQ 1 IQ 2 Q 1ON Q 2ON Q 1ON Q 2ON (B) (C) Sauraion (A) = Equal vol second is applied across he primary (B) = Unequal vol second applied across he primary bu sill in safe region (C) = Highly unbalance vol second applied across he secondary and core is near o sauraion DS01114A -page Microchip Technology Inc.

19 AVOIDING SHOOT-THROUGH In a push-pull converer, boh swiches canno urn ON a he same ime. Turning boh swiches on a he same ime will generae an equal and opposie flux in he ransformer core, which resuls in no ransformer acion and he windings will behave as if hey have a shor. This condiion offers a very low impedance beween he inpu source VIN and ground, and here will be a very large shoo-hrough curren hrough he swich, which could desroy i. To avoid shoo-hrough, an inducor is placed beween he ransformer primary and he inpu supply, as shown in Figure 15. The resuled converer is known as a curren-source push-pull converer. When boh swiches are on, he volage across he primary becomes zero and he inpu curren builds up and energy is sored in he inducor. When only one of he wo swiches is ON, he inpu volage and sored energy in he inducor supplies energy o he oupu sage. The relaion beween he oupu and inpu in Coninuous Conducion mode is given by Equaion 15. EQUATION 15: FIGURE 15: + - Q 2 V OUT D Half-Bridge Converer CURRENT SOURCE PUSH-PULL CONVERTER VOUT/VIN RELATIONSHIP N S = ( 1 D) N P CURRENT FED PUSH-PULL CONVERTER Q 1 N P2 N P1 N S2 N S1 V OUT The half-bridge converer is a ransformer-isolaed converer based on he basic forward opology. The basic schemaic and swiching waveforms are shown in Figure 16. D D 6 D 5 I OUT N P1 = N P2 = N P N S1 = N S2 = N S The swiches Q1 and Q2 form one leg of he bridge, wih he remaining half being formed by he capaciors C3 and C4. Therefore, i is called a half-bridge converer. The swiches Q1 and Q2 creae pulsaing AC volage a he ransformer primary. The ransformer is used o sep down he pulsaing primary volage, and o provide isolaion beween he inpu volage source VIN and he oupu volage. In he seady sae of operaion, capaciors C3 and C4 are charged o equal volage, which resuls in he juncion of C3 and C4 being charged o half he poenial of he inpu volage. When he swich Q1 is ON for he period of TON, he do end of he primary connecs o posiive VIN, and he volage across he capacior C4 (VC4) is applied o he ransformer primary. This condiion resuls in half of he inpu volage being VIN, which is applied o he primary when he swich Q1 is ON, as shown in Figure 16 (C). The diode D4 becomes reverse-biased, and he diode D3 becomes forward-biased, which carry he full inducor curren hrough he secondary winding NS1. The difference of he primary volage refleced on he secondary NS1 and oupu volage VOUT is applied o he oupu inducor L in he forward direcion. Therefore, he inducor curren IL rises linearly from is presen value of IL1 o IL2, as shown in Figure 16 (E). During his TON period, he refleced secondary curren, plus he primary magneizing curren flows hrough he swich Q1. As he volage is applied o he primary in he forward direcion during his TON period, and when he swich Q1 is ON, he flux densiy in he core changes from is iniial value of B1 o B2, as shown in Figure 13. A he end of he TON period, he swich Q1 urns OFF, and remains off for he res of he swiching period TS. The swich Q2 will be urned ON afer half of he swiching period TS/2, as shown in Figure 16 (B); herefore, during he TOFF period, boh swiches are off. When swich Q1 is urned off, he body diode of he swich Q2 provides he pah for he leakage energy sored in he ransformer primary, and he oupu recifier diode D4 becomes forward-biased. As he diode D4 become forward-biased, i carries half of he inducor curren hrough he ransformer secondary NS2 and half of he inducor curren is carried by he diode D3 hrough he ransformer secondary NS1, as shown in Figure 16 (E). Therefore, he equal and opposie volage is applied a he ransformer secondary, assuming boh secondary windings NS1 and NS2 have an equal number of urns. As a resul, he ne volage applied across he secondary during he TOFF period is zero, which keeps he flux densiy in he ransformer core consan o is value of B2. The oupu volage VOUT is applied o he inducor L in he reverse direcion when boh swiches are OFF. Therefore, he inducor curren IL decreases linearly from is iniial value of IL2 o IL1, as shown in Figure 16 (E). The body diodes of swiches Q1 and Q2 provide he pah for he ransformer leakage energy Microchip Technology Inc. DS01114A -page 19

20 Afer he ime period TS/2 when he swich Q2 urns ON, he do end of he primary connecs o he negaive of VIN, and he volage across he capacior C3 (VC3) is applied o he ransformer primary. Therefore, half of he inpu volage VIN is applied o he primary when he swich Q2 is ON in he reverse direcion, as shown in Figure 16 (C). The value of he magneic flux densiy in he core is changed from is iniial value of B2 o B1, as shown in Figure 13. Assuming he number of secondary winding urns of NS1 is equal o NS2, and o avoid magneic sauraion in he ransformer core, he TON period of boh swiches should be he same. Afer he TON period, Q2 urns OFF and remains off for he res of he period TS, as shown in Figure 16 (B). Please noe ha when eiher of he swiches urn ON for he TON period, i affecs he enire inpu volage VIN of he oher swich. FIGURE 16: HALF-BRIDGE CONVERTER + IQ 1 (A) V C4 C 4 C B Q 1 IQ 2 + N P V P - D 3 N S1 N S2 I L + L - IOUT V OUT V C3 C 3 Q 2 D 4 - T S Q 1PWM T ON T OFF (B) Q 2PWM (C) V P /2 (D) I SW IQ 1 IQ 2 IQ 1 IQ 2 I L2 (E) I L I L1 I D4 (A) = Half-Bridge Converer (B) = Gae pulse waveform of Q 1 (C) = Volage across ransformer primary (D) = Curren hrough he swich Q 1 and Q 2 (E) = Oupu inducor and diode D 4 curren DS01114A -page Microchip Technology Inc.

21 EQUIVALENT TRANSFORMER The equivalen ransformer model is shown in Figure 17. During he TOFF period, when boh swiches are OFF, ideally, he secondary currens flowing hrough he diode D3 and he diode D4 should be equal. However, in he pracical sense, because of he presence of he non-zero magneizing curren IM, ID3 and ID4 are no equal. This magneizing curren IM(), as shown in Figure 17, may flow hrough he ransformer primary, hrough one of he secondaries, or i may divide beween all hree of he windings. FIGURE 17: I 1 () + V P - I M () I 1 () TRANSFORMER EQUIVALENT MODEL The division of he magneizing curren depends on he I-V characerisics of he swiches, he diode and he leakage of he ransformer windings. Assuming negligible leakage in he ransformer and ha boh diodes have similar I-V characerisics, he curren flowing hrough he diode D3 and D4 is given by Equaion 16. EQUATION 16: N P N S2 N S1 N S = N S1 = N S2 Transformer Equivalen Model I 1 = 0 = I D3 for I M ()<< i() D 3 i d3 I D4 D 4 OUTPUT DIODES AND MAGNETIZING CURRENT RELATIONSHIP 0.5 i () ( 0.5 n) I M () I D4 = 0.5 i () ( 0.5 n) I M () I D3 = I D4 = 0.5 i () DC BLOCKING CAPACITOR A small DC blocking capacior is placed in series wih he ransformer primary, o block he DC flux in he ransformer core. The value of he DC blocking capacior is given by Equaion 17. EQUATION 17: DC BLOCKING CAPACITOR I C PRIM T ONMAX B = ΔV where: T ONMAX = maximum ON ime of eiher MOSFET I PRIM = maximum primary curren ΔV = permissible droop in primary volage because of he DC blocking capacior PREVENTING SHOOT-THROUGH A half-bridge converer is also prone o magneic imbalance of he ransformer core when he flux creaed by he swiches Q1 and Q2 during he TON period is no equal. To preven saircase sauraion, he peak curren mode conrol echnique is used o decide he TON period of he swiches Q1 and Q2. The maximum duy cycle of 45% wih a dead-ime beween he wo swiches is used o preven shoo-hrough curren from he ransformer primary. APPLICATION CONSIDERATIONS The complee uilizaion of he magneic and maximum volage sress on eiher of he swiches is equal o he inpu volage VIN. However, only half of he inpu volage is applied across he primary when eiher of he swiches is ON for he TON period. Therefore, double he primary swich curren is required o have he same oupu power as he push-pull converer. This makes he half-bridge opology bes suied for applicaions up o 500 was. This is especially suied for European and Asian regions where he AC is 230 VAC line volage. The power raing of he half-bridge converer can be increased up o was if fron-end PFC is used. The peak primary curren and he maximum ransien OFF sae volage sress of he swich deermine he pracical maximum available oupu power in he half-bridge converer opology Microchip Technology Inc. DS01114A -page 21