A high efficiency and power factor, segmented linear constant current LED driver

Transcription

1 Vol. 36, No. 4 Journal of Semiconductors April 2015 A high efficiency and power factor, segmented linear constant current LED driver Li Yongyuan( 励 勇 远 ), Guo Wei( 过 伟 ), and Zhu Zhangming( 朱 樟 明 ) School of Microelectronics, Xidian University, Xi an , China Abstract: A high efficiency, high power factor, and linear constant current LED driver based on adaptive segmented linear architecture is presented. When the input voltage varied, the proposed LED driver automatically switched over LED strings according to the segmented LED voltage drop, which increased the LED lighting time. The efficiency and power factor are improved, while the system design is simplified by this control scheme. Without the usage of electrolytic capacitor and magnetic components, the proposed driver possesses advantages of smaller size, longer lifetime and lower cost over others. The proposed driver is implemented in 0.8 m 5 V/40 V HVCMOS process, which occupies an active area of m 2. The measured results show that the average value of the internal reference voltage is mv, with a standard deviation of only mv, thus LED current can be set accurately. Under 220 V root mean square 50 Hz utility voltage and the number ratio of the three LED strings being 47 : 17 : 16, the system can realize a high power factor of and power conversion efficiency of 93.4%. Key words: high efficiency; high power factor; segmented; linear constant; LED driver DOI: / /36/4/ EEACC: Introduction LEDs lighting has a broad prospect for its long lifetime, environmental friendliness and high lighting efficiency, which has achieved a premier position in a new generation of lighting systems. Generally, there are two approaches to realize a constant current in LEDs. The first approach is the switchmode constant current source Œ1 3, which regulates the current by peak current control (PCC) or hysteretic current control (HCC). These methods have advantages of high accuracy, low output current ripple and high efficiency. But the circuit complexity will greatly increase the cost. Moreover, an electrolytic capacitor and inductor are inevitably used in those topologies to reduce lifetime and increase the system area. The second approach is the linear constant current source to regulate the current by a negative feedback control Œ4 6, shaping the output current in proportion to the input voltage. Compared with the switch-mode constant current source, the linear constant current source can accurately control the output current and achieve better consistency. It is particularly important that the linear constant current mode solves the problem of flicker, which has become the main scheme for low cost, low power and flicker-free LEDs lighting. Therefore, it is of great value to carry out research on the linear constant current LED driver. An LED driver consists of a main controller and constant current sources in Reference [5], which samples the input voltage directly, controls switches to light the LED string step by step, and shapes the output current in proportion to the input voltage. However, the maximum power conversion efficiency is 90.68%. In Reference [7], the controller can switch over two different output currents to improve the power factor, which realizes a low cost linear constant current LED driver. However, the maximum power factor is just 95% and the output current is not precise enough. In this paper, a high efficiency, high power factor (PF), and segmented linear constant current LED driver is presented. The number of turn-on LEDs is varied in accordance with rectified input voltage, which shapes the output current in proportion to the input voltage and increases the LED lighting time. The output current accuracy can be controlled strictly by precise internal reference voltage. 2. Proposed control scheme Figure 1 shows the schematic of the control scheme. Figure 1(a) is a system diagram of the proposed linear constant current LED driver. The proposed LED driver drives N LED strings and each LED string consists of a certain number of LEDs in series. I LED is a voltage-controlled current source, which is controlled by input voltage V REC. The typical waveform of the input voltage V REC and LED current I LED is shown in Figure 1(b). In order to make the switches turn on, the input voltage V REC should satisfy the condition as follows: where V LN can be derived as V REC > V LN ; (1) V LN D.N 1 C N 2 C C N i /V F ; (2) where N i represents the number of LEDs in series of each LED string in line i. V F is the forward voltage of LED. During a whole period, the input voltage V REC rises from 0 to the peak voltage V peak, the output current can be divided into N stages. In low AC input condition, V REC does not satisfy Equation (1), hence power transistors S 1 S N are all cut-off. S 1 will turn on when V REC rises to V L1, which is controlled to turn the LED1 string on. If V REC rises to V L2, then S 1 turns off and S 2 turns on * Project supported by the National Natural Science Foundation of China (Nos , , , ). Corresponding author. [email protected] Received 24 November 2014, revised manuscript received 12 January Chinese Institute of Electronics

2 Figure 1. Schematic of the control scheme. to turn the LED1 C LED2 strings on. So power transistors from S 3 to S N turn on in turn as V REC increases, shaping the LED current I LED in proportion to the input voltage. This control scheme minimizes the power loss and improves efficiency. If the V REC decreases, the operation process is contrary to what has been described above. As shown in Figure 1(a), where i.t/ and V.t/ are the input current and input voltage, respectively, it is assumed that V.t/ is a distortion-free input voltage which can be expressed as V.t/ D V sin.!t/; (3) where V is the peak value of the input voltage;! is the angular frequency of the input voltage. Because the diode in the rectifier circuit is nonlinear, input current is a non-sinusoidal periodic, which can be represented by Fourier series: i.t/ D X n I n sin.n!t C a n /; n D 1; 2; ; (4) where I n is the amplitude of nth-order component, a n is the phase shift of nth-order component. According to the PF definition from Reference [8], the associated PF can be described as PF D.I i1 =I i / cos a 1 ; (5) where I i and I i1 are the root mean square (RMS) value of input current and fundamental input current, respectively. I i1 /I i is expressed as a distortion degree of the current waveform, which can be measured by the total harmonic distortion, referred to as the distortion factor. a 1 is the phase angle between the fundamental input current and sinusoidal input voltage; cos a 1 is the displacement factor. According to the presented scheme: the number of turn-on LEDs is varied in accordance with rectified input voltage V REC, and shapes the LED current I LED in proportion to the input voltage V REC. So the more LED segments, the higher PF and efficiency are. However, the cost increases as the LED segment numbers rise. Therefore, the proposed LED driver is designed with 3 LED strings. 3. The proposed LED driver Figure 2 shows the system diagram of the proposed segmented linear constant current LED driver. V CC is the internal power supply voltage, which is produced by V DD. UVLO is the output signal of the undervoltage lockout (UVLO) circuit. V BG is reference voltage, which is produced by bandgap reference circuit. Buffer is used to increase the load ability and generate voltage reference V TH of 1 V. Operational amplifier OP1, transistor N 1 and resistor R 0 constitute a voltage follower configuration to make V X D V TH. So the current through R 0 is V TH /R 0. Transistor N 4 turns on when the over-temperature protection (OTP) occurs. Then N 4 extracts a fraction of the current to reduce output current. Hence the rise of temperature is limited. The reference voltage circuit is shown in Figure 2. Y N is the aspect ratio of the transistor P N /N N. During the normal operation, N 4 is cut-off. V Y D 2V REF D V X D V TH due to Y 1 : Y 2 D 1 : 1, Y 3 : Y 4 D 1 : 1 and R 0 : R 2 : R 3 D 2 : 1 : 1. The value of V REF is 500 mv as an internal reference voltage. Once the system is powered up, the input voltage V REC is very low. Because S 1, S 2 and S 3 are all cut-off, there is no feedback voltage at CS PIN. When V REC rises to V L1, S 1 will turn on to turn the LED1 string on and CS feeds back into the chip. CS and FB1 are the input and output of the feedback loop 1 in Figure 2, respectively. FB1 and CS are both the input signals of the feedback loop 2 while FB2 is the output signal in the feedback loop 2. The principle of the feedback loop 1 and the feedback loop 2 is similar to that of the reference voltage circuit. Because Y 11 : Y 12 D 1 : 1, Y 13 : Y 14 D 1 : 1, Y 6 : Y 7 D 1 : 1, Y 8 : Y 9 D 1 : 2, R 8 : R 10 D 2 : 1, and R 4 : R 6 D 3 : 4, FB1 and FB2 are equal to 1.5CS and 3CS, respectively. Since the transconductance amplifier 1 (OTA1), S 1, R CS, feedback loop 1 and feedback loop 2 constitute a negative feedback loop, the LED peak current is equal to V REF /3CS. S 1 turns off and S 2 turns on when V REC reaches V L2, which is controlled to turn the LED1 C LED2 strings on. OTA2, S 2, R CS, and feedback loop 1 constitute a negative feedback loop, the LED peak current becomes V REF /1.5CS. When V REC rises to V L3, S 2 turns off and S 3 turns on. Similar to the above description, LED

3 Figure 2. System diagram of the proposed segmented linear constant LED driver. C LED2 C LED3 strings are turned on, with peak current of V REF /CS. Thus the LED peak current can be precisely set by an external resistor R CS. In order to optimize the PF and THD, the current proportion is 1 : 2 : 3 when the LED string is segmentally turned on. If V REC decreases, the operation process is contrary to what has been described above. 4. The key circuit block 4.1. Bandgap reference voltage Figure 3 shows the schematic of the bandgap reference circuit, which consists of startup circuit, 1st-order curvature compensation circuit and 3rd-order curvature compensation circuit. The startup circuit is made up of N 32 N 37, R 20 R 21 and R 26. EN is the enable signal of the bandgap reference circuit. I M and I QNB are the current, which flows through the M and base of bipolar transistor Q N, respectively. The basic principle of the bandgap reference circuit is: I R25 increases as the temperature rises, these currents transfuse into R 24 for compensating the 3rd-order negative temperature coefficient of V BE. Q 3 Q 6 and R 22 R 24 serve as a 1st-order curvature compensation circuit. Q 5, Q 6 and R 22 work as a PTAT current of the typical bandgap reference I PTAT D.V BE5 V BE6 /=R 22 D kt ln N=qR 22 ; (6) where N is the emitter area ratio of the Q 6 and Q 5 ; V XYn is the voltage difference between X and Y of Q n. Figure 3. Schematic of the bandgap reference voltage with 3rd-order curvature compensation. Since R 23 C R 24 and I Q5 C I Q6 are equal to R 27 and I R24, respectively, it can be drawn: I R27 D I Q5 C I Q6 : (7) I Q10 mirrors I PTAT as a self-biased of the whole reference circuit. Since Y 50 : Y 51 D 2 : 1, it can be seen I Q7 D I P50 I Q5 I Q6 D I Q5 C I Q6 : (8)

4 Figure 4. Schematic of the undervoltage lockout (UVLO) circuit. I Q7B flows through Q 5 while I Q3B and I Q4B flow through Q 6. Setting I Q7B D I Q3B C I Q4B to prevent the current mismatch between Q 5 and Q 6. V CE3 is clamped to V BE7 by Q 7. Therefore, V CE3 D V BE D V CE4, which avoids the current mismatch caused by the Early Effect. To sum up, I Q5 and I Q6 have a great matching capacity to improve the reference precision. Q 9 and R 23 R 25 are used as a 3rd-order curvature compensation circuit. I R23 is twice the PTAT current. At the normal temperature, because I R23 is very low, there is almost no current through Q 9. The bandgap reference circuit was only required to achieve 1st-order compensation. Q 9 turns on due to I R23 increasing as the temperature rises, then I R24 is I R24 D I R23 C I R25 D I R23 C I S9 exp V BEQ9 =V T ; (9) where V T is the thermal voltage; I S9 represents the reverse saturation current of Q 9. Then the output voltage V BG of the bandgap reference circuit is V BG D V BE5 C 2kTR 23 qr 22 ln N C R 24 2kT ln N C I R25 : qr 22 (10) Equation (10) shows that I R25 alleviates descending speed of the reference voltage by compensating the logarithmic term of V BE5 as the temperature rises, which plays a significantly effect on 3rd-order compensation Undervoltage lockout (UVLO) circuit The schematic of the UVLO circuit base on a bandgap structure is shown in Figure 4. The proposed UVLO circuit uses a current comparator to detect voltage without an extra voltage reference circuit. P 23 P 24, P 29 P 31, P 34, N 12 N 13, and C 5 C 6 serve as a startup biasing circuit. The biasing circuit of the UVLO circuit uses transistors with large channel lengths to suppress the channel-length modulation effect. The clamping circuit is composed of P 35, P 37 P 39, N 20 N 23, and C 8 C 9. P 25 P 28, P 53 and C 7 are used as a protection circuit to prevent the mal-operation from the glitch of the input voltage V REC. SMT is the Schmitt trigger. The resistive divider is made up of R 17 R 19 in series. The resistive divider, the inverter INV2, and the switches N 14 N 15 are used as hysteresis control. The emitter area of Q 2 is four times that of Q 1, and the feedback of their emitter resistors R 15 and R 16, so the equivalent transconductances of Q 1 and Q 2 are as follows: G m2 D D G m1 D g Q1 1 C g Q1 R 15 C 1 g P34 ; (11) g Q2 1 C g Q2 R 15 C R 16 C 1 g P34 g Q1 : (12) 1 C g Q1 R 15 C 1 g P34 C g Q1 R 16 C Generally, G m1 > G m2 due to g Q1 R 16 > 1. That is to say, the variable rate of I C1 is much more rapid than that of I C2 as the V REC fluctuates. The operating principle of this UVLO circuit is described as follows. When V REC is low, V A > V B due to I C1 < I C2. I C1 and I C2 increase as the input voltage V REC rises. But the curve slope of I C1 is steeper than that of I C2. So I C1 will be equal to I C2 when V REC rises, the comparator reaches the threshold voltage V THS (V A D V B /. If V REC continuously rises, the comparator changes its state (V A < V B /. Based on a typical principle

5 Figure 5. Schematic of the smoothing circuit 3. of bandgap reference, when V REC rises to make I C1 D I C2, the PTAT current I C1 or I C2 is I C1 D I C2 D V T ln 4=R 16 : (13) When the comparator reaches the threshold voltage V THS, I C7 D 2I C2 C I C3. Because P 53 always cut-off before the UVLO circuit reverses, so I C8 D 2I C2 C I C3 D I C7. Moreover P 52 and P 34 have the same ratios of width to length. As for V in, the gate-source voltage V GS52 of P 52 merely offset the gate source voltage V GS34 of P 34, which is equivalent to apply V in directly to the base of Q 1 and Q 2. Therefore, the expression for this threshold voltage V THS : V THS D V BE1 C 2 R 15 R 16 V T ln 4; (14) where V BE1 and V T show a negative and a positive temperature coefficient, respectively. So a low temperature coefficient of V THS can be obtained exactly by choosing proper ratios of R 15 to R Smoothing circuit Transient response of the linear constant current driver is dependent on the slew-rate at the gate of the power transistor. The slew-rate at the gate of the power transistor can be improved by reducing the power transistor size or increasing a tail current of the OTA. If the output current mutates, the loop circuit must be fast enough to regulate the output current. Therefore, once the power transistor is fixed, the tail current of the OTA determines the slew-rate at the gate of the power transistor. The dynamic biasing technique is used in this LED driver Œ9; 10. The bias current of the OTA is increased by smoothing circuit for a short duration to improve transient response of the LED driver. Therefore, the transient response of the LED driver can be greatly improved due to the slew-rate enhancement at the gate of the power transistor. Figure 5 shows the schematic of the smoothing circuit 3 and the structure of the three smoothing circuits is the same. When a large overshoot appears in the output current of the LED driver, P 48 cuts off due to the voltage of the CS being bigger than V REF. Thus, the drain voltage of P 48 is increased. N 24, N 26, N 28, N 30 and N 31 are turned off. I N25 is Figure 6. Die photograph of the implemented chip. increased and due to a positive feedback path through transistors N 27!P 47!P 46, I N25 will be further increased. I bias6 mirrors I N25 and adds to the tail current of the OTA. Hence, the bias current of the OTA is increased to improve the transient response of the LED driver. In a similar way, when a large undershoot occurs in the output current of the LED driver, I bias5 will be further increased due to the positive feedback path through transistors N 26!P 45!P 44. Thus, the bias current of the OTA is increased to improve the transient response of the LED driver. 5. Measured results Some experimental results are offered to verify the effectiveness of the proposed LED driver, which is implemented in 0.8 m 40 V HVCMOS process. The die photograph of the implemented chip is shown in Figure 6, which occupies an active area of m 2. Figure 7 shows the measured power factor and efficiency versus input voltage. Under 220 V RMS 50 Hz utility voltage, the number ratio of the three LED strings is 47 : 17 : 16. The input power and output power are W and 8.23 W, respectively; the system can realize a high power factor of and power conversion efficiency of 93.4%

6 Table 1. Performance summary and comparison. Design This work Reference [3] Reference [5] Reference [7] Process (m) 0.8 N/A 1 1 PF THD (%) > > 30 Efficiency (%) Output power (W) Figure 7. Measured power factor and efficiency versus input voltage. Figure 9. Waveform of I LED, G1, G2 and G3 under 220 V RMS 50 Hz utility voltage. Figure 8. Waveform of V REC and I LED under 220 V RMS 50 Hz utility voltage. The experimental waveform of V REC and I LED under 220 V RMS 50 Hz utility voltage is shown in Figure 8. During each half-wave cycle, the waveform of I LED is obviously divided into 3 steps, which is in proportion to V REC. Figure 8 shows the output current overshoot/undershoot can be significantly decreased by smoothing circuit. Figure 9 shows the experimental waveform of I LED, G1, G2 and G3 under 220 V RMS 50 Hz utility voltage. Power transistors from S 1 to S 3 turn on in turn as V REC increases, which switches over three output current I LED, respectively. The importance of the internal voltage reference is taken into account, which demands for low temperature drift, high accuracy and good consistency. Therefore, layout optimization techniques are used while the bandgap reference circuit with 3rd-order curvature compensation is proposed. Tapeout sample test result shows that this design achieves a better internal voltage reference. Figure 10. Measured accuracy of V REF in batches. Measured accuracy of V REF in batches is shown in Figure 10. The average value of the V REF is mv, with a standard deviation of only mv. R CS is an external sense resistor, which can be adjusted flexibly and easily to heat dissipation, according to the definition of the output current, which can be obtained precisely by accurate internal reference voltage. Table 1 summarizes the performance of this work and shows the comparison summary with the other designs. Under 220 V RMS 50 Hz utility voltage, the system can realize a PF of higher than 0.95 mentioned in Reference [3]. The THD of 24.68% is smaller than that achieved in Reference [7]. The power conversion efficiency of 93.4% is the biggest compared with References [3, 5, 7]

7 6. Conclusion In order to simplify system design, reduce chip cost and achieve high PF and efficiency, the linear constant current LED driver based on adaptive segmented linear architecture is presented. The driver drives three LED strings, where the number of turn-on LEDs is varied in accordance with rectified input voltage, which shapes the output current in proportion to the input voltage and increases LED lighting time to improve PF and efficiency. Without the usage of an electrolytic capacitor and magnetic components, the proposed driver possesses the advantage of smaller size, longer lifetime and lower cost over others. The 3rd-order curvature compensated bandgap circuit ensures accurate output current. The output current overshoot/undershoot at the extreme load transients are significantly reduced by smoothing circuit. The proposed linear constant current LED driver is suitable for LED lighting applications. References [1] Wu C, Hui S Y R. Elimination of an electrolytic capacitor in AC/DC light-emitting diode (LED) driver with high input power factor and constant output current. IEEE Trans Power Electron, 2012, 27(3): 1598 [2] Wang B, Ruan X, Yao, K et al. A method of reducing the peak-toaverage ratio of LED current for electrolytic capacitor-less AC DC drivers. IEEE Trans Power Electron, 2010, 25(3): 592 [3] Wang Y, Jiang J, He L. High-precision constant current controller for primary-side feedback LED drivers. IEEE 23rd International Symposium on Industrial Electronics (ISIE), 2013: 1 [4] Hu Y, Huber L, Jovanovic M M. Single-stage universal-input AC/DC LED driver with current-controlled variable PFC boost inductor. IEEE Trans Power Electron, 2012, 27(3): 1579 [5] Wang C, Xi J, He L. A linear constant current LED driver with no off-chip inductor or capacitor. IEEE 23rd International Symposium on Industrial Electronics (ISIE), 2014: 2524 [6] Park C, Rim C T. Filter-free AC direct LED driver with unity power factor and low input current THD using binary segmented switched LED strings and linear current regulator. IEEE 28th Applied Power Electronics Conference and Exposition (APEC), 2013: 870 [7] Kang E, Kim J, Oh D, et al. A 6.8-W purely-resistive AC lightingemitting diode driver circuit with 95% power factor. IEEE 8th International Conference on Power Electronics and ECCE Asia, 2011: 778 [8] Hwu K I, Tu W C. Controllable and dimmable AC LED driver based on FPGA to achieve high PF and low THD. IEEE Trans Industrial Informatics, 2013, 9(3): 1330 [9] Man T Y, Mok P K T, Chan M. A high slew-rate push pull output amplifier for low-quiescent current low-dropout regulators with transient-response improvement. IEEE Trans Circuits Syst II, Exp Briefs, 2007, 54(9): 755 [10] Or P Y, Leung K N. An output-capacitorless low-dropout regulator with direct voltage-spike detection. IEEE J Solid-State Circuits, 2010, 45(2):