Installation Considerations for Multi - Motor AC Drives & Filters Used In Metal Industry Applications

Transcription

1 Installation Considerations for Multi - Motor AC Drives & Filters Used In Metal Industry Applications G. Skibinski*, D. Dahl*, K. Pierce*, R. Freed**, and D. Gilbert** *Rockwell Automation 6400 W. Enterprise Drive Mequon, WI (414) (414) fax Abstract- This paper investigates the zero sequence current that develops in the ground circuit of a multi-motor drive application. The motivation was to reduce the magnitude of common mode electrical noise in the system ground grid and to reduce occurrence of ground fault sensor trips as a result of this noise. The phenomenon of line to ground cable charging current (I lg ) during drive switching is a main component of zero sequence current and is discussed with basic equations describing its magnitude. Results of a high frequency characterization study of all components in the zero sequence path was required to obtain estimates of surge impedance for these equations. Measured surge impedance was also used in a simulation program designed to predict I lg magnitude for various system conditions. A PWM output filter and an input isolation transformer with the neutral high resistance grounded were two solutions investigated to reduce the zero sequence current that were both simulated and measured on site. I. INTRODUCTION Over the years, the metals industry has found that a single Variable Frequency ac Drive (VFD) synchronously powering multiple ac induction motors is both an economical and flexible configuration for steel processing requirements. In a typical application, a single high horsepower ac drive may synchronously power up to 163 low hp induction motors on the conveyer of a 400 foot long steel process tunnel furnace, as shown in Fig. 1 and Fig. 2. The use of drive output contactors on each motor allows flexibility in selecting the number of motors required for a given iron slab run through the 400 ft long furnace. In normal operation, each 250 hp drive may use 30 to 80 motors to control a slab in a particular section of the oven. The slab is transferred to the next drive section and caught on the fly where its particular speed and temperature may be readjusted for steel processing requirements. The output contactor switching arrangement also provides process uptime and redundancy in the event of drive, cable or motor failure. A failed cable or motor may be quickly switched out, while a failed drive may have its load motors transferred to another drive. Finally, a single 250 hp drive is required to power all 163 motors if a large slab is processed. The inherent length of the tunnel furnace dictates long drive to motor output cables with substantial line to ground capacitance as shown in Fig. 3. This creates the potential for large zero sequence cable charging currents during switching instants of the VFD output waveform. Previous generation six step drive inverters used Bipolar Junction Transistor (BJT) semiconductors with slow 1-2 µs switch times which tended to reduce the peak line to ground current. Also, BJTs were gate current driven devices which beta limited the maximum output current allowed. The BJT would stay in the active region under beta limit control during switching, until either the cable was fully charged or a transistor desaturation protection circuit was faulted. Some BJT applications upsized the drive hp requirements or would require a simple 3 phase output line reactor to prevent faults induced by cable charge currents. Use of Insulated **Bricmont Inc. 395 Valley Brook Road McMurray, Pa (412) Ext. 236 Gate Bipolar Transistor (IGBT) inverters with 50 ns switch times and higher switching rates (2 khz to 12 khz) requires a careful investigation of the zero sequence cable charging current problem for multi-motor drives using long drive output cables. The IGBT is a gate voltage driven device which can source high magnitudes of transient output current, limited only by external drive impedance. The fast risetimes create a line to ground impulse voltage source during switching, which tends to maximize the peak line to ground current seen. It is shown that each long cable leaving the drive may have a 3 Apk line to line current as well as an 8 Apk current from phase wire capacitance to ground. A plant system ground current problem now exists with the newer drives because all 163 motors are switched synchronously resulting in 8 Apk times 163 cables or 1,300 amps peak transient current to ground possible during every switch instant! This current is associated with approximately 8 miles of three phase drive output cable. This paper describes the use of a special PWM output filter to reduce the effect zero sequence current on drive ground fault trips, system ground fault indicator mis-operation and increased Electro Magnetic Interference (EMI) noise in the plant ground. The installation of the special PWM output filter substantially reduced the peak ground current for synchronous operation of 72 one hp motors. However, the drive ground fault trip returned as power to the last 91 one hp motors was applied. To circumvent this problem, the original plant one line power diagram was changed from a single 2.5 MVA solid ground transformer feeding three 250 hp drives, to a solid ground 2.5 MVA plant transformer feeding separate 250 kva resistive ground isolation transformers for each 250 hp drive as shown in Fig. 1. The remaining sections discuss the cable charge phenomenon, equations describing its magnitude, characterization results of component high frequency surge impedance s used in determining charge current magnitude and system simulation. Measured field site data is compared to system simulation results. Solutions to the cable charge phenomenon and pre-installation application guidelines are provided to insure successful utilization of the multi-motor drive topology that is typically used in the metals industry. II. CABLE CHARGING CURRENT PHENOMENON Cable charge current consists of transient line to line (I ll ) and line to ground (I lg ) components. Fig. 4 defines these current paths in motor & Power Equipment (PE) ground wires for a grounded conduit configuration. Wires are represented by distributed line inductance (Lo1), distributed line to line stray capacitance (C ll ) and distributed line to conduit ground cable capacitance C lg. The C lg component additionally consists of stator winding to motor PE frame ground capacitance (C sg ). A. Line to Line Cable Charge Current Each IGBT switching interval sets a dv/dt transition of the line to line PWM voltage pulse (V ll ) in Fig. 5 and induces a transient line to line cable charge current I ll from the drive dc bus capacitor through

2 the gated IGBT. The traveling I ll wave enters distributed cable L ol, into C ll and returns through another phase directly back to the drive bus capacitor. The V ll zero voltage dwell time prior to the next dv/dt transition to the dc bus voltage (V bus ) level dictates whether cable C ll is fully discharged to zero volts following the initial transient. Initially uncharged cable conditions are typical for drives with low PWM carrier frequencies (f c ) or short output cables. High f c or long cable conditions may have a positive or negative residual cable voltage (V res ) prior to V bus transitions, depending on interaction of the PWM modulator and output cable characteristics [1]. Peak transient I ll cable charge current associated with V ll traveling wave voltage is estimated in (1) using cable line to line surge impedance (Z ll ), V bus and a (+/-V res ), dependent on whether V res aids or opposes the incoming V bus value. Section III further discusses typical magnitudes for Z ll. I ll = V bus ( + / ) V res Zll where B. Line to Ground Cable Charge Current Z ll = Lo1 (1) Cll Each IGBT switching transition in a given phase also sources a line to ground cable charging current path in Fig. 4 from line inductance L o1 through C lg and C sg. The zero sequence voltage waveform (V ng ) at the drive PWM output is the source of I lg current. Fig. 6 shows V ng and I lg waveforms for a single 480V drive (V bus = 650 Vdc) operating at f c = 4 khz and output 30 Hz frequency and connected to a single motor cable of 300 ft. V ng was measured between the neutral point of a wye configured 1 Meg-ohm resistor network connected to phase A, B, C and drive PE ground point. V ng contains a dc bus related 180 Hz ripple voltage and 4 khz modulation voltage component. Fig. 7 is an expanded V ng version showing the 4 khz step like modulation component along with associated I lg transient current at each IGBT transition. I lg is the common mode line to ground current summation of all 3 phases, measured by passing all three motor phase wires thru a Pearson high frequency current transformer. In contrast to the V ll waveform of Fig. 5, there is no dwell time in Fig. 7 V ng waveform. Highest I lg transients (8 Apk) in Fig. 7 occur when capacitance C lg and C sg have previously attained a steady state charge of opposite polarity to the incoming step magnitude change. Assuming the theoretical neutral of the PWM inverter is the V bus midpoint, then the maximum line to ground step magnitude change is (V bus /2) at the drive output. Fig. 7 shows highest I lg transients occur when C lg is charged to a -250V (-V bus /3) plateau and a charge reversal step of V bus /2 = 325 V is applied. Because of the charge reversal phenomenon, the maximum transient line to ground forcing voltage is roughly V bus. The I lg magnitude is limited by line to ground surge impedance (Z lg ) in (2). Section III further discusses typical magnitudes for Z lg. ~ Vbus = where Ilg Z lg = Lo1 Zlg Clg C. Zero Sequence Current Paths Fig. 8 shows system I lg current paths taken for a configuration consisting of 3 phase output wires plus ground wire enclosed in a conduit and an input tray cable from a wye grounded feeder transformer. The conduit is bonded to drive cabinet and motor junction box and the green ground PE wire is connected to ground stud in the motor junction box and drive cabinet PE bus. Transient I lg current sourced from the drive flows thru cable capacitance to the grounded conduit wall, to green PE ground wire and partly thru motor stator winding capacitance to frame ground. The conduit, and (2) to a lesser extent the internal green PE ground wire, absorb most I lg current and return it back to the drive out of the ground grid, thereby reducing "ground noise" for the length of Potential #1 - Potential #2 run. However, a conduit may have accidental contact with grid ground structure due to straps, support, etc. that bypasses I lg back into the ground grid. How I lg current divides between the conduit, green wire or ground grid is dependent on the variable and unpredictable ac resistance characteristics of earth ground at the application site. Upon arriving at drive PE, I lg must enter the ground grid, since a physical connection between drive PE ground and power structure does not exist. I lg must bypass drive PE ground, and remain in the ground grid until the feeder transformer secondary grounded neutral (Xo) is found. At this point a metallic path back to the drive source can occur on input phase R, S or T. Once inside the drive, I lg current path selects the bridge rectifier diode that is conducting back to the dc bus source. The I lg ground current path spans both output and input cable lengths, in contrast to the I ll current path which was solely confined to the output wires. It is also evident that placing the feeder transformer close to the drive results in less ground grid noise. Using conduit with a PE ground wire bonded from feeder transformer Xo to the drive input is also beneficial in reducing ground grid noise. It is seen in Fig. 8 that one method to reduce I lg is to insert a high resistance type grounding resistor in the series I lg path of the transformer Xo lead. This is discussed in Section IV. D. Zero Sequence Current Problems Fig. 9 shows I lg and I ll current spikes adding to the fundamental output phase current of a single low hp motor. In this multi-motor application, I lg and I ll currents travel synchronously down each of the 160 motor cables, so that peak zero sequence current magnitude summed in the ground grid becomes enormous. The drive Ground Fault Circuit prevented continuous operation because of the sum total of the I lg magnitude. An additional problem encountered is that the RMS value of the I lg and I ll currents must be added to the one hp rated load current value to prevent the output overload heater from malfunctioning. The RMS value of the I lg and I ll currents may be decreased by reducing f c. However, peak I lg current remains unchanged and may still cause system noise problems. Common mode noise in the system ground grid created by the zero sequence current may also effect sensitive electronic equipment referenced to ground and is discussed in [2-5]. The solution to these problems was the addition of a special PWM output filter and an input high resistance ground isolation transformer for each drive discussed in Section V. III. CHARACTERIZATION OF SYSTEM COMPONENTS This section characterizes the high frequency surge impedance of components that determine peak current magnitudes for I ll and I lg in (1) and (2). The high frequency impedance of other system components in the I lg path are also characterized for use in Section IV simulation. Simulation is used for analysis as well as design of the component parameters required to solve the cable charge current problem. Component impedance was measured with HP 4284A and HP 4285A R-L-C Impedance Analyzers with frequency ranges of 20 Hz - 1 MHz and 75 khz to 30 MHz, respectively. Component surge impedance at the specific frequencies corresponding to drive transition time (fn) and I lg oscillation frequency (fo) are of importance. Equivalent frequency fn is defined in [3] and Fig. 5 as fn = 1/π trise and is ~ 1.3 MHz for the 250 hp output voltage risetime transition of 250 ns. The Z ll at fn is used to calculate I ll and I lg in (1) and (2). The I lg oscillation frequency fo typically varies between 100 khz and 1 MHz.

3 A. Normal Mode Z ll Impedance of Wires in Conduit Normal mode and common mode impedance of the application s #12 AWG Poly Vinyl Chloride (PVC) wires in a grounded conduit was experimentally measured with impedance analyzers for a 10 ft conduit section. Measured surge impedance Z ll between two conductors was determined by measuring open circuit capacitance C oc between two phase wires and by measuring short circuit inductance L sc between the two wires at the conduit input side with the same two wires shorted at the output side of the conduit [6]. The 60 Hz capacitance C oc between phases was 40 pf and is much lower than that of a tightly bundled cable since the inherent air space between wire insulation reduces the effective dielectric constant closer to that of air (ε r = 1) rather than the PVC insulation (ε r = 5.6) value. It was important to ground the test conduit during normal mode measurements. Otherwise, the C oc value doubled to an erroneous 80 pf. value. The 60 Hz C oc value variation with frequency in Fig. 10 decreases ~ 10 % at 100 khz and ~ 50 % during the drive transition equivalent frequency of 1.3 MHz. Z ll = Lsc Coc High frequency inductance and resistance variation in Fig. 10 follows the classical skin and proximity effect profile for two wires with equal and opposite current flow that are semi-adjacent to each other. The inductance decreases to 60 % to 70 % of the 60 Hz value over the 100 khz to 2 MHz range. The [ac/dc] resistance ratio is [6x] or 0.24 Ω at 100 khz and [40x] or 1.6 Ω at 2 MHz. Fig. 12 shows line to line surge impedance Z ll variation with frequency calculated from (3) and measured L sc - C oc data of Fig. 10. The Z ll = 300 Ω at fn, so peak I ll transient using (1) is ~ 2 Apk for a 650 V dc bus. While the 1.6 Ω ac resistance at fn provides transient damping, it is seen the 300 Ω surge impedance approximation of a lossless line dictates I ll magnitude. Peak line to line charge current magnitude of PVC wires in a conduit is 4x to 5x lower than for tightly bundled armor cable, since Z ll range for these cables is ~ 50 Ω - 80 Ω [6]. The 250 hp drive without any output filter must transiently source an I ll = 2.2 Apk*163 cables = 358 Apk. B. Common Mode Z lg Impedance of Wires in Conduit Measured surge impedance Z lg between phase conductor and conduit ground was determined by measuring open circuit capacitance C oc between phase wire and the conduit and by measuring short circuit inductance L sc between phase A wire input and conduit input with opposite end of phase A bonded to the conduit output side. The test conduit was floating during common mode measurements. The 60 Hz capacitance C oc between phase and conduit was 250 pf and is much higher than the line to line value. The 60 Hz C oc value variation with frequency in Fig. 11 decreases ~ 15 % at 100 khz and also ~ 15 % during the drive transition equivalent frequency of 1.3 MHz. High frequency inductance and resistance variation in Fig. 11 follows the classical skin and proximity effect profile for a common mode I lg current path that goes down a copper phase wire and returns in the opposite direction down a coaxial steel tube. Inductance decreases to 40% of the 60 Hz value over the 100 khz to 2 MHz range. The [ac/dc] resistance ratio is [4x] or 0.3 Ω at 100 khz and [20x] or 1.5 Ω at 2 MHz. Fig. 12 shows line to line surge impedance Z lg variation with frequency calculated using (3) but measured L sc - C oc data of Fig. 11. The Z lg = 80 Ω at fn, so peak I lg transient using (2) is ~ 8 Apk for a 650 V dc bus. At frequencies > 5 MHz, the Z lg of Fig. 12 tends toward a 50 Ω coaxial value. The 1.5 Ω ac resistance at fn for the measured 10 ft section provides transient damping and is later seen to affect I lg magnitude. The 250 hp drive without any output filter or damping (3) considered must theoretically transiently source an I lg = 8 Apk*163 cables = 1,300 Apk. Common mode surge impedance testing with the insulated PE ground wire bonded to both ends of the conduit was also performed. Results show that Z lg of Fig. 12 decreased by only 5% over the 100 khz to 5 MHz range. This implies that the PE wire does not carry the high frequency zero sequence current. Fig. 13 shows test results of an isolated 300 ft section which plots I lg components in the conduit and the PE ground wire during a Fig. 7 step transition in common mode voltage. Fig. 13 verifies that the conduit, although made of steel, looks like a low inductance coaxial tube to high frequency, while the insulated PE wire appears a high value inductor at high frequency. The transient I lg component of 1,300 Apk during switching will be modified to lower value dependent on the surge impedance of the input transformer, output filter and motor components in the I lg ground path, which spans both output cable and input cable lengths. The I ll = 358 Apk component estimate will also be modified to lower value dependent only on the output filter surge impedance addition. C. Input Transformer Impedance The equivalent circuit R-L-C parameters at high frequency are estimated in Table 1 for both 2.5 MVA and 250 kva transformers used. The R (60Hz) - L (60Hz) parameters of Table 1 are per phase values referred to the secondary side and determined using Fig. 14 transformer (X/R) ratio and (%Z) data 60 Hz and equations (4) through (6). The L (60Hz) parameter corresponds to air core coil leakage reactance of both primary and secondary coils. V ll( ac) = V base 3 ; Ibase phaseva rating = 3 V base ; = (4) 3 V Zbase ll( ac) Ibase 2 Z phase Z Zbase X l R 2 2 = % * = + = ( K) R (5) X R = K from Fig.14 ; L phase X l = (6) 2π f Determining accurate estimates for high frequency R-L parameters (R (1MHz) - L (1MHz) ) at f n would itself be study subject outside the scope of this paper. Also, time did not permit on-site testing of the two transformers used. High frequency R-L approximations at f n were determined by adjusting R (60Hz) - L (60Hz) with the two wire results of Fig. 10 with (Rac/Rdc = 40x) pu factor and (Lac/Ldc = 0.7x) pu factor. It is assumed ½ of the leakage is in the primary and ½ in the secondary for both transformers. Fig. 8 shows that for the 2.5 MVA transformer, only the leakage inductance and resistance of the secondary side conduct zero sequence current I lg, and that I lg, may flow on all three phases in parallel ( 1/3 pu phase value) back to the drive. Thus, the 2.5 MVA (R (1MHz) - L (1MHz) ) parameters are multiplied by (1/6) to get (R 1(Equiv) - L 1(Equiv) ) parameters used in the system P-SPICE model of Fig. 18 for I lg magnitude determination. Table 1 Equivalent R-L-C Parameters for 250 kva & 2.5 MVA transformers kva R (60Hz) L (60Hz) R (1MHz) L (1MHz) R (Eqiuv) L (Equiv) C (Equiv) mω 86 µη 307 mω 60 µη 50 mω 10 µη 20 nf R2 & R4 L2 & L3 2, µω 14 µη 30 mω 10 µη 5 mω 2 µη R 1 L 1 In a similar approach, L 2(Equiv) and L 3(Equiv), each represents the zero sequence high frequency primary and secondary non-saturable air

4 core leakage inductance of the 250 kva 1:1 isolation transformer and R 2(Equiv), R 4(Equiv), each represents the zero sequence high frequency primary and secondary coil resistance. The C (Equiv) parameter was measured between a primary phase lead and Xo of the secondary side on the 250 kva transformer at the field site. The C (Equiv) parameter simulates the inherent high frequency coupling capacitor between primary and secondary coils that can partially transfer part of the I lg current. D. Output Filter Impedance The primary filter component of Fig. 15 consists of a 250 hp 3 phase reactor rated at 5% impedance. The reactor is beneficial in reducing both I ll and I lg charge current components. The apparent 60 Hz inductance at rated current is determined using (7) and by measuring phase resistance R phase and applying V rms until rated I rms occurs. V rms 2 2 = Z phase = X phase + R phase ; I L rms phase X phase = (7) 2π f The 5% Z rating is the highest inductance allowed that does not significantly decrease the 60 Hz output voltage to the motor. However, the apparent 60 Hz inductance of these reactors drops significantly at the important f n frequency because of the magnetic skin effect of the core laminations. Fig. 16 shows impedance analyzer results indicating only 18% of the 60 Hz inductance (~ 20 µη) exists in the 1 MHz to 2 MHz region where it is desirable to maintain inductance to limit I lg during drive switching. Reactor core loss resistor R core was measured at f n to be 250 Ω. The ac skin effect resistance of each phase coil at f n was measured to be 60 mω. An external resistor R damp was added to each phase to provide additional damping in the output circuit. Only phase values are used in the output circuit simulation since I lg is sourced from one switching instant in an inverter phase. A common mode core made of high frequency ferrite material in Fig. 15 was also added in series to help compensate for the drop in iron core inductance during switching. E. Motor Common Mode Surge Impedance The transient V ll waveform entering the motor is propagated and attenuated through the stator line to line winding, which is modeled as a distributed transmission line [4]. There have been simple and more complex methods attempts to model the high frequency line to line surge impedance characteristics of the motor winding [7,8]. However, there is little literature on the common mode surge impedance of the motor. Of interest in calculating I lg magnitude is the motor common mode surge impedance when the zero sequence traveling wave enters the motor winding to frame ground. The Z lg of the motor winding was measured using the designated R-L-C analyzers and appears as a phase to ground capacitor with impedance X c = (1 / 2πf C mg ) at 60 Hz. Fig. 17 shows the measured 60 Hz C mg values for various motor hp sizes. The C mg of a 1 hp motor is between 1 nf to 3 nf. The Z lg at the f n frequency is also highly capacitive, but the C mg value is reduced and the skin effect resistance of the winding becomes evident. The P-SPICE model used at f n,, C 4 in Fig. 18, is thus a simple capacitor to ground with a series resistor accounting for ac skin effect resistance of the winding. F. I lg Estimate using System Surge Impedance The total inductance in the series I lg path at f n is ~ 20 µη for the filter inductor, 2 µη for the 2.5 MVA transformer, 2 µη stray inductance in the drive. The output cable inductance is estimated as a lump sum of 95 µη per motor line for 500 ft length (Fig. 11; 4.75 µη * 40% effect skin factor per 10 ft length). However, all lines are in parallel for 161 motors so the net lumped cable inductance is 0.58 µη. The values indicate the high frequency inductance of the filter dominates over any other component. The total lumped capacitance in the series I lg path at f n is ~ C mg * 161 motors = 484 nf plus cable capacitance. The output cable capacitance is estimated as a lump sum of 10.6 nf per motor line for 500 ft length (Fig pf * 85% effect skin factor per 10 ft length). However, all lines are in parallel for 161 motors so the net lumped cable capacitance is 1.71 µf. The values indicate cable capacitance dominates over motor to ground capacitance. An I lg estimate using a lumped parameter surge impedance model of the components in the I lg path is calculated using a lumped L ol of µη and 2.19 µf for lumped capacitance C lg in (1). This results in Z lg of 3.35 ohms and peak I lg of 190 Apk without taking into the effect of damping resistance. IV. SIMULATION OF FIELD CONDITIONS A. Zero Sequence Equivalent Circuit Fig. 18 shows a P-SPICE equivalent circuit model of all components in the I lg zero sequence path defined at f n. The 2.5 MVA solid Xo ground transformer parameters R 1 & L 1 were defined previously. The 250 kva transformer parameters R 2, R 4 and L 2, L 3 were defined previously and resistor R 3 = 250 Ω that is connected to Xo. The 250 kva transformer and R 3 combination was a final solution that was not initially at the application site. The input conduit consisted of 75 ft of 320 amp conductors in a large conduit with lossless transmission line parameters L 4 and C 2. Resistor R 5 simulates high frequency skin effect resistance losses in the cable. The drive is represented by a step voltage change through an IGBT with ON state resistance and some stray inductance in the drive bus work. The output filter lumped parameters R 6 & L 5 were defined previously. A P-SPICE lossless transmission line model for the single output cable was previously defined using C 3 & L 6 values per meter. Resistor R 7 is a lumped value for the cable skin effect loss. A more sophisticated transmission line model that models distributed cable losses is detailed in [9] using the SIMULINK program. Capacitor C 4 is the line to ground C mg of 3 nf for one motor. The remaining 162 motors are grouped into one lossless transmission line, loss resistor and motor model. C 6 is the line to ground C mg of 162 motors or 480 nf. The skin effect resistor R 8 is a lumped value calculated as (R 7 /162) and line inductance is (L 6 /162). The value of the model lies in its ability to predict current for any system combination of components without expensive field testing. Fig. 19 through Fig. 22 simulation results are for various cases of synchronously switching 163 motors at once and measuring I lg. B. Charging Current Simulation Fig. 19a simulation results are shown with the 2.5 MVA and 250 kva transformer, input conduits, output filter and cable skin effect resistor R 7 and R 8 all removed from the circuit. An I lg value of 1,310 Apk with 500 khz cable oscillation ring frequency that never decays is shown and agrees with the Section III-B 1,300 Apk prediction. Fig. 19b shows just the addition of cable ac skin effect resistors R 7 and R 8 alone was enough to decrease peak I lg magnitude to 489 Apk at the same 500 khz oscillation frequency of the cable parameters, but now with a decay to zero in 6 µs. Fig. 19c shows the addition of the 2.5 MVA transformer, input conduit and non-ideal IGBT drive model, to the existing ac cable skin effect resistors R 7 and R 8. These additions were enough to decrease peak I lg magnitude to 319 Apk, at much different and lower 42 khz oscillation frequency, but now with a much longer decay to zero in over 100 µs.

5 C. Simulation with Output Filter Added Fig. 20 shows the addition of the 2.5 MVA transformer, input conduit, non ideal IGBT drive model and PWM output filter models along with existing ac cable skin effect resistor R 7 and R 8. This condition simulates the system originally shipped to the customer and corresponds to the brute force peak I lg 190 Apk estimate of Section III-F. The peak I lg magnitude simulated is decreased to 110 Apk at a lower 14 khz oscillation frequency, but now with a much longer decay to zero in over 250 µs as compared to Fig. 19c. The graph of Fig. 20 also shows a reduced I lg component that goes through the motor stator winding to ground. The output line common mode surge impedance is predominant in determining peak I lg out of the drive. The addition of filter inductance may reduce I lg, but it also lengthens the oscillation period, which may lead to higher peak I lg, if it has not decayed to zero before the next drive switching. The drive ground fault sensor trips at exactly 100 amp and also caused random trips due to the 110 Apk. D. Simulation with Isolation Transformer & Output Filter Fig. 21 shows the simulation of the final solution involving all of Fig. 18. The addition of a standard high resistance grounding resistor R 3 = 250 Ω connected to Xo of the 250 kva one to one isolation transformer added provided the substantial damping required in the zero sequence ground path. The leakage reactance provided extra non-saturable inductance s to limit current. The primary to secondary series capacitance actually changed the circuit oscillation frequency. Fig. 20 shows the 250 kva transformer primary zero sequence current still conducts a large percentage of the I lg component at the drive output. The equivalent circuit of Fig. 18 shows that R 3 is actually damping resistor in parallel with the 2.5 MVA secondary side and the 250 kva primary side. The peak I lg magnitude simulated is decreased to a tolerable 24 Apk at a higher 120 khz oscillation frequency, and now with a drastically reduced decay to zero in 20 µs to 40 µs as compared to Fig. 20. E. Simulation with Grounded Transformer & Output Filter The effect of varying R 3 = 0 Ω to represent a grounded wye 250 kva isolation transformer with all the components of Fig. 18 intact was investigated by simulation. Fig. 22 shows the peak I lg magnitude in the 2.5 MVA secondary side and the 250 kva primary side goes to zero. The I lg oscillation frequency reverts back to 18 khz, since C 1 is essentially shorted. The peak magnitude increases to an unacceptable 141 Apk with a 250 µs decay time due to no damping resistance in the zero sequence path. Thus, a solid grounded isolation transformer is unacceptable for drive ground fault indicator operation. V. FIELD SITE MEASUREMENTS Fig. 23a shows I lg common mode current at the drive output that was measured at the field site for 163 motors starting up to 30 Hz output frequency. Fig. 23a corresponds to conditions of a 2.5 MVA source transformer with a PWM output filter that was simulated in Section IV-C and Fig. 20. The 120 Apk, 14 khz oscillation frequency and 250 µs decay time agree well with the simulated 110 Apk, 15 khz oscillation frequency and 250 µs decay time. Fig. 23b also shows measured I lg common mode current at the drive output with six slabs of steel on the rollers. The I lg was measured at the field site for 163 motors starting and running to an output frequency of 12 Hz. Fig. 23b corresponds to conditions of a 2.5 MVA source transformer, 250 kva isolation transformer with 250 Ω wye ground resistor and the PWM output filter that was simulated in Section IV-D and Fig. 21. The 32 Apk, 120 khz oscillation frequency and 20 µs decay time agree well with the simulated 24 Apk, 120 khz oscillation frequency and 20 µs to 30 µs decay time. Fig. 24 is a long term time scale of Fig. 23a showing both I lg common mode current and phase current at the drive output that was measured at the field site for 163 motors starting up to 30 Hz output frequency. Note the high RMS content of the zero sequence current. The phase current does not represent the I lg spikes due to aliasing at 5 ms per division. Fig. 25 is a long term time scale of Fig. 23b showing the PE insulated green ground wire I lg common mode current component vs. the V ll switching waveform that was measured at the field site for 163 motors at 30 Hz output frequency. Note the lower RMS content of the PE zero sequence current as compared to the total common mode current of 32 Apk. This agrees with previous lab measurements showing that most of I lg current conducts within the conduit. VI. CONCLUSION This paper described how line to ground cable charging current is generated, the paths it takes and the system impact when there are multiple parallel output cables attached to single drive output. High frequency measurement techniques were outlined on how to determine the impedance s of the components in the path of the cable and motor line to ground charging current. Knowing these impedance s, simple equations were shown to give rough estimates of the cable charge current magnitude. A P-SPICE model of the zero sequence circuit was shown to be accurate tool and had good agreement with measured data taken at the field site. In general, the use of an output filter alone, consisting of mostly inductance, tends to reduce the cable charge magnitude but spreads the resulting oscillation over a longer time. Selection of a filter damping resistor, in parallel with the output filter, would not significantly affect the oscillation decay time. The use of a high resistance grounded input isolation transformer in addition to the output filter, added extra line inductance and tended to reduce the cable charge magnitude. The transformer high resistance ground resistor in series with the ground path through Xo provided a place for a filter damping resistor, that is not in the main normal mode current path, and which could significantly decrease the oscillation decay time and peak charge current to ground magnitude. The use of a solidly grounded input isolation transformer, with or without an output filter, does not significantly decrease the oscillation decay time nor the peak charge current to ground magnitude. ACKNOWLEDGMENT The authors wish to thank D. Schlegel, J. Pankau and H. Jelinik for impedance measurement and paper support activities. REFERENCES [1] Kerkman,R.,Leggate,D.,Skibinski,G.,"Interaction of drive modulation & cable parameters on AC motor transients",ieee-ias-1997 IAS Conf [2] G. Skibinski, J. Pankau, & W. Maslowski,"Installation considerations for IGBT ac drives",1997 IEEE Annual Textile,Fiber & Film Ind. Conf [3] G. Skibinski, J. Pankau, R. Sladky, J. Campbell, Generation, control and regulation of EMI from ac drives, IEEE-IAS-1997 Annual Conf. [4] Anderson,Kerkman,Saunders,Schlegel, and Skibinski, "Modern Drives Application Issues & Solutions Tutorial", 1996 IEEE-IAS-(PCIC),PA [5] Rendusara,D.,Enjeti,P., Inverter output filter reduces common mode & differential mode dv/dt at motor terminals in PWM drives, 1998 PESC [6] Skibinski, G., "Design methodology of a cable terminator to reduce reflected voltage on AC motors", IEEE Ind. Appl. Soc. Conf., 1996 [7] Skibinski,Kerkman,Leggate,Pankau,Schlegel, Reflected wave modeling techniques for PWM ac motor drives, 1998 IEEE-APEC, Anaheim,CA [8] Grandi,G.,Casade,D., Reggiani,U., Analysis of common mode & differential mode hf current in PWM fed ac motors,1998 PESC, Japan [9] Leggate, Pankau, Schlegel, Kerkman, Skibinski, Reflected waves and their associated current, 1998 IEEE-IAS annual Conf.

6 Fig. 2. Picture showing tunnel furnace section with motors Fig. 3. Picture showing potential for high zero sequence currents flowing in the 163 multiple conduits and wireway V ng Fig. 1. One line diagram of steel process furnace (400 ft long) showing a 2.5 MVA feeder transformer, three 250 Hp IGBT PWM drives, output filters & isolation transformers, output contactors, wireways, and conduits to 163 one hp motors. A I MOTOR B L 01 C ll C lg C lg Motor Drive C L 02 C mg PE L 03 Conduit Fig. 6. Measured V ng & I lg of a 480V 30 Hz, 300' of motor cable [ "0" at offest 4 div; 2 ms/div; 2 A/div; 100V/div] Fig. 4. Line to line I ll and line to ground I lg cable charging current paths t rise f c t fall f n =.3 18 / t ri se DC Bus I lg V ll τ t I ll t V ng Fig. 5. Line to line V ll PWM voltage and I ll charging current Fig. 7. Expanded V ng & I lg of a 480V 30 Hz, 300' of motor cable [ "0" at offest 4 div; 50 µs/div; 2 A/div; 100V/div]

7 R AC Drive + U Motor Frame 450 PE X O S T Common Mode Current PE A+ A- - Logic V W PE Some HF Accidental Contact Of Conduit Motor C Slot Motor PE Motor Capacitance PE Ground If Required By Code Line to Line TE Potential 4 Potential #1 Potential #2 Potential #3 Fig. 8. System path of l Ig zero sequence current 150 Line to Conduit Ground Frequency [ Hz ] Fig. 12. Measured Z lg & Z ll surge impedance's of three #12 awg PVC wires and a PVC insulated PE wire in a grounded conduit Fig. 9. Phase current to one hp cable showing I ll & I lg transient current spikes pu R =40 mohm 1 pu L = 3.75 uh 1 pu C = 40 pf Normal Mode Resistance Line to Line 8 Apk I lg of Conduit Normal Mode Inductance Line to Line Normal Mode Capacitance Line to Line Frequency [ Hz ] Fig. 10. Measured normal mode R-L-C characteristics of three #12 awg PVC wires and a PVC insulated PE wire in a 10 grounded conduit pu R = 80 mohm 1 pu L = 4.75 uh 1 pu C = 250 pf Common Mode Resistance Wire to Conduit 1 Apk I lg of PE Wire V ng step Fig. 13. I lg of Conduit & I lg of PE ground wire during V ng step change [ "0" at offset 3 div; 5 µs/div ; 2 A/div; 2 A/div; 100V/div] Transformer ( X/R ) Ratio Transformer % Z Rating 1 1 Common Mode Capacitance Wire to Conduit Transformer kva Rating 0.1 Common Mode Inductance Wire to Conduit Frequency [Hz] Fig. 11. Measured common mode R-L-C characteristics of three #12 awg PVC wires and a PVC insulated PE wire in a 10 grounded conduit Fig. 14. Estimated transformer (% Z) and (X/R) Ratio vs. kva rating

8 Rext Lcommon mode Rcore Lcoil Rcoil Rext Rcore Lcoil Rcoil Lcoil Rext Rcore Rcoil Fig. 19a. I lg simulation results for a V ng step input into 163 cables at time zero with the 2.5 MVA and 250 kva transformer, input conduits, output filter and cable skin effect resistors R 7 R 8 all removed from Fig. 18 circuit path. [500 A/div ; 1µs/div] Fig. 15. Special PWM output filter with iron core inductor, external damping resistor and common mode inductor of high frequency ferrite material pu phase inductance = 80 uh Frequency [ Hz ] Fig. 16. Inductance vs. frequency of the iron core reactor used in the PWM filter. The inductance is only 18 % of the 60 Hz value during switching. Fig. 19b. I lg simulation results for a V ng step input into 163 cables at time zero with the 2.5 MVA and 250 kva transformer, input conduits and output filter all removed from Fig. 18 circuit path but with incorporating cable skin effect resistors R 7 R 8. [100 A/div; 1 µs/div] 1.0E E E E E AC Motor Horsepower Rating [ hp ] Fig. 17. Measured stator winding to ground capacitance vs. hp rating Fig. 19c. I lg simulation results for a V ng step input into 163 cables at time zero with only the 250 kva transformer and output filter removed from Fig. 18 circuit path. [100 A/div; 20 µs/div] 2.5 MVA XFMR 250 kva XFMR Xo I2 R1 L1 I1 Xo R2 L2 C1 L3 R4 I3 R3 I4 INPUT CONDUIT L4 R5 C2 I5 V1 IGBT OUTPUT FILTER R6 I7 L5 I8 OUTPUT CONDUIT ONE MOTOR I9 L6 R7 C3 C4 COMBINED MODEL OF QTY 162 OUTPUT CONDUITS & MOTORS I10 L7 R8 C5 C6 Fig. 18. P-SPICE equivalent circuit model of the zero sequence path with all components defined at f n. Fig. 20. I lg simulation results for a V ng step input into 163 cables at time zero (middle cursor) with only the 250 kva transformer removed from Fig. 18 circuit path. Also shown is total current into the combined 162 motor s line to ground C mg capacitance or C 6 [50 A/div; 50µs/div]

9 Fig. 21. Final solution I lg simulation results (top trace) for a V ng step input into 163 cables at time zero (middle cursor) with all components of Fig. 18 circuit path, including the resistive wye ground isolation transformer & output filter. Also shown is the 250 kva X 0 resistor current (middle trace) and 2.5 MVA secondary current (bottom trace) [10 A/div; 20 µs/div] Fig. 23b. Final solution I lg on site measured results for a V ng step input (top trace) into 163 cables at time zero (near middle cursor) with all components of Fig. 18 circuit path, including the resistive wye ground isolation transformer & output filter. [10 A/div; 20 µs/div] Fig. 22. I lg simulation results for a V ng step input into 163 cables at time zero with all final solution components of Fig. 18 circuit path, except the 250 kva transformer s wye X 0 neutral is solidly grounded. Also shown is 250 kva transformer primary current ( o level trace) [50 A/div; 20 µs/div] Fig. 24. Long term view of Fig. 23a conditions showing I lg on site measured results (top trace) for 163 cables with all components of Fig. 18 circuit path, except the 250 kva isolation transformer with a 250 ohm resistive wye ground. [50 A/div; 5 ms/div]. Bottom trace is phase current at drive output frequency of 30 Hz [200 A/div; 5 ms/div] Fig. 23a. Final solution I lg on site measured results for a V ng step input (top trace) into 163 cables at time zero (near middle cursor) with all components of Fig. 18 circuit path, except the 250 kva isolation transformer with a 250 ohm resistive wye ground [50 A/div; 50 µs/div] Fig. 25. Long term view of final solution Fig. 23b conditions showing V ll waveform switching into 163 cables at 30 Hz vs. the low I lg on site measured current in the PE insulated ground wire (bottom trace) [10 A/div; 500V/div 10 ms/div].