14 MHz Single Side Band Transmitter

1 14 MHz Single Side Band Transmitter 1. Objectives. The objective of this work is to calculate and adjust the key elements of an Upper Side Band Transmitter in the 20m range. 2. Devices to study.. - Architecture of a double heterodyne transmitter. - Microphone audio frequency amplifier. - Quartz local oscillator. - Analog multiplier mixer. - Quartz selective filter. - Various types of LC filters. - Wide band HF power amplifier. 3. Bibliography. [5] Circuits et techniques HF et VHF, Prof. C. Dehollain. [2] Traité d électricité volume VIII, Électronique, Prof. R. Dessoulavy et J.-D. Chatelain, PPR. [3] Schematics of the LEG board "Emetteur - Récepteur USB" in appendix. [4] Data sheets of the components on the intranet.

2 4. Bloc diagram. 4.1. Description. Micro Amp. BF M1 FQ M2 FLC1 Amp. HF FLC2 Antenne LO VFO This is the classic architecture of a double heterodyne transmitter. The base-band audio signal from the microphone is amplified, and modulates a fixed Intermediate Frequency given by a Local Oscillator, then it passes through a sharp selective filter which keeps only one of the side band (in this case the upper side band). The useful signal at IF is then brought at the desired transmission frequency by mixing it with a sinus at variable frequency given by a Variable Frequency Oscillator. Finally this HF signal is amplified to give power to the antenna. 4.2. Theoretical forecasts. 4.2.1. Explain how the SSB signal is generated in this schematics. 4.2.2. Knowing that the audio band to transmit is going from 300 Hz to 3000 Hz, explain why the filter FQ at IF = 9 MHz must be so selective. 4.2.3. Calculate the exact frequency of the local oscillator if the quartz filter is centred at 9.000 MHz with a bandwidth of 2800 Hz 4.2.4. Calculate the frequency range covered by the VFO if the transmitted signal is in the band from 14 MHz to 14,5 MHz. 4.2.5. What is the function of the first LC filter (FLC1)? 4.2.6. What is the function of the second LC filter (FLC2)?

3 5. The output filter (FLC2). 5.1. Schematics. C63 56pF In C61 390pF L61 470nH C62 390pF Out This filter is designed to work with 50 Ω source and load resistances. 5.2. Theoretical forecasts. 5.2.1. What is the goal of this filter? 5.2.2. What is the type and order of this filter? 5.2.3. What is the limiting factor of the power transmitted by this filter? 5.3. Measurements. 5.3.1. With a network analyser, measure the bode plot of this filter from 1 MHz to 100 MHz. Measure the attenuation for the harmonics 2 and 3 of the transmitted signal. 5.3.2. Measure precisely the response and the insertion losses in the band from14 MHz to 14.5 MHz. 5.3.3. Calculate the level needed at the amplifier's output to have 0.4 W on an ideal antenna of 50 Ω impedance.

4 6. The 0.5 W HF Power Amplifier (Amp. HF). 6.1. Schematics. +15 R53 1kΩ R54 470Ω C55 22nF C56 L55 22µH C510 P51 20kΩ L52 22µH C54 C59 T51 2N5179 P52 2kΩ L53 33µH C52 C57 P53 2kΩ L51 33µH L54 33µH C58 T53 2N3553 T52 2N3553 Out In R51 C512 R52 C511 This amplifier is designed to have a large power gain without becoming unstable. Stability problems arose because of parasitic high frequency capacitive or inductive coupling between elements, or at low frequency by coupling between stages through the power supply. Self oscillation requires that both phase and gain conditions are satisfied. This amplifier is thus wide band to avoid phase condition. This amplifier has two stages: a class A common emitter around T51, and a class A cascode made of the common emitter T52 and the common base T53. 6.2. Theoretical forecasts. 6.2.1. What are the advantage of the cascode architecture in an wide band amplifier. 6.2.2. Calculate the RMS and peak values of the output current and voltage to obtain 0.5 W in a 50 Ω load. 6.2.3. Calculate the minimum quiescent collector current in T52 and T53. If, for a good thermal stability we choose to drop 1 V across R52, calculate R52 and the base potential of T52 and T53. Calculate the power dissipation in each transistor. 6.2.4. Calculate the input resistance, the voltage gain and the power gain of the cascode stage.

5 6.2.5. Calculate R51 to have 2 V across it with a quiescent collector current of 10 ma in T51. 6.2.6. Calculate the input resistance, the voltage gain and the power gain of the common emitter stage built around T51. 6.2.7. Calculate the power gain of the complete power amplifier. 6.2.8. Calculate the power level needed at the input of this power amplifier to have 0.4 W at the antenna. 6.3. Measurements and tuning. 6.3.1. To avoid the destruction of the amplifier, follow these rules : - Connect a source with 50 Ω impedance and a load of 50 Ω impedance before applying the power supply voltage. - Start with potentiometers P51, P52 and P53 turned fully anti-clockwise (base voltages at 0 V) before applying the power supply voltage. - During the measurement be careful not overloading the spectrum analyser's input. - During the measurement, continuously check that the amplifier is not self-oscillating. 6.3.2 With the input closed on a 50 Ω coax resistor (no input signal): -measure the DC voltage across R51 and adjust P51 to have a quiescent current of 10 ma through T51. -turn P53 fully clockwise, measure the DC voltage across R52 and adjust P52 to have a quiescent current of 200 ma through the cascode stage. -adjust P53 to have on the base of T53 the voltage calculated in 6.2.3, check the quiescent current in the cascode and readjust P52 if necessary. 6.3.3. Apply a sine signal of 14.25 MHz at the input and observe the spectrum of the output signal. Search the best compromise between input level, quiescent current in T51, quiescent current in T52-T53 and T53 base polarisation to obtain the desired output power with minimum harmonic distortion. 6.3.4. Measure the gain and the 1 db compression point. 6.3.5. With a dual tone signal at the input, measure the intermodulation distortion at various levels and calculate the 3 rd order intercept point. 6.3.6. Shut down the supply of the power amplifier before inserting the filter FLC2 between the amplifier's output and the 50 Ω load (either a coax resistor, or the spectrum analyser's input or 50 Ω antenna. If necessary, readjust the various parameters to obtain 0.4 W in the load with minimum intermodulation distortion. 6.3.7. Measure the input level needed to have 0.5 W at the antenna.

6 7. The first HF filter (FLC1). 7.1. Schematics. In C44 R41 2.7kΩ C41 L41 2.2µH C43 L42 2.2µH C42 +15 U41 LM6361-15 R42 2.7kΩ C45 Out This filter is a dual LC resonators with capacitive coupling. The voltage follower allows to give enough current to the low impedance input of the power amplifier without damping the filter too much. 7.2. Theoretical forecasts. 5.2.1. What is the goal of this filter? 7.2.2. Assuming an output impedance of the preceding stage much lower than R41, calculate C41, C42 and C43 to obtain a centre frequency of 14.25 MHz with a 2 MHz bandwidth at -3 db. 7.3. Measurements and tuning. 7.3.1. The measurements will be made with the following stage connected and using a high impedance HF probe to respect the effective load in the circuit. 7.3.2. Measure the transfer function of this filter with a network analyser and adjust L41, L42 and C43 to obtain an optimum flat response centred at 14.25 MHz. 7.3.3. Measure the insertion loss. 7.3.4. Measure the level needed at the input of this stage to have 0.4 W at the antenna.

7 8. The second mixer (M2). 8.1. Schematics. VFO In 5 MHz IF In 9 MHz R31 50Ω R34 1kΩ C31 C33 1 6 X Y U31 AD734 X*Y 1 12 C32 RF Out 14 MHz This mixer is made with the integrated analog multiplier AD734. This circuit is easy to implement and has a great linearity. The impedance at the inputs of the chip being very high, they are reduced by parallel resistors. R31 is matched to the source resistance of a lab waveform generator and R34 is the ideal load resistance for the quartz IF filter which precedes this mixer. 8.2. Theoretical forecasts. 8.2.1. Assuming 0.5 Veff at the VFO input and knowing the signal level needed at the input of the following stage, calculate the input level required at the IF input of this mixer. 8.3. Measurements. 8.3.1. The measurements will be made with the following stage connected and using a high impedance HF probe to respect the effective load in the circuit. 8.3.2. Apply the forecast signals at the inputs of this mixer and measure the spectrum of the output signal. Measure also the spectrum of the signal at the output of FLC1. 8.3.3. Measure the conversion gain of this mixer. 8.3.4. Measure the level needed at the IF input of this stage to have 0.4 W at the antenna.

8 9. The quartz filter (FQ). 9.1. Schematics. In C101 5-40pF L' 6.8µH 1 3 F101 A09F24A 4 2 L' 6.8µH C102 5-40pF Out The goal of this filter is to suppress as much as possible the lower side band while transmitting the upper side band with minimum attenuation. These two side bands are separated by 600 Hz around 9 MHz. Such a selective filtering is only achievable with a quartz network. To obtain a minimum ripple in pass-band, this filter needs to be driven by a source with precise impedance, in this case R21 in mixer M1, and loaded with a precise impedance, in this case R34 in mixer M2. The reactive part of these impedances are made minimum by adjusting C101 and C102 so that the total capacitance (parasitic + adjustable) is cancelled by the inductance L'. 9.2. Theoretical forecasts. 9.2.1. Other than the suppression of the lower side band, what is the second function of this filter? 9.3. Measurements and tuning. 9.3.1. Replace the IC AD734 in mixer M1 by the small set in the schematics : point de calibration BF In C21 100nF 1 6 7 50Ω 12 C22 100nF R21 1kΩ IF Out 9 MHz Connect the filter between mixers M1 and M2. Connect the source of the network analyser to BF In of M1. Connect the input of the network analyser to IF In of M2 using an active probe with high input impedance. These cautions are taken to avoid to disturb the impedances at the filter's accesses. 9.3.2. In these conditions, measure the frequency response of the filter and adjust C101 and C102 to minimise ripple in the pass-band and insertion losses. 9.3.3. Measure precisely bandwidth at 3 db and insertion losses. 9.3.4. Measure the level needed at the filter's input to have 0.4 W at the antenna.

9 10. The local oscillator (LO = Osc3). 10.1. Schematics. +7V R162 47kΩ L161 C161 5-105pF C166 22nF XT161 9MHz L162 1µH C163 2-15pF R161 560 T161 BFR91 C162 C164 9 MHz LO Out This oscillator is built around a bipolar transistor in common emitter. The parallel LC resonator in the collector allows to select the fundamental frequency. The inductor L162 allows to widen the frequency range of the quartz. 10.2. Theoretical forecasts. 10.2.1. What is the maximum peak to peak voltage at the output? 10.2.2. How is made the positive feedback which allows self-oscillation. 10.2.3. Calculate L161 for an oscillation at 9 MHz (quartz's fundamental). 10.3. Measurements and tuning. 10.3.1. The adjustment of such an oscillator is delicate. Make this tuning with LO Out connected to LO In of M1, because the parasitic capacitance, mainly due to the coax cable, influences the oscillator. With C163 in the middle position, adjust C161 to have an oscillation of maximum amplitude and a frequency near the desired value. Then push the frequency precisely to 8.9985 MHz by adjusting C163. If not successful, readjust C161 then C163. 10.3.2. Measure the spectrum of the output signal from 1 MHz to 150 MHz. Measure the level of the useful component and the relative levels of its main harmonics.

10 11. The LF input amplifier. 11.1. Schematics. +15 R12 5.6kΩ P11 500kΩ C11 10µF R11 10kΩ D11 6.2V micro C12 10µF R13 +15-15 U11 LF356 BF Out The électret microphone has an built in preamplifier made of a JFET with R11 as the drain DC load. Circuit made of R12-D11-C11 create a regulated 6.2 V power supply for this preamplifier. The LF signal given by this microphone is amplified by the inverting amplifier built around U11, which gain is given by the ratio P11/R13. 11.2. Theoretical forecast. 11.2.1. What is the AC load of the microphone. The optimum load being 1 kω, calculate the resistor R13 needed. 11.3. Measurements and tuning. 11.3.1. The nominal bandwidth of the input is from 300 Hz to 3 khz, the nominal input level is 10 mv RMS. Apply a dual tone signal at the auxiliary input (BF In Ext.), adjust P11 to have 20 V peak to peak at BF Out, and measure the intermodulation distortion.

11 12. The first mixer (M1). 12.1. Schematics. BF In LO In 9 MHz C21 100nF C23 100nF 1 6 X Y U21 AD734 X*Y 10 12 C22 100nF R21 1kΩ IF Out 9 MHz This mixer is also made with the integrated analog multiplier AD734. The very low output resistance of this circuit is increased to 1 kω by the serie resistor R21, thus presenting a source resistance of optimum value for the quartz filter following this mixer. 12.2. Theoretical forecasts. 12.2.1.Knowing The amplitude of the LO signal at 9 MHz and the signal level needed at the input of the following stage, calculate the input level required at the BF input of this mixer. 12.3. Measurements and tuning. 12.3.1. Apply a signal at the auxiliary input (BF In Ext.), adjust its level and the amplifier's gain to have the level theoretically needed at the BF input of M1 12.3.2. Measure the spectrum of the signal at the output of the multiplier (pin 12) from 1 MHz to 100 MHz. 12.3.3. Measure in details the spectrum of this signal around 9 MHz. 12.3.4 In these same conditions, measure the details of the spectrum of the signal at the quartz filter's output around 9 MHz. Measure the rejection ratio of the FI carrier and the lower side band.

12 13. Final measurements and tuning. 13.1. Apply a dual tone audio signal at the auxiliary input (BF In Ext.), adjust its level and the amplifier's gain to have 0.4 W at the antenna. If its impossible to obtain this power with less than 20 Vp-p at the output of the LF amplifier, increase the level at the VFO input of M2. 13.2 Measure at the output of each stage the level of the signal and the intermodulation distortion. 13.3. Calculate, for the nominal output power, the ratio of harmonic and intermodulation distortion at the antenna. 13.4. Measure the audio (LF) bandwidth of the transmitter. 13.5. Measure the carrier rejection ratio. 13.6. Measure the suppressed side band rejection ratio.