Fig. 1 :Block diagram symbol of the operational amplifier. Characteristics ideal opamp real opamp


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1 Experiment: General Description An operational amplifier (opamp) is defined to be a high gain differential amplifier. When using the opamp with other mainly passive elements, opamp circuits with various characteristics result. Lie the transistor, the opamp belongs to the standard elements of circuit design []. In electronic circuit diagrams the opamp is drawn as bloc diagram symbol in accordance to Fig.. Terminals for operating voltage, offset voltage compensation, frequency response compensation etc. frequently are not drawn. Fig. :Bloc diagram symbol of the operational amplifier When designing circuits with opamps, they normally are based on the "ideal operational amplifier", whose characteristics are represented in Table. For comparison the data areas of real opamps were listed as well. Characteristics ideal opamp real opamp DC voltage difference amplification o 5 o 5 common mode rejection CM 9 db CM 4dB Input impedances Z P and Z 6 Ω Z P,Z 5 Ω Input currents I P and I,pA I P, I,5µA Output resistance Z a 5 Ω Z a 5Ω Slew rate S,2 /µs S 5 /µs Transit frequency f T, MHz f T 6 MHz Table : Characteristics of ideal and real operational amplifiers
2 . Basic Circuits Circuits with opamps mainly are based on two basic circuits, the inverting amplifier according to Fig. 2a and the noninverting amplifier according to Fig. 2b. Fig. 2: Basic circuits with operational amplifiers a) inverting amplifiers b) noninverting amplifier.. Calculation of the Inverting Amplifier The following equations result when setting mesh and node equations according to Fig.2a: Z I I I a 2 + a Z e d d a + d + a + a a 2 a a a I + I + I 2
3 The gain i of the inverting opamp circuit is: i a + ( + ) + Z (.) e In Eq. (l.l) Z e can be neglected. When solving for the gain we receive: i +, (.2) using + feedbac factor. For a large differential gain, i becomes: I ( ). (.3)..2 Calculation of the oninverting Amplifier For the noninverting amplifier according to Fig. 2b we receive:. + (.4) If the differential gain is very large, Eq. (.4) simplifies to: I with +. (.5).2 Offset The differential amplifier stages of the opamps are implemented with bipolar transistors or fieldeffect transistors. For the operating point adjustment of these transistors base or gate currents are necessary. With MOSFETS, parasitic currents occur. Examples for I P and I are given in Table. The linear average value of these currents is called idle or bias current: I B,5 ( I P + I ), (.6) while the difference is called offset current: I off I P  I. (.7) 3
4 The current I from Fig.2 through the resistors and causes voltages acting as additional input voltages. For the inverting amplifier this influence can be compensated by putting a resistor p onto the noninverting input as seen in Fig. 7. With the current I P a further input voltage is generated. It compensates for the input voltage given as I, if I P and I are equal and p. The offset current I OFF prevents a complete alignment. Although the differential voltage d of a real operational amplifier is zero, DC voltage can occur at the output of the opamp as well. In order to compensate this output voltage, DC voltage must be applied at the input. This offset voltage is called Off which is in the order of some m and subject to thermal drift. Besides the offset compensation circuits, which produce additional voltages in front of the operational amplifier, there are some opamps with special pins to perform this compensation. They can be connected to a trimming potentiometer according to Fig.3a and Fig. 3b. ot all applications of operational amplifiers do rely on correct offset compensation (e.g., pure AC amplifiers with AC coupling). In these cases an offset is omitted. Fig. 3: Some possibilities of offset compensation.3 CommonMode Gain and CommonMode ejection If both inputs of the opamp are connected to the same voltage eg, the differential voltage between both inputs is zero. In case of offsetcompensation the output voltage ag should be zero as well. In reality this will not occur. This results in the definition of a socalled commonmode gain g : g ag. (.8) e g The common mode rejection ratio G relates the openloop gain to the commonmode gain: G g. (.9) 4
5 .4 Slew ate In a real opamp both, output and internal voltages and currents are limited. Therefore, internal parasitic capacitors can only be loaded with finite currents. That is one reason for the output voltage change delay of an opamp. The amplifier responds to an input voltage step with a ramp shaped output signal. This signal's gradient is called 'slew rate' (S). The slew rate influences in particular the reaction to large signals: A sine output voltage of the amplitude Û A and the frequency f can reach a maximum gradient: d dt t f a ( ) 2π $. max A (.) For a given signal amplitude Û A the maximum frequency f max is limited by the slew rate S to: S f. 2π (.) max $ A When these boundaries are not respected, nonlinear distortions occur..5 Frequency esponse and Stability Apart from the slew rate the small signal behavior of an opamp limits the bandwidth where the circuits can be used. We have to consider that an opamp usually consists of several amplifier stages (differential amplifier stage, intermediate stage, output stage). Each of these stages shows a lowpass behavior of first order with threshold frequency ω gn. The gain of the opamp can therefore be given in complex form: ( jω ). ω ω ω ( + j ) ( + j ) ( + j ) ω ω ω g g 2 g3 (.2) The transit frequency is defined as the frequency f T of an opamp at which the gain drops below, i.e.: ( 2ω ). (.3) f T This value can only be regarded as a very preliminary criteria for the selection of opamps. In Fig. 4, the frequency and phase response are drawn logarithmically with respect to the frequency (Bode diagram) and are approximated by line segments. This approximation maes it easier to understand how frequency and phase characteristics are constructed. The linear approximation is possible only with logarithmically scaled frequency and gain axis in db..5. Stability For dimensioning opamp  circuits a further characteristic of the opamp  model is important. 5
6 The angle arg{((jω)} taes values between and 27 for positive frequencies. A real feedbac ratio according to Eq. (.2) to Eq. (.5) leads to instability of the circuit, if no suitable measures are taen. For the loop gain of these opamp circuits we can state: ( jω ). ω ω ω (+ j ) ( + j ) (+ j ) ω ω ω (.4) g Instability occurs whenever the amount of the gain '(jω) is greater or equal to, and the phase shift is large enough that the inverse feedbac degenerates to a positive feedbac, i.e. the amount of the phase shift arg { (jω)} becomes smaller or equal 8. An opamp circuit is stable for all frequencies ω that fullfil: or equivalently: (jω) < for all ω with arg{ (jω)} 8 (.5) arg{ (jω)} < 8 for all ω with (jω) > (.6) The difference between 8 and the actual angle for which (jω) is called phase reserve Φ r. It is a quantity which determines the transient behavior of an opamp circuit. g 2 g3 Fig. 4: Typical frequency and phase response of an operational amplifier depicted in a logarithmically scaled frequency and gain axis in db(bode diagram) 6
7 .5.2 Frequency esponse Compensation If we want to avoid instability of the opamp circuit, frequency response compensation has to be applied. Essentially, two principles can be introduced. The easiest method is lag compensation. Here the opamp is followed by a lowpass filter with low cutoff frequency. This causes the gain to drop below unity before achieving the critical phase shift of 8. The openloop gain results from: T ( jω ) ( jω ). (.7) ω + j ω The cutoff frequency f of the compensating lowpass depends on the feedbac ratio. If (no feedbac), then there is no need of compensation. On the other hand, if (full feedbac), then f is to be selected at least in such a way that (jω), if Φ 8 is achieved. If this is fulfilled exactly, the opamp would have no phase reserve and the amplifier circuit would operate on the boundary between stability and instability. In Fig. 5, the lowpass compensation for a phase reserve Φ in the case of < < is drawn (solid line). A compensation with a phase reserve of 65 is considered to be the optimum in most cases. Following Eq. (.5), the noninverting opamp circuit reaches the amplification >. The product (jω)* equals to unity if (jω T ) has dropped to the value /. According to Fig. 5, the compensation frequency f has to be chosen such that at the frequency f T an angular phase shift of: is reached. Φ K Φ r 8 with Φ r phase reserve (.8) For the graphic construction of f we often can assume f T > f. This means that at frequency f T the compensating lowpass filter shifts the phase by  9 additionally. The amplitude gain decreases by 2 db/decade. 7
8 Fig. 5: Frequency response compensation A few opamps have a special pin, at which an external compensating capacitor can be attached, see Fig. 6. The resistor x is integrated into the opamp chip. Fig. 6: Compensation circuit of an opamp To determine the value of the compensating capacitor from Fig. 6, Eq.(.9) is used: C 2π. (.9) f x 8
9 For most of all of opamp chips, the compensation is not achieved by insertion of an additional lowpass of low cutoff frequency, but rather by shifting the cutoff frequencies f g and f g2 (Miller compensation). Here, the compensating capacitor is not attached to ground as shown in Fig. 6, but connected to two pins of the integrated circuit of the opamp. This method of compensation maes use of the socalled "Millereffect" of the capacitive feedbac [2]. The first cutoff frequency f g of the noncompensated opamp is shifted to the frequency f M fm (.2) 2π C x which causes no change in Eq.(.9) to calculate C. However, f g disappears in the Bode diagram of the compensated opamp. The frequency f g2 is shifted with C to a higher frequency f M2 f M 2 2π x2 ( C + C C C 2. (.2) ) Here, x, x2, C and C 2 are integrated in the opamp IC s and could not be altered from outside. Their values are not found in data sheets. If these elements must be considered in the compensation, their values have to be determined by frequency response measurements. In Fig. 5, the frequency response curve for an opamp using this compensation technique is drawn (dotted line). When comparing lowpass with frequency shift compensation, a larger phase reserve can be attained with this frequency shift, and thus a larger bandwidth of the amplifier can be used. In the experiment, an opamp with " Millercompensation " is used. 2 eferences [] Tietze, Ch. Schen: Halbleiterschaltungstechni, Springer erlag Berlin, Heidelberg, ew Yor [2]. Fliege: Lineare Schaltungen mit Operationsverstärern, Springer erlag Berlin, Heidelberg, ew Yor 979 [3] W.Bauer, H.H. Wagener Bauelemente und Grundschaltungen der Eletroni, Band 2: Grundschaltungen, Carl Hanser erlag München,Wien 98. Elt 52 9
10 3 The Experiment 3. Equipment dual channel oscilloscope Hz...MHz multimeter function generator Hz... 5MHz frequency counter test circuit 3.2 Tass ) Before the experiment (this means: at home) the resistors and have to be calculated for the ideal and real opamp for a DC voltage amplification of 4dB, both in the non inverting and in the inverting circuit. Additionally, the compensating capacitor for a phase reserve Φ r 45 has to be calculated, if the cutoff frequencies of the noncompensated opamp are given by f g 6Hz and f g2 37Hz. The openloop gain is determined to db and the resistance x 9 Ω. Calculate a lowpass compensation. 2) The offset voltage of the opamp has to be measured with the circuit in Fig.7. The resistor p is used only for compensation of the bias current influence. The equation for the offset voltage given here has to be derived from the theoretical equations. Fig. 7: Circuit for offset voltage detection 3) In the circuit configuration in accordance to Fig. 7, a variable resistor has to be inserted into the socet mared "offset", and the offset voltage compensation has to be carried out. For zero indication a multimeter can be used. 4) With the circuit according to Fig. 8 the offset correction has to be done. Closed switches S and S2 and tae into account the effect. For further experiments, the offset compensation should not be changed. ow the input currents as well as the offset current have to be determined. The following currents result:
11 Fig. 8: Circuit for measuring the input currents 5) According to Fig. 9, the common mode rejection ratio G has to be measured for a frequency of Hz. This is done by using the equation: 2 GdB 2log( ) a This has to be done for, 5 and 8 Fig. 9: Circuit for measurement of the commonmode rejection ratio 6) With the circuit given in Fig., the frequency response curve of an opamp without feedbac branch can be measured and drawn on logarithmically spaced paper in a Bodediagram. Here, the openloop gain (jω) in db and the phase curve Φ(jω) should be approximated linearly. With an adjustable DC voltage of +/ 6 m the output voltage drift always has to be corrected. It should be adjusted during the experiment continuously, so that the average value of the output voltage is zero. The output signal of the opamp is connected with channel 2 of
12 the oscilloscope. The frequency of the input signal has to be read from the frequency counter. Always be aware of the connection between grounds in experimental setup, the wave generator and the oscilloscope. The input voltage has to be determined in such a way that the opamp output is not clipped. This can be detected on the oscilloscope, when the upper amplitude values of the output voltage are cut off (dynamic range exceeded) or when at higher frequencies the sine signal is distorted to a triangle signal (slew rate exceeded). Fig. : Circuit for measurement of the openloop gain (jω) 7) The experiment according to Fig. has to be implemented with a capacitor of 22pF. Then, a new Bode diagram has to be drawn. With the newly detected cutoff frequency f the value of the resistor x can be calculated. For this, Eq.(.9) is used. 8) For an inverting amplifier according to Fig., calculate the compensating capacitor C and the resistors / for a DC gain of 8.3 db. The phase reserve should be 65. Fig. : Circuit of an inverting amplifier 2
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