Current-Mode Class-D Power Amplifiers for High-Efficiency RF Applications

Transcription

1 2480 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 49, NO. 12, DECEMBER 2001 Current-Mode Class-D Power Amplifiers for High-Efficiency RF Applications Hidenori Kobayashi, Member, IEEE, Jeffrey M. Hinrichs, and Peter M. Asbeck, Fellow, IEEE Abstract We show that current-mode class-d (CMCD) power amplifiers can achieve high efficiency at RF frequencies. In contrast with conventional voltage-mode class-d amplifiers, the CMCD features zero voltage switching, which eliminates the output capacitance discharge loss. Experimental CMCD amplifiers with 76.3% power-added efficiency (PAE) at 290-mW output and 71.3% PAE at 870-mW output are demonstrated using GaAs FETs at 900 MHz. Index Terms Analog circuits, MESFET power amplifiers, power amplifiers, wireless communications. I. INTRODUCTION SWITCHING-MODE power amplifiers have the potential for high efficiency, with drain efficiency theoretically approaching 100% [1]. However, available transistors are not ideal switches because of parasitic reactances, finite on-resistance, and limited gain. Class-D amplifiers are popular switching-mode amplifiers for audio frequencies. However, they exhibit poor efficiency at higher frequencies because parasitic reactances lead to substantial loss [2]. If transistor switches have output shunt capacitance (and not too large series inductance), then energy is dissipated each cycle, where is the voltage across the transistor at switch closure. The class-e approach can solve these problems [3]. When the transistors turn on and the switches close, the voltage across the devices is always zero so output capacitance discharge loss is avoided [i.e., zero-voltage-switching (ZVS)]. This approach works well up to the megahertz frequency range, but is less effective in the gigahertz range. Uncertain duty cycle due to sinusoidal driving waveforms, nonlinear capacitances, and other parasitic components degrade the class-e operation. Multiharmonic load termination [4], class-f [5], and inverse class-f (tuned class B) [6], [7] are other less well-developed techniques to achieve ZVS (or zero-current-switching (ZCS), a related condition that minimize series inductances losses). Here, we describe another approach that can achieve ZVS, the current-mode class-d (CMCD) amplifier. This is related to the more frequently used voltage-mode class-d amplifier by an interchange between voltage and current waveforms. In this Manuscript received March 30, 2001; revised August 17, This work was supported by the University of California at San Diego Center for Wireless Communications and by the Office of Naval Research (monitored by Dr. J. Zolper). H. Kobayashi and P. M. Asbeck are with the Department of Electrical and Computer Engineering, University of California at San Diego, La Jolla, CA USA ( kobayashi@ece.ucsd.edu; asbeck@ece.ucsd.edu). J. M. Hinrichs is with the Radio Systems Division, TRW, San Diego, CA USA ( jeffrey.hinrichs@trw.com). Publisher Item Identifier S (01) Fig. 1. VMCD circuits and waveforms. VMCD circuit 1. VMCD circuit 2. (c) Current and voltage waveform. paper, the CMCD is demonstrated with GaAs FETs to achieve 75% power-added efficiency (PAE) at 900 MHz. II. CONCEPT OF CMCD Fig. 1 shows the basic schematic of a voltage mode class-d (VMCD) amplifier (sometimes referred to only as class-d ). The fundamental circuit and a more practical realization are shown in Fig 1 and Fig 1, respectively. Two transistors are driven 180 out-of-phase. A series filter is employed, with a resonant frequency set to the center frequency of the signal. Ideal drain voltage and drain current waveforms are shown in Fig 1(c). The voltage across the transistors is a square wave and the transistor current becomes a half-wave rectified sine wave. However, if the transistors have some output capacitance, this capacitance must be charged or discharged to or ground through the transistor. That means that the voltage waveform cannot have a perfect square shape and some transient current spikes occur when the transistor turns on. The overlapping of voltage and current cannot be avoided. This loss can be described by an energy loss per cycle with given by where is the drain source capacitance and is the drain source voltage when the transistor is turned on. This output capacitance loss is independent of the channel resistance of the transistor and becomes the dominant loss mechanism at (1) /01$ IEEE

2 KOBAYASHI et al.: CMCD POWER AMPLIFIERS FOR HIGH-EFFICIENCY RF APPLICATIONS 2481 loss is eliminated and inductance loss is unavoidable without ZCS at node A in Fig. 3. In practice, transistors are modeled more accurately as Fig. 3 than Fig. 3. Additionally, in suitable integrated-circuit (IC) implementation, the drain inductance can be minimized. Another advantage of the CMCD amplifier is ease of implementation at high frequency. The VMCD amplifier typically requires complementary devices or a center-tapped transformer to function properly. The CMCD amplifier only requires a simple balun structure. Fig. 2. CMCD circuits and waveforms. CMCD circuit 1. CMCD circuit 2. (c) Current and voltage waveform. III. THEORETICAL ANALYSIS OF CMCD In an ideal situation, the drain source voltage, are half-rectified sine waves and the drain currents, are square waves. The current sources are ideal chokes tied to voltage source, as shown in Fig. 2. The peak drain voltage can be calculated by (2a) (2b) Let us suppose the transformer ratio : is 2 : 1 in Fig. 2. Power output can be calculated by (3) Fig. 3. Switch models. In an ideal situation, and are the same, thus, the dc drain current can be expressed as (4) high frequencies starting at hundreds of megahertz. This is one of the reasons that class-d is not a popular gigahertz amplifier. Fig. 2 and shows the CMCD circuits and Fig. 2(c) shows the waveforms for each transistor. For the CMCD, we use current sources instead of voltage sources, and the two switching transistors control the current instead of the voltage. There is a parallel-connected filter, with resonant frequency set to the carrier frequency. Due to the filter resonance, there is no voltage across the transistors at each switching time and so-called ZVS is achieved. Even if the transistors have some output capacitance, the output capacitance can become part of the output parallel filter; voltage waveforms are still as shown in Fig. 2(c). This ZVS feature is a key advantage of the CMCD architecture. As mentioned above, another important switching condition, i.e., ZCS, avoids inductance losses by making the current zero at the instant of switching. VMCD and class-f achieve this condition. However, ZCS is less important than ZVS. Fig. 3 and shows examples of switches with parasitic inductance and capacitance in different topologies. In Fig. 3, the capacitance voltage and the switch voltage are identical. Thus, if there is voltage when the switches turn on, energy is lost, and ZVS is needed at node A to avoid this loss. However, current can continue to flow to the capacitor when switches turn off and the inductance loss is eliminated. On the contrary, the capacitance However, (2a) (4) are too ideal to properly represent real RF amplifiers. High-order harmonic frequencies are needed for perfect half-rectified sine or square waves. In RF frequency operation, it is very difficult to control impedance for the high-order harmonics due to the frequency dependence of each component. A more realistic estimate results from ignoring harmonic components of and greater than third order. The parallel filter and balanced circuits are still considered ideal in this calculation. In this case, the load line appears open to second harmonics and short to third harmonics. The voltage and current can be described as (5a) (5b) We suppose the transistors drain voltage and current never becomes negative. This can be expressed by To represent the switching operation, we consider and with the restriction that the waveform must be monotonically increasing during one half cycle and monotonically decreasing (6)

3 2482 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 49, NO. 12, DECEMBER 2001 during the other half cycle. The coefficients that achieve the above condition can be calculated from (7a) (7b) From (5)(7b),,,, and can be determined to be (8a) (8b) (8c) (8d) If or are increased from these values, the waveforms develop multiple peaks and valleys. Suppose the transformer ratio : is 2 : 1 in Fig. 2. The relationship of fundamental current and voltage can be expressed as (9) Total dc input current, dc input power, and output power are given by (10a) (10b) (10c) Transistors (switches) are not ideal in this case. There are simultaneous drain currents and voltages causing the transistors to dissipate energy. Waveforms of voltage and current are shown in Fig. 4. Table I shows corresponding values, along with efficiency for the case V and. Further analysis, including higher harmonics up to fifth harmonics, can be considered using (11a) and (11b), which are derived by similar considerations as follows: (11a) (11b) Results for this case are also shown in Fig. 4 and Table I. As we expect, efficiency increases if appropriate higher order harmonic terminations are used. Also, we can see that output power is very sensitive to harmonics terminations even when the funda- Fig. 4. CMCD analytical voltage and current waveforms. Voltage waveforms. Current waveforms (V = 3 V, R =50). mental load is fixed. On the contrary, dc input current (dc input power) is almost constant. IV. PA DESIGN AND SIMULATION The CMCD amplifier was implemented at 900 MHz with commercially available GaAs MESFETs (Infineon CLY5). Harmonic-balance simulations were performed with HP ADS. Statz MESFET model parameters of the chip transistor and the SOT223 package model were used. The CMCD circuits are redesigned using a 50- coaxial balun and are shown in Fig. 5 (minor package parasitic components are omitted in this figure). Due to practical board layout constraints, 25- transmission lines and are inserted between components. We optimized the lengths of and, and the values of and to achieve ZVS at node A (not at the drain terminal of the transistor package) by simulation. Fig. 6 shows simulated voltage and current waveforms. Solid lines indicate chip drain source voltage and current, and dotted lines are voltage and current including package parasitic components (mainly and ). Due to device parasitic drain source capacitance, the drain current does not look like a square wave, but becomes negative when the transistor is off. If we extract all output capacitances (including device capacitances), the current waveform should be more rectangular. In this simulation, we achieved 84% PAE at 25.4 dbm (0.35 W) with V.

4 KOBAYASHI et al.: CMCD POWER AMPLIFIERS FOR HIGH-EFFICIENCY RF APPLICATIONS 2483 TABLE I THEORETICAL EFFICIENCY (V = 3 V AND R = 50 ) Fig. 5. CMCD circuits for high frequency. V. EXPERIMENTAL RESULTS A picture of the CMCD power amplifier is shown in Fig. 7. A Mini-Circuits 180 hybrid is used to generate the balanced signal for the input and input impedance-matching circuits are tuned for maximum gain. and in Fig. 5 are optimized for higher efficiency. Fig. 8 shows the efficiency and RF output power when the gate-bias voltage is 1.7 V and is 3 V. The input power does not include the 180 -hybrid loss, but the output power includes output balun loss. The drain efficiency reaches 80% when the input power is above 11 dbm. The PAE reaches a peak value of 76.3% under these conditions. The corresponding power gain is approximately 12 db. The gain decreases with increasing input power since the output power is nearly constant (and determined only by the power supply voltage). Fig. 9 shows the output power and efficiency when is 5 V. Drain efficiency of 75.6% and PAE of 72.5% was achieved with the input power set to 14.7 dbm and output set to 28.6 dbm (0.73 W). In both cases, output power is much smaller than the theoretical value we expected. One of the reasons is that we can only achieve up to third harmonic terminations, and even second and third harmonic terminations are not perfect because of device parasitic Fig. 6. Simulated voltage and current waveforms. Voltage waveform. Current waveform (operational frequency = 900 MHz). Fig. 7. CMCD PA circuit. components. We believe that the nonideal RF choke especially decreases output power.

5 2484 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 49, NO. 12, DECEMBER 2001 Fig. 8. Measured output power and efficiency (V = 3 V). Fig. 11. Output power and efficiency versus VDD (P =12:7 dbm). Fig. 9. Measured output power and efficiency (V =5V). Fig. 10. Output and PAE versus gate bias voltage (V =3 V, P = 12:7 dbm). In the regime of low input power (less than 6 dbm for low, less than 10 dbm for high ), the CMCD works as a class-b or class-ab push pull amplifier (because the parallel filter is an open circuit at the operational frequency). Thus, the structure is an approximately linear amplifier in this regime. Fig. 10 shows the gate-bias voltage versus dc input power, output power, and PAE at V and dbm. PAE stays over 70% when dc bias is below 1.6 V. In the regime of dc bias 1.6 V, the transistor cannot be in the off state and losses increase rapidly. Fig. 11 shows versus output power and PAE in a decibel scale at dbm. and are linear on a decibel scale, which means is proportional to the square of. At higher s, the output power becomes compressed a little. This is because the transistor power gain is limited and the device does not work well as a switch in this region. Fig. 12 shows the frequency dependence of output power and PAE. The dotted line shows the CMCD optimized for 900 MHz. The solid line shows the case where the input matching Fig. 12. Output power and PAE versus frequency. circuits were tuned with variable capacitors at every frequency. This eliminates some of the frequency dependence of the input matching circuits. In this case, the power output becomes almost flat and PAE is greatly improved. This indicates that much of the circuit s bandwidth is a result of input impedance circuits. From the standpoint of harmonic tuning, second harmonics should be presented with an open and third harmonics should be shorted for the CMCD. Balanced circuits are intrinsically open at even harmonics and output capacitances of the transistors and filter tend to be shorted at high frequency. These two are less dependent on the operational frequency and the output load impedance. We believe the CMCD is capable of a large bandwidth when broad-band impedance input matching is used. VI. SYSTEM APPLICATION The output of the CMCD amplifier has a nearly constant envelope, dictated by the power supply voltage. Thus, it is appropriate for use with modulated signals of constant envelope. Fig. 13 shows the output spectrum when a global system for mobile communications (GSM) modulated input signal was used

6 KOBAYASHI et al.: CMCD POWER AMPLIFIERS FOR HIGH-EFFICIENCY RF APPLICATIONS 2485 Hidenori Kobayashi (M 00) received the B.S. and M.S. degrees in electric engineering from Meiji University, Kanagawa, Japan, in 1994 and 1996, respectively. In 1996, he joined the Semiconductor Division, Fuji Electric Company Ltd., Tokyo, Japan, where he was engaged in research and development of analog and digital IC design. In 1999, he joined the Center for Wireless Communications (CWC), Department of Electrical and Computer Engineering, University of California at San Diego, La Jolla, where he is currently a Visiting Scholar. He is currently involved with RF power-amplifier circuit design. Fig. 13. Output spectrum of the GSM signal. ( dbm, and 5). No major distortion is observed in both cases. The drain efficiency and PAE values were equal to those measured for sinusoidal signals, as shown in Figs. 8 and 9. To apply the CMCD to QPSK or related waveforms with a time-varying envelope, a modulated power supply voltage is required (such as in the Kahn technique). VII. CONCLUSION We have demonstrated for the first time the possibility of current-mode class-d power amplifiers at RF frequencies. Experimental results confirm that high efficiency can be attained and show the possibility of high bandwidth. REFERENCES [1] Krauss, Bostlan, and Rabb, Solid State Radio Engineering. New York: Wiley, [2] S. El-Hamamsy, Design of high-efficiency RF class D power amplifier, IEEE Trans. Power Electron., vol. 9, pp , May [3] N. O. Sokal and A. D. Sokal, Class E-A new class of high efficiency tuned single-ended power amplifiers, IEEE J. Solid State Circuits, vol. SC-10, pp , June [4] J. Staudinger, Multiharmonic load termination effects on GaAs MESFET power amplifiers, Microwave J., pp , Apr [5] C. Duvanaud, S. Dietsche, G. Pataut, and J. Obregon, High-efficient class F GaAs FET amplifier operating with very low bias voltages for use in mobile telephones at 1.75 GHz, IEEE Microwave Guided Wave Lett., vol. 3, pp , Aug [6] A. Inoue, T. Heima, A. Ohta, R. Hattori, and Y. Mitsui, Analysis of class-f and inverse class-f amplifiers, in Proc. IEEE MTT-S Int. Microwave Symp. Dig., vol. 2, 2000, pp [7] C. J. Wei, P. DiCarlo, Y. A. Tkachenko, R. McMorrow, and D. Bartle, Analysis and experimental waveform study on inverse class-f mode of microwave power FETs, in Proc. IEEE MTT-S Int. Microwave Symp. Dig., vol. 1, 2000, pp Jeffrey M. Hinrichs received the B.S. and M.S. degrees in electrical engineering from the University of California at San Diego, La Jolla, in 1999 and 2001, respectively. Since 2000, he has been involved in research with the High Speed Device Group, University of California at San Diego. His research interests include digital signal processing, communication theory, and their application to the generation of RF signals utilizing high-speed digital to analog conversion techniques. He has studied switching-mode power amplifiers driven with delta sigma modulated data streams, with potential application to wireless transmitter design. He has been an Intern for five summers at the Los Alamos National Laboratory, Los Alamos, NM, where he has contributed to projects ranging from scientific visualization to microwave pulse power. He is currently a member of the systems engineering team for the Radio Systems Division, TRW, San Diego, CA. Peter M. Asbeck (M 75 SM 97 F 00) received the B.S. and Ph.D. degrees in electrical engineering from the Massachusetts Institute of Technology (MIT), Cambridge, in 1969 and in 1975, respectively. He was with the Sarnoff Research Center, Princeton, NJ, and at the Philips Laboratory, Briarcliff Manor, NY, where he was involved in the areas of quantum electronics and GaAlAs/GaAs laser physics and applications. In 1978, he joined the Rockwell International Science Center, where he was involved in the development of high-speed devices and circuits using III V compounds and heterojunctions. He pioneered the effort to develop HBTs based on GaAlAs/GaAs and InAlAs/InGaAs materials, and has contributed widely in the areas of physics, fabrication, and applications of these devices. In 1991, he joined the University of California at San Diego (UCSD), La Jolla, as Professor in the Department of Electrical and Computer Engineering. His research interests are in development of high-speed heterojunction transistors and opto-electronic devices, and their circuit applications. Dr. Asbeck is a distinguished lecturer of the IEEE Electron Device Society and of the IEEE Microwave Theory and Techniques Society (IEEE MTT-S). He is a member of the Department of Defense Advisory Group on Electron Devices. He was the general chairman of the 1996 Device Research Conference, and the chairman of the 1999 IEEE Topical Workshop on Power Amplifiers for Wireless Communications.