LT3478/LT A Monolithic LED Drivers with True Color PWM Dimming DESCRIPTIO FEATURES APPLICATIO S TYPICAL APPLICATIO

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1 FEATURES True Color Dimming Delivers Constant LED Color with Up to 3:1 Range Wide Input Voltage Range: 2.8V to 36V 4.5A, 6mΩ, 42V Internal Switch Drives LEDs in Boost, Buck-Boost or Buck Modes Integrated Resistors for Inductor and LED Current Sensing Program LED Current: 1mA to 15mA (LT3478-1) (1mV to 15mV)/R SENSE (LT3478) Program LED Current De-Rating vs Temperature Separate Inductor Supply Input Inrush Current Protection Programmable Soft-Start Fixed Frequency Operation from 2kHz to 2.25MHz Open LED Protection (Programmable OVP) Accurate Shutdown/UVLO Threshold with Programmable Hysteresis 16-Pin Thermally Enhanced TSSOP Package APPLICATIO S U High Power LED Driver Automotive Lighting, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Patents Pending. TYPICAL APPLICATIO 4.7µF 45.3k 54.9k 13k Automotive TFT LCD Backlight 8V TO 16V DIMMING CONTROL U V S L SW SHDN V REF OUT LT LED OVPSET CTRL1 SS V C R T 1µF 1µH.1µF 69.8k 1µF.1Ω R SENSE (LT3478) 7mA 15W 6 LEDs (WHITE) EFFICIENCY (%) I LED = 7mA f OSC = 5kHz DUTY CYCLE = 1% 1 4.5A Monolithic LED Drivers with True Color Dimming DESCRIPTIO U The LT 3478/LT are 4.5A step-up DC/DC converters designed to drive LEDs with a constant current over a wide programmable range. Series connection of the LEDs provides identical LED currents for uniform brightness without the need for ballast resistors and expensive factory calibration. The LT reduces external component count and cost by integrating the LED current sense resistor. The LT3478 uses an external sense resistor to extend the maximum programmable LED current beyond 1A and also to achieve greater accuracy when programming low LED currents. Operating frequency can be set with an external resistor from 2kHz up to 2.25MHz. Unique circuitry allows a dimming range up to 3:1 while maintaining constant LED color. The are ideal for high power LED driver applications such as automotive TFT LCD backlights, courtesy lighting and heads-up displays. One of two CTRL pins can be used to program maximum LED current. The other CTRL pin can be used to program a reduction in maximum LED current vs temperature to maximize LED usage and improve reliability. Additional features include inrush current protection, programmable open LED protection and programmable soft-start. Each part is available in a 16-pin thermally enhanced TSSOP Package. Effi ciency vs 6 LEDs LUXEON III (WHITE) 12 (V) TA1b 3478 TA1 1

2 ABSOLUTE AXI U RATI GS (Note 1) W W W SW...42V V OUT, LED...42V, V S, V L, S H D N (Note 5)...36V...15V CTRL1, 2...6V SS, R T, V C, V REF, OVPSET...2V Operating Junction Temperature Range (Notes 2, 3, 4)... 4 C to 125 C Storage Temperature Range C to 15 C Lead Temperature (Soldering, 1 Sec)... 3 C U U U W PACKAGE/ORDER I FOR ATIO SW SW V S L V OUT LED OVPSET TOP VIEW SS 15 R T CTRL1 11 SHDN 1 V REF 9 V C FE PACKAGE 16-LEAD PLASTIC TSSOP T JMAX = 125 C, θ JA = 35 C/W EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB. ORDER PART NUMBER LT3478EFE LT3478EFE-1 LT3478IFE LT3478IFE-1 FE PART MARKING 3478FE 3478FE FE 3478FE-1 Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The denotes the specifi cations which apply over the full operating temperature range, otherwise specifi cations are at T A = 25 C. SW = open, = V S = L = V OUT = S H D N = 2.7V, LED = open, SS = open, = CTRL1, = 1.25V, V REF = open, V C = open, R T = 31.6k. PARAMETER CONDITIONS MIN TYP MAX UNITS Minimum Operating Voltage (Rising) V Operational Input Voltage V S (Note 5) V V Quiescent Current V C = V (No Switching) 6.1 ma Shutdown Current S H D N = V 3 6 µa S H D N Pin Threshold (V SD_µp ) (Micropower) V S H D N Pin Threshold (V SD_UVLO ) (Switching) V S H D N Pin Current S H D N = V SD_UVLO 5mV S H D N = V SD_UVLO + 5mV µa µa V REF Voltage I(V REF ) = µa, V C = V V V REF Line Regulation I(V REF ) = µa, 2.7V < < 36V.5.15 %/V V REF Load Regulation < I(V REF ) < 1µA (Max) 8 12 mv Frequency: f OSC 2kHz R T = 2k MHz Frequency: f OSC 1MHz R T = 31.6k MHz 2

3 ELECTRICAL CHARACTERISTICS The denotes the specifi cations which apply over the full operating temperature range, otherwise specifi cations are at T A = 25 C. SW = open, = V S = L = V OUT = S H D N = 2.7V, LED = open, SS = open, = CTRL1, = 1.25V, V REF = open, V C = open, R T = 31.6k. PARAMETER CONDITIONS MIN TYP MAX UNITS Frequency: f OSC 2.25MHz R T = 9.9k MHz Line Regulation f OSC R T = 31.6k, 2.7V < < 36V.5.2 %/V Nominal R T Pin Voltage.64 V Maximum Duty Cycle R T = 31.6k R T = 2k R T = 9.9k LED Current to V C Current Gain (Note 6) 77 µa/a LED Current to V C Voltage Gain (Note 6) 4 V/A V C to Switch Current Gain 13 A/V V C Source Current (Out of Pin) CTRL1 =.4V, V C = 1V 4 µa V C Sink Current CTRL1 = V, V C = 1V 4 µa V C Switching Threshold.65 V V C High Level (V OH ) CTRL1 =.4V 1.5 V V C Low Level (V OL ) CTRL1 = V.2 V Inductor Current Limit 2.7V < V S < 36V A Switch Current Limit A Switch V CE SAT I SW = 4.5A 27 mv Switch Leakage Current SW = 42V, V C = V 1 µa V OUT Overvoltage Protection (OVP) (Rising) OVPSET = 1V OVPSET =.3V Full Scale LED Current (LT3478-1) CTRL1 = V REF, Current Out of LED Pin ma 7mA LED Current (LT3478-1) CTRL1 = 7mV, Current Out of LED Pin ma 35mA LED Current (LT3478-1) CTRL1 = 35mV, Current Out of LED Pin ma 1mA LED Current (LT3478-1) CTRL1 = 1mV, Current Out of LED Pin ma Full Scale LED Current V SENSE (LT3478) CTRL1 = V REF, V SENSE = V VOUT V LED mv CTRL1 = 7mV, V SENSE (LT3478) CTRL1 = 7mV, V SENSE = V VOUT V LED mv CTRL1 = 35mV, V SENSE (LT3478) CTRL1 = 35mV, V SENSE = V VOUT V LED mv CTRL1 = 1mV, V SENSE (LT3478) CTRL1 = 1mV, V SENSE = V VOUT V LED mv CTRL1, 2 Input Currents CTRL1 = 1mV, = 1.25V or 4 na = 1mV, CTRL1 = 1.25V (Current Out of Pin) OVPSET Input Current OVPSET = 1V, V OUT = 41V (Current Out of Pin) 2 na Switching Threshold V V C Pin Current in Mode V C = 1V, = 1 5 na OUT Pin Current in Mode = 1 1 na SS Low Level (V OL ) I (SS) = 2µA.15 V SS Reset Threshold V C = V.25 V SS High Level (V OH ) V C = V 1.5 V Soft-Start (SS) Pin Charge Current SS = 1V, Current Out of Pin, V C = V 12 µa Soft-Start (SS) Pin Discharge Current SS =.5V, V C = V 35 µa % % % V V 3

4 ELECTRICAL CHARACTERISTICS Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3478EFE/LT3478EFE-1 are guaranteed to meet performance specifications from C to 125 C junction temperature. Specifications over the 4 C to 125 C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3478IFE/LT3478IFE-1 are guaranteed over the full 4 C to 125 C operating junction temperature range. Note 3: This IC includes over-temperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125 C when over-temperature protection is active. Continuous operation above the specifi ed maximum operating junction temperature may impair device reliability. Note 4: For maximum operating ambient temperature, see the Thermal Calculations section in the Applications Information section. Note 5: The maximum operational voltage for is limited by thermal and effi ciency considerations. Power switch base current is delivered from and should therefore be driven from the lowest available power supply in the system. See Thermal Calculations in the Applications Information section. Note 6: For LT3478, parameter scales (R SENSE /.1Ω). TYPICAL PERFOR A CE CHARACTERISTICS LED CURRENT (ma) LED Current vs CTRL1 T A = 25 C = V REF (FOR LT3478 SCALE BY.1Ω/R SENSE ) LT U W V REF LED CURRENT (ma) LED Current vs Temperature (FOR LT3478 SCALE BY.1Ω/R SENSE ) I LED = 15mA, CTRL1 = = V REF LT I LED = 1mA, CTRL1 = 1mV, = V REF LED CURRENT (ma) LED Current vs Duty Cycle Wide Dimming Range (3:1) T A = 25 C = V S = 12V 6 LEDS AT 5mA FREQ = 1Hz CTRL1 =.5V = V REF F OSC = 1.6MHz L = 2.2µH CTRL1 (V) DUTY CYCLE (%) 3478 G G G3 5 CTRL1 Pin Current vs Temperature 24 Switch V CE (SAT) vs Switch Current T A = 25 C 7. Switch and Inductor Peak Current Limits vs Temperature CTRL1 PIN CURRENT X ( 1) (na) = V REF CTRL1 AND PINS INTERCHANGEABLE CTRL1 =.1V CTRL1 =.35V CTRL1 =.7V SWITCH V CE (SAT) (mv) CURRENT LIMIT (A) SWITCH INDUCTOR 5 CTRL1 =.9V SWITCH CURRENT (A) G G G6 4

5 TYPICAL PERFOR A CE CHARACTERISTICS U W 1.28 V REF vs Temperature 1.6 S H D N Threshold vs Temperature 15 S H D N Pin (Hysteresis) Current vs Temperature V REF (V) SHDN (V) SHDN PIN CURRENT (µa) 1 5 JUST BEFORE PART TURNS ON AFTER PART TURNS ON G G G9 CURRENT (µa) Shutdown Current vs Quiescent Current vs Temperature Quiescent Current vs Temperature 5 SHDN = V = 36V = 2V = 2.8V CURRENT (ma) T A = 25 C V C = V (V) CURRENT (ma) = 2.8V V C = V G G G12 PIN CURRENT (µa) 4 2 V S, L, SW Shutdown Currents vs Temperature 5 SHDN = V V S = L = SW = 36V I(V S PIN) = I(L PIN) I(SW PIN) G18 SWITCH PEAK CURRENT LIMIT (A) Switch Peak Current Limit vs Duty Cycle T A = 25 C DUTY CYCLE (%) 3478 G19 5

6 TYPICAL PERFOR A CE CHARACTERISTICS U W SWITCHING FREQUENCY (khz) 1 Switching Frequency vs R T 1 T A = 25 C SWITCHING FREQUENCY (MHz) Switching Frequency vs Temperature R T = 31.6k V OUT CLAMP (V) Open-Circuit Output Clamp Voltage vs Temperature OVPSET = 1V R T (kω) G G G15 14 SS Pin Charge Current vs Temperature 1.8 V C Pin Active and Clamp Voltages vs Temperature SS PIN CURRENT (µa) (OUT OF PIN) V C (V) V C CLAMP V C ACTIVE THRESHOLD G G17 6

7 PI FU CTIO S U U U SW (Pins 1, 2): Switch Pin. Collector of the internal NPN power switch. Both pins are fused together inside the IC. Connect the inductor and diode here and minimize the metal trace area connected to this pin to minimize EMI. (Pin 3): Input Supply. Must be locally bypassed with a capacitor to ground. V S (Pin 4): Inductor Supply. Must be locally bypassed with a capacitor to ground. Can be shorted to if only one supply is available (see L (Pin 5) function). L (Pin 5): Inductor Pin. An internal resistor between V S and L pins monitors inductor current to protect against inrush current. Exceeding 6A immediately turns off the internal NPN power switch and discharges the soft-start pin. Input current monitoring can be disabled by connecting the inductor power supply directly to the L pin and leaving the V S pin open (requires local bypass capacitor to GND on L pin; not V S pin). V OUT (Pin 6): Output voltage of the converter. Connect a capacitor from this pin to ground. Internal circuitry monitors V OUT for protection against open LED faults. LED (Pin 7): Connect the LED string from this pin to ground. An internal (LT3478-1)/external(LT3478) resistor between the V OUT and LED pins senses LED current for accurate control. OVPSET (Pin 8): Programs V OUT overvoltage protection level (OVP) to protect against open LED faults. OVP = (OVPSET 41)V. OVPSET range is.3v to 1V for an OVP range of typically 12.3V to 41V. V C (Pin 9): Output of the transconductance error amplifi er and compensation pin for the converter regulation loop. V REF (Pin 1): Bandgap Voltage Reference. This pin can supply up to 1µA. Can be used to program CTRL1,, OVPSET pin voltages using resistor dividers to ground. S H D N (Pin 11): The S H D N pin has an accurate 1.4V threshold and can be used to program an undervoltage lockout (UVLO) threshold for system input supply using a resistor divider from supply to ground. A 1µA pin current hysteresis allows programming of undervoltage lockout (UVLO) hysteresis. S H D N above 1.4V turns the part on and removes a 1µA sink current from the pin. S H D N = V reduces current < 3µA. S H D N can be directly connected to. If left open circuit the part will be turned off. CTRL1 (Pin 12): CTRL1 pin voltage is used to program maximum LED current ( = V REF ). CTRL1 voltage can be set by a resistor divider from V REF or an external voltage source. Maximum LED current is given by: (LT3478-1) Max LED Current = Min(CTRL1, 1.5) Amps ( LT3478) Max LED Current = 1. Min( CTRL, 15. ) Amps R SE NSE (linear for.1v < CTRL1<.95V ; = V REF ) For maximum LED current, short CTRL1 and pins to V REF. (Pin 13): The pin is available for programming a decrease in LED current versus temperature (setting temperature breakpoint and slope). This feature allows the output LED(s) to be programmed for maximum allowable current without damage at higher temperatures. This maximizes LED usage and increases reliability. A voltage with negative temperature coeffi cient is created using an external resistor divider from V REF with temperature dependant resistance. If not used, should be tied to V REF. (Pin 14): Input pin for dimming control. Above 1V allows converter switching and below 1V disables switching with V C pin level maintained. With an external MOSFET placed in series with the ground side of the LED string, a signal driving the pin and MOSFET gate provides accurate dimming control. The signal can be driven from V to 15V. If unused, the pin should be connected to V REF. R T (Pin 15): A resistor to ground programs switching frequency between 2kHz and 2.25MHz. SS (Pin 16): Soft-Start Pin. Placing a capacitor here programs soft-start timing to limit inductor inrush current during start-up due to the converter. When inductor current 7

8 PI FU CTIO S U U U exceeds 6A or V OUT exceeds OVP, an internal soft-start latch is set, the power NPN is immediately turned off and the SS pin is discharged. The soft-start latch is also set if and/or S H D N do not meet their turn on thresholds. The SS pin only recharges when all faults are removed and the pin has been discharged below.25v. Exposed Pad (Pin 17): The ground for the IC and the converter. The FE package has an Exposed Pad underneath the IC which is the best path for heat out of the package. Pin 17 should be soldered to a continuous copper ground plane under the device to reduce die temperature and increase the power capability of the. BLOCK DIAGRA W V 3 SHDN + 1µA REF 1.24V UVLO V S mΩ 57mV L 5 INRUSH CURRENT PROTECTION + V C SS 16 SOFT-START OVERVOLTAGE DETECT OVPSET SW 1, 2 1Ω R SENSE.1Ω (INTERNAL FOR LT3478-1) V OUT 6 LED 7 R SENSE (EXTERNAL FOR LT3478) V REF 1 DETECT OSC S Q Q1 R LED CTRL V Ω Q2 SLOPE COMP Σ GM + 14 LED LED LED R S 1V TO OVERVOLTAGE DETECT CIRCUIT 8 OVPSET 15 R T 17 EXPOSED PAD (GND) 9 V C 3478 F1 Figure 1 8

9 OPERATIO U The are high powered LED drivers with a 42V, 4.5A internal switch and the ability to drive LEDs with up to 15mA for LT and up to 15mV/R SENSE for LT3478. The work similarly to a conventional current mode boost converter but use LED current (instead of output voltage) as feedback for the control loop. The Block Diagram in Figure 1 shows the major functions of the. For the part to turn on, the pin must exceed 2.8V and the S H D N pin must exceed 1.4V. The S H D N pin threshold allows programming of an undervoltage lockout (UVLO) threshold for the system input supply using a simple resistor divider. A 1µA current flows into the S H D N pin before part turn on and is removed after part turn on. This current hysteresis allows programming of hysteresis for the UVLO threshold. See Shutdown Pin and Programming Undervoltage Lockout in the Applications Information Section. For micropower shutdown the S H D N pin at V reduces supply current to approximately 3µA. Each LED driver is a current mode step-up switching regulator. A regulation point is achieved when the boosted output voltage V OUT across the output LED(s) is high enough to create current in the LED(s) equal to the programmed LED current. A sense resistor connected in series with the LED(s) provides feedback of LED current to the converter loop. The basic loop uses a pulse from an internal oscillator to set the RS flip-flop and turn on the internal power NPN switch Q1 connected between the switch pin, SW, and ground. Current increases in the external inductor until switch current limit is exceeded or until the oscillator reaches its maximum duty cycle. The switch is then turned off, causing inductor current to lift the SW pin and turn on an external Schottky diode connected to the output. Inductor current flows via the Schottky diode charging the output capacitor. The switch is turned back on at the next reset cycle of the internal oscillator. During normal operation the V C voltage controls the peak switch current limit and hence the inductor current available to the output LED(s). As with all current mode converters, slope compensation is added to the control path to ensure stability. The CTRL1 pin is used to program maximum LED current via Q2. The pin can be used to program a decrease in LED current versus temperature for maximum reliability and utilization of the LED(s). A voltage with negative temperature coeffi cient can be created using an external resistor divider from V REF with temperature dependant resistance. Unused is tied to V REF. For True Color dimming, the provide up to a 3:1 wide dimming range by allowing the duty cycle of the pin (connected to the IC and an external N-channel MOSFET in series with the LED(s)) to be reduced from 1% to as low as.33% for a frequency of 1Hz. Dimming by duty cycle, allows for constant LED color to be maintained over the entire dimming range. For robust operation, the monitor system performance for any of the following faults : or S H D N pin voltages too low and/or inductor current too high and/or boosted output voltage too high. On detection of any of these faults, the stop switching immediately and a soft-start latch is set discharging the SS pin (see Timing Diagram for SS pin in Figure 11). All faults are detected internally and do not require external components. When all faults no longer exist, an internal 12µA supply charges the SS pin with a timing programmed using a single external capacitor. A gradual ramp up of SS pin voltage limits switch current during startup. For optimum component sizing, duty cycle range and efficiency the allow for a separate inductor supply V S and for switching frequency to be programmed from 2kHz up to 2.25MHz using a resistor from the R T pin to ground. The advantages of these options are covered in the Applications Informations section. 9

10 APPLICATIO S I FOR Inductor Selection ATIO U W U U Several inductors that work well with the are listed in Table 1. However, there are many other manufacturers and inductors that can be used. Consult each manufacturer for more detailed information and their entire range of parts. Ferrite cores should be used to obtain the best efficiency. Choose an inductor that can handle the necessary peak current without saturating. Also ensure that the inductor has a low DCR (copper-wire resistance) to minimize I 2 R power losses. Values between 4.7µH and 22µH will suffice for most applications. Inductor manufacturers specify the maximum current rating as the current where inductance falls by a given percentage of its nominal value. An inductor can pass a current greater than its rated value without damaging it. Aggressive designs where board space is precious will exceed the maximum current rating of the inductor to save space. Consult each manufacturer to determine how the maximum inductor current is measured and how much more current the inductor can reliably conduct. Capacitor Selection Low ESR (equivalent series resistance) ceramic capacitors should be used at the output to minimize the output ripple voltage. Use only X5R or X7R dielectrics, as these materials retain their capacitance over wider voltage and temperature ranges than other dielectrics. A 4.7µF to 1µF output capacitor is suffi cient for most high output current designs. Some suggested manufacturers are listed in Table 2. Diode Selection Schottky diodes, with their low forward voltage drop and fast switching speed, are ideal for applications. Table 3 lists several Schottky diodes that work well. The diode s average current rating must exceed the application s average output current. The diode s maximum reverse voltage must exceed the application s output voltage. A 4.5A diode is suffi cient for most designs. For dimming applications, be aware of the reverse leakage current of the diode. Lower leakage current will drain the output capacitor less, allowing for higher dimming range. The companies below offer Schottky diodes with high voltage and current ratings. Table 1. Suggested Inductors MANUFACTURER PART NUMBER IDC (A) INDUCTANCE (µh) MAX DCR (mω) L W H (mm) MANUFACTURER CDRH14R-1NC CDRH13RNP-4R7NC-B Sumida CDRH124R-1MC CDRH14R-5R2NC FDV63-4R7M Toko UP4B Cooper Table 2. Ceramic Capacitor Manufacturers MANUFACTURER PHONE NUMBER WEB Taiyo Yuden (48) AVX (83) Murata (714) Table 3. Suggested Diodes MANUFACTURER PART NUMBER MAX CURRENT (A) MAX REVERSE VOLTAGE WEB UPS Microsemi 1 B52C B53C B34A B54C PDS Diodes, Inc.

11 APPLICATIO S I FOR ATIO U W U U Shutdown and Programming Undervoltage Lockout The have an accurate 1.4V shutdown threshold at the S H D N pin. This threshold can be used in conjunction with a resistor divider from the system input supply to define an accurate undervoltage lockout (UVLO) threshold for the system (Figure 2). S H D N pin current hysteresis allows programming of hysteresis voltage for this UVLO threshold. Just before part turn on, 1µA fl ows into the S H D N pin. After part turn on, µa flows from the S H D N pin. Calculation of the on/off thresholds for a system input supply using the S H D N pin can be made as follows: V SUPPLY OFF = 1.4 [1 + R1/R2)] V SUPPLY ON = V SUPPLY OFF + (1µA R1) An open drain transistor can be added to the resistor divider network at the S H D N pin to independently control the turn off of the. Programming Switching Frequency The switching frequency is programmed using an external resistor (R T ) connected between the R T pin and ground. The internal free-running oscillator is programmable between 2kHz and 2.25MHz. Table 4 shows the typical R T values required for a range of switching frequencies. Selecting the optimum switching frequency depends on several factors. Inductor size is reduced with higher frequency but effi ciency drops due to higher switching losses. In addition, some applications require very high duty cycles to drive a large number of LEDs from a low supply. Low switching frequency allows a greater operational duty cycle and hence a greater number of LEDs to be driven. In each case the switching frequency can be tailored to provide the optimum solution. When programming the switching frequency the total power losses within the IC should be considered. See Thermal Calculations in the Applications Information section. V SUPPLY 1 T A = 25 C OFF ON R1 R2 11 SHDN 1.4V 1µA + SWITCHING FREQUENCY (khz) F2 Figure 2. Programming Undervoltage Lockout (UVLO) with Hysteresis R T (kω) 3478 F3 With the S H D N pin connected directly to the pin, an internal undervoltage lockout threshold exists for the pin (2.8V max). This prevents the converter from operating in an erratic mode when supply voltage is too low. The provide a soft-start function when recovering from such faults as S H D N <1.4V and/or <2.8V. See details in the Applications Information section Soft-Start. Figure 3. Switching Frequency vs R T Resistor Value Table 4. Switching Frequencies vs R T Values SWITCHING FREQUENCY (MHz) R T (kω)

12 APPLICATIO S I FOR Programming Maximum LED current Maximum LED current can be programmed using the CTRL1 pin with tied to the V REF pin (see Figures 4 and 5). The maximum allowed LED current is defined as: (LT3478-1) Max LED Current = Min(CTRL1, 1.5) Amps ( LT3478) Max LED Current = 1. Min( CTRL115,. ) Amps 12 LED CURRENT (ma) R S ENSE LED current vs CTRL1 is linear for approximately.1v < CTRL1 <.95V For maximum possible LED current, connect CTRL1 and to the V REF pin. T A = 25 C = V REF (FOR LT3478 SCALE BY.1Ω/R SENSE ) LT CTRL1 (V) ATIO U W U U V REF 3478 F4 Figure 5. Programming LED Current 1.4 Figure 4. LED Current vs CTRL1 Voltage R2 R1 1 V REF V OUT CTRL1 LED 3478 F5 (LT3478) R SENSE Programming LED Current Derating vs Temperature A useful feature of the is the ability to program a derating curve for maximum LED current versus temperature. LED data sheets provide curves of maximum allowed LED current versus temperature to warn against exceeding this current limit and damaging the LED (Figure 6). I f FORWARD CURRENT (ma) Luxeon V (Maximum) and LT (Programmed) Current Derating Curves vs Temperature LUXEON V EMITTER CURRENT DERATING 6 CURVE 5 4 EXAMPLE LT PROGRAMMED LED 2 CURRENT DERATING CURVE T A AMBIENT TEMPERATURE ( C) LUXEON V EMITTER (GREEN, CYAN, BLUE, ROYAL BLUE) θ JA = 2 C/W 3478 F6 Figure 6. LED Current Derating Curve vs Ambient Temperature Without the ability to back off LED current as temperature increases, many LED drivers are limited to driving the LED(s) at only 5% or less of their maximum rated currents. This limitation requires more LEDs to obtain the intended brightness for the application. The allow the output LED(s) to be programmed for maximum allowable current while still protecting the LED(s) from excessive currents at high temperature. This is achieved by programming a voltage at the pin with a negative temperature coeffi cient using a resistor divider with temperature dependent resistance (Figures 7 and 8). voltage is programmed higher than CTRL1 voltage. This allows initial LED current to be defi ned by CTRL1. As temperature increases, voltage will fall below CTRL1 voltage causing LED currents to be controlled by pin voltage. The choice of resistor ratios and use of temperature dependent resistance in the divider for the pin will defi ne the LED current curve breakpoint and slope versus temperature (Figure 8). A variety of resistor networks and NTC resistors with different temperature coeffi cients can be used for programming

13 APPLICATIO S I FOR R2 R1 R4 R V REF CTRL1 R Y R NTC R NTC R X R NTC R NTC A B OPTION A TO D C ATIO U W U U to achieve the desired curve vs temperature. The current derating curve shown in Figure 6 uses the resistor network shown in option C of Figure 7. D R Y 3478 F7 Figure 7. Programming LED Current Derating Curve vs Temperature (R NTC Located on LEDs PCB) CTRL1, PIN VOLTAGES (mv) CTRL LED CURRENT = MINIMUM 1 OF CTRL1, R3 = OPTION C T A AMBIENT TEMPERATURE ( C) 3478 F8 Figure 8. CTRL1, 2 Programmed Voltages vs Temperature Table 5 shows a list of manufacturers/distributors of NTC resistors. There are several other manufacturers available and the chosen supplier should be contacted for more detailed information. To use an NTC resistor to indicate LED temperature it is only effective if the resistor is connected as close as possible to the LED(s). LED derating curves shown by manufacturers are listed for ambient temperature. The NTC resistor should be submitted to the same ambient temperature as the LED(s). Since the temperature dependency of an NTC resistor can be nonlinear over a wide range of temperatures it is important R X to obtain a resistor s exact values over temperature from the manufacturer. Hand calculations of voltage can then be performed at each given temperature and the resulting curve plotted versus temperature. Several iterations of resistor value calculations may be required to achieve the desired breakpoint and slope of the LED current derating curve. Table 5. NTC Resistor Manufacturers/Distributors MANUFACTURER Murata Electronics North America TDK Corporation Digi-key If calculation of voltage at various temperatures gives a downward slope that is too strong, alternative resistor networks can be chosen (B, C, D in Figure 7) which use temperature independent resistance to reduce the effects of the NTC resistor over temperature. Murata Electronics provides a selection of NTC resistors with complete data over a wide range of temperatures. In addition, a software tool is available which allows the user to select from different resistor networks and NTC resistor values and then simulate the exact output voltage curve ( behavior) over temperature. Referred to as the Murata Chip NTC Thermistor Output Voltage Simulator, users can log onto and download the software followed by instructions for creating an output voltage V OUT () from a specifi ed V CC supply (V REF ). At any time during selection of circuit parameters the user can access data on the chosen NTC resistor by clicking on a link to the Murata catalog. The following example uses hand calculations to derive the resistor values required for CTRL1 and pin voltages to achieve a given LED current derating curve. The resistor values obtained using the Murata simulation tool are also provided and were used to create the derating curve shown in Figure 6. The simulation tool illustrates the non-linear nature of the NTC resistor temperature coeffi cient at temperatures exceeding 5 C ambient. In addition, the resistor divider technique using an NTC resistor to derive voltage inherently has a fl attening characteristic (reduced downward slope) at higher temperatures. To avoid LED current exceeding a maximum 13

14 APPLICATIO S I FOR 14 ATIO U W U U allowed level at higher temperatures, the voltage curve may require a greater downward slope between 25 C and 5 C to compensate for that loss of slope at higher temperatures. Example: Calculate the resistor values required for generating CTRL1 and from V REF based on the following requirements: (a) I LED = 7mA at 25 C (b) I LED derating curve breakpoint occurs at 25 C (c) I LED derating curve has a slope of 2mA/25 C between 25 C and 5 C ambient temperature Step1: Choose CTRL1 = 7mV for I LED = 7mA CTRL1 = V REF /(1 + R2/R1) R2 = R1 [(V REF /CTRL1) 1] For V REF = 1.24V and choosing R1 = 22.1k, R2 = 22.1k [(1.24/.7) 1] R2 = 17k (choose 16.9k) CTRL1 = 1.24/(1 + (16.9/22.1)) CTRL1 = 73mV (I LED = 73mA) Step 2: Choose resistor network option A (Figure 7) and = CTRL1 for 25 C breakpoint start with R4 = R2 = 16.9k, R NTC = 22k (closest value available) = 71mV (I LED = Min(CTRL1, ) 1A = 71mA) Step 3: Calculate slope between 25 C and 5 C (T) = 1.24/(1 + R4/R NTC (T)) at T = T O = 25 C, = 71mV at T = 5 C, R NTC (T) = R NTC (T O ).e x, x = B [(1/(T + 273) 1/298)] (B = B-constant; linear over the 25 C to 5 C temperature range) For R NTC B-constant = 395 and T = 5 C x = 395 [(1/323) 1/298] = 1.26 R NTC (5 C) = R NTC (25 C).e 1.26 R NTC (5 C) = 22k.358 R NTC (5 C) = 7.9k (5 C) = 1.24/( /7.9) = 395mV slope (25 C to 5 C) = [(5 C) (25 C)]/25 C = (395 71)/25 = 36mV/25 C I LED slope = 36mA/25 C The required I LED slope is 2mA/25 C. To reduce the slope of versus temperature it is easier to keep the exact same NTC resistor value and B-constant (there are limited choices) and simply adjust R4 and the type of resistor network used for the pin. By changing the resistor network to option C it is possible to place a temperature independent resistor in series with R NTC to reduce the effects of R NTC on the pin voltage over temperature. Step 4: Calculate the resistor value required for R Y in resistor network option (c) (Figure 7) to provide an I LED slope of 2mA/25 C between 25 C and 5 C ambient temperature. (25 C) =.7V = 1.24/(1 + (R4/(R NTC (25 C)+ R Y )) R4 =.77 (R NTC (25 C) + R Y ) (a) for 2mA/25 C slope (5 C) =.7.2 =.5 (5 C) =.5V = 1.24/(1 + (R4/(R NTC + R Y )) R4 = 1.48 (R NTC (5 C) + R Y ) (b) Equating (a) = (b) and knowing R NTC (25 C) = 22k and R NTC (5 C) = 7.9k gives,.77 (22k + R Y ) = 1.48 (7.9k + R Y ) 17k +.77 R Y = 11.7 k R Y R Y = (17k 11.7k)/( ) R Y = 7.5k

15 APPLICATIO S I FOR ATIO U W U U The value for R4 can now be solved using equation (a) where, R4 =.77 (R NTC (25 C) + R Y ) =.77 (22k + 7.5k) R4 = 22.7k (choose 22.6k) I LED slope can now be calculated from, I LED slope = [(5 C) (25 C)]/25 C where (5 C) = 1.24/( /( )) = 53mV and (25 C) = 1.24/( /( )) = 699mV giving I LED slope (from 25 C to 5 C) = 53mV 699mV/25 C = 196mV/25 C => I LED slope = 196mA/25 C Using the Murata simulation tool for the resistor network and values in the above example shows a voltage curve that flattens out as temperatures approach 1 C ambient. The final resistor network chosen for the derating curve in Figure 6 used option C network with R4 = 19.3k, R NTC = 22k (NCP15XW223JSRC) and R Y = 3.1k. Although the downward slope is greater than 2mA/25 C initially, the slope is required to avoid exceeding maximum allowed LED currents at high ambient temperatures (see Figure 6). Dimming Many LED applications require an accurate control of the brightness of the LED(s). In addition, being able to maintain a constant color over the entire dimming range can be just as critical. For constant color LED dimming, the provide a pin and special internal circuitry to allow up to a 3:1 wide dimming range. With an N-channel MOSFET connected between the LED(s) and ground and a signal connected to the gate of the MOSFET and the pin (Figure 9), it is possible to control the brightness of the LED(s) based on signal duty cycle only. This form of dimming is superior to dimming control using an analog input voltage (reducing CTRL1 voltage) because it allows constant color to be maintained during dimming. The maximum current for the output LED(s) is programmed for a given brightness/color and chopped over a duty cycle range (Figure 1) from 1% to as low as.33%. SHDN V REF OVPSET V S L SW LT3478/ LT V C V OUT Figure 9. Dimming Control Using the INDUCTOR CURRENT LED CURRENT CTRL1 R T LED DIMMING CONTROL (LT3478) R SENSE 3478 F9 Figure 1. Dimming Waveforms Using the D2 T TON (= 1/f ) MAX I LED D1 C OUT 3478 F1 Some general guidelines for LED Current Dimming using the pin (see Figure 1): (1) Dimming Ratio (PDR) = 1/( duty cycle) = 1/(TON f ) (2) Lower f allows higher Dimming Ratios (use minimum f = 1Hz to avoid visible fl icker and to maximize PDR) (3) Higher f OSC value improves PDR (allows lower TON ) but will reduce effi ciency and increase internal heating. In general, minimum operational TON = 3 (1/f OSC ). (4) Lower inductor value improves PDR 15

16 APPLICATIO S I FOR ATIO U W U U (5) Higher output capacitor value improves PDR (6) Choose the schottky diode (D2, Figure 9) for minimum reverse leakage See Typical Performance Characteristics graph LED Current vs Duty Cycle. Soft-Start To limit inrush current and output voltage overshoot during startup/recovery from a fault condition, the LT3478/ LT provide a soft-start pin SS. The SS pin is used to program switch current ramp up timing using a capacitor to ground. The monitor system parameters for the following faults: <2.8V, S H D N <1.4, inductor current >6A and boosted output voltage >OVP. On detection of any of these faults, the stop switching immediately and a soft-start latch is set causing the SS pin to be discharged (see Timing Diagram for the SS pin in Figure 11). When all faults no longer exist and the SS pin has been discharged to at least.25v, the soft-start latch is reset and an internal 12µA supply charges the SS pin. A gradual ramp up of SS pin voltage is equivalent to a ramp up of switch current limit until SS exceeds V C. The ramp rate of the SS pin is given by: ΔV SS /Δt = 12µA/C SS SW SS FAULTS TRIGGERING SOFT-START LATCH WITH SW TURNED OFF IMMEDIATELY: < 2.8V OR SHDN < 1.4V OR V OUT > OVP OR I (INDUCTOR) > 6A SOFT-START LATCH SET:.65V (ACTIVE THRESHOLD).25V (RESET THRESHOLD).15V SOFT-START LATCH RESET: SS <.25V AND > 2.8V AND SHDN > 1.4V AND V OUT < OVP AND I (INDUCTOR) < 6A 3478 F11 Figure 11. LT3478 Fault Detection and SS Pin Timing Diagram To limit inductor current overshoot to <.5A when SS charges past the V C level required for loop control, the C SS capacitor should be chosen using the following formula: C SS(MIN) = C C (7.35.6(I LED V OUT /V S )) Example: V S = 8V, V OUT = 16V, I LED = 1.5A, C C =.1µF, C SS(MIN) =.1µF (7.35.6(1.5 16/8)) =.612µF (choose.68µf). High Inductor Current Inrush Protection The provide an integrated resistor between the V S and L pins to monitor inductor current (Figure 1). During startup or hotplugging of the inductor supply, it is possible for inductor currents to exceed the maximum switch current limit. When inductor current exceeds 6A, the protect the internal power switch by turning it off and triggering a soft-start latch. This protection prevents the switch from repetitively turning on during excessive inductor currents by delaying switching until the fault has been removed. To defeat inductor current sensing the inductor supply should be connected to the L pin and the V S pin left open. See details in the Applications Information section Soft-Start. LED Open Circuit Protection and Maximum Dimming Ratios The LED drivers provide optimum protection from open LED faults by clamping the converter output to a programmable overvoltage protection level (OVP). In addition, the programmable OVP feature draws zero current from the output during = to allow higher dimming ratios. This provides an advantage over other LED driver applications which connect a resistor divider directly from V OUT. An open LED fault occurs when the connection to the LED(s) becomes broken or the LED(s) fails open. For an LED driver using a step-up switching regulator, an open circuit LED fault can cause the converter output to exceed the voltage capabilities of the regulator s power switch, causing permanent damage. When V OUT exceeds OVP, the 16

17 APPLICATIO S I FOR ATIO U W U U immediately stop switching, a soft-start latch is set and the SS pin is discharged. The SS latch can only be reset when V OUT falls below OVP and the SS pin has been discharged below.25v (Figure 11). If the LED(s) simply go open circuit and are reconnected, however, the OVP used to protect the switch might be too high for the reconnected LED(s). The therefore allow OVP to be programmable to protect both the LED driver switch and the LED(s). (The minimum allowable OVP for normal operation for a given LED string depends on the number of LEDs and their maximum forward voltage ratings.) OVP is programmed using the OVPSET pin (front page), given by, OVP = (OVPSET 41)V where the programmable range for the OVPSET pin is.3v to 1V resulting in an OVP range of 12.3V to 41V. The OVPSET pin can be programmed with a single resistor by tapping off of the resistor divider from V REF used to program CTRL1. If both CTRL1 and are connected directly to V REF (maximum LED current setting) then OVP- SET requires a simple 2 resistor divider from V REF. Thermal Calculations To maximize output power capability in an application without exceeding the 125 C maximum operational junction temperature, it is useful to be able to calculate power dissipation within the IC. The power dissipation within the IC comes from four main sources: switch DC loss, switch AC loss, Inductor and LED current sensing and input quiescent current. These formulas assume a boost converter architecture, continuous mode operation and no dimming. (1) Switch DC loss = P SW(DC) = (R SW I 2 L(AVE) D) R SW = switch resistance =.7Ω (at T J = 125 C) I L(AVE) = P OUT /(η V S ) P OUT = V OUT I LED η = converter efficiency = P OUT /(P OUT + P LOSS ) V S = inductor supply input D = switch duty cycle = (V OUT + V F V S )/(V OUT + V F V SAT ) V F = forward voltage drop of external Schottky diode V SAT = I L(AVE) R SW (2) Switch AC loss = P SW(AC) = t EFF (1/2)I L(AVE) (V OUT + V F )(F OSC ) t EFF = effective switch current and switch V CE voltage overlap time during turn on and turn off = 2 (t ISW + t VSW ) t ISW = I SWITCH rise/fall time = I L(AVE) 2ns t VSW = SW fall/rise time = (V OUT + V F ).7ns f OSC = switching frequency (3) Current sensing loss = P SENSE = P SENSE(IL) + P SENSE(ILED) P SENSE(IL) = I 2 L(AVE) 9.5mΩ P SENSE(ILED) = I 2 LED.1Ω (4) Input quiescent loss = P Q = I Q where I Q = (6.2mA + (1mA D)) Example (Using LT3478-1): For = V S = 8V, I LED = 7mA, V OUT = 24.5V (7 LEDs), V F =.5V and f OSC =.2Mhz, η =.89 (initial assumption) I L(AVE) = (24.5.7)/(.89 8) = 2.41A D = ( )/( ) =.684 T EFF = 2 ((2.41 2)ns + ( ).7)ns = 45ns Total Power Dissipation: P IC = P SW(DC) + P SW(AC) + P SENSE + P Q P SW(DC) =.7 (2.41) =.278W P SW(AC) = 45ns MHz =.271W P SENSE = ((2.41) 2.95) + ((.7) 2.1) =.14W P Q = 8 (6.2mA + (1mA.684)) =.597W P IC = = 1.25W 17

18 APPLICATIO S I FOR Local heating from the nearby inductor and Schottky diode will also add to the final junction temperature of the IC. Based on empirical measurements, the effect of diode and inductor heating on the LT junction temperature can be approximated as: ΔT J (LT3478-1) = 5 C/W (P DIODE + P INDUCTOR ) P DIODE = (1 D) V F I L(AVE) 1 D =.316 V F =.5V I L(AVE) = 2.41 P DIODE = =.381W P INDUCTOR = I 2 L(AVE) DCR DCR = inductor DC resistance (assume.5ω) P INDUCTOR = (2.41) 2.5 =.29W The use a thermally enhanced FE package. With proper soldering to the Exposed Pad on the underside of the package combined with a full copper plane underneath the device, thermal resistance (θ JA ) will be about 35 C/W. For an ambient temperature of T A = 7 C, the junction temperature of the LT for the example application described above, can be calculated as: T J (LT3478-1) = T A + θ JA (P TOT ) + 5(P DIODE + P INDUCTOR ) = (1.25) + 5(.671) = = 118 C In the above example, efficiency was initially assumed to be η =.89. A lower efficiency (η) for the converter will increase I L(AVE) and hence increase the calculated value for T J. η can be calculated as: η = P OUT /(P OUT + P LOSS ) P OUT = V OUT I LED = 17.15W P LOSS (estimated) = P IC + P DIODE + P INDUCTOR = 1.92W η = 17.15/( ) =.9 ATIO U W U U If an application is built, the inductor current can be measured and a new value for junction temperature estimated. Ideally a thermal measurement should be made to achieve the greatest accuracy for T J. Note: The junction temperature of the IC can be reduced if a lower supply is available separate from the inductor supply V S. In the above example, driving from an available 3V source (instead of V S = 8V) reduces input quiescent losses in item(4) from.597w to.224w, resulting in a reduction of T J from 118 C to 15 C. Layout Considerations As with all switching regulators, careful attention must be given to PCB layout and component placement to achieve optimal thermal,electrical and noise performance (Figure 12). The exposed pad of the (Pin 17) is the only GND connection for the IC. The exposed pad should be soldered to a continuous copper ground plane underneath the device to reduce die temperature and maximize the power capability of the IC. The ground path for the R T resistor and V C capacitor should be taken from nearby the analog ground connection to the exposed pad (near Pin 9) separate from the power ground connection to the exposed pad (near Pin 16). The bypass capacitor for should be placed as close as possible to the pin and the analog ground connection. SW pin voltage rise and fall times are designed to be as short as possible for maximum efficiency. To reduce the effects of both radiated and conducted noise, the area of the SW trace should be kept as small as possible. Use a ground plane under the switching regulator to minimize interplane coupling. The schottky diode and output capacitor should be placed as close as possible to the SW node to minimize this high frequency switching path. To minimize LED current sensing errors for the LT3478, the terminals of the external sense resistor R SENSE should be tracked to the V OUT and LED pins separate from any high current paths. 18

19 APPLICATIO S I FOR ATIO U W U U C VS C VIN V S SCHOTTKY DIODE V OUT (CONNECT MULTIPLE GROUND PLANES THROUGH VIAS UNDERNEATH THE IC) OUTPUT CAPACITOR GND SOLDER THE EXPOSED PAD (PIN 17) TO THE ENTIRE COPPER GROUND PLANE UNDERNEATH THE DEVICE SW SW SW 1 2 POWER GND SS R T C SS R T INDUCTOR V S L R R L 5 12 CTRL1 R R SENSE (LT3478 ONLY) R R V OUT LED OVPSET EXPOSED PAD PIN 17 ANALOG GND C BYPASS CAP 11 SHDN 1 V REF 9 V C C F R C R C C 3252 F8 Figure 12. Recommended Layout for (Boost Confi guration) TYPICAL APPLICATIO S U 15W, 6 LEDs at 7mA, Boost LED Driver L1 1µH D1 8V TO 16V C1 4.7µF 25V V S L SW SHDN OUT C2 1µF 25V LT Dimming Waveforms R1 45.3k R4 54.9k R2 13k L1: CDRH14R-1NC D1: PDS56 Q1: Si2318DS LEDs: LUXEON III (WHITE) V REF OVPSET CTRL1 LT LED SS V C R T C SS 1µF C C.1µF f OSC = 5kHz 7mA R T 69.8k 5V/DIV f = 1Hz INDUCTOR CURRENT 1A/DIV I LED.5A/DIV 2µs/DIV DIMMING RATIO = 1:1 (SEE EFFICIENCY ON PAGE 1) 3478 TA2b 3.3V V 1Hz DIMMING RATIO = 1:1 R3 1k Q TA2a 19

20 TYPICAL APPLICATIO S U 17W, 15 LEDs at 35mA, Boost LED Driver plus LT33 V S 8V TO 14V 3.3V C1 4.7µF 16V C3 3.3µF 1V L1: CDRH14R-5R2 D1: PDS56 LEDs: LUXEON I (WHITE) R1 24k R2 1k 3.3V V 1Hz V S L SW SHDN V REF OUT CTRL1 LT LED OVPSET SS V C R T C SS 1µF DIMMING RATIO = 3:1 L1 5.2µH V C C C.1µF f OSC = 1MHz V OUT D1 1.5A R T 31.6k V OUT C2 3.3µF 25V LED1 LED2 LED3 V MAX LT33 OT1 OT2 GND V EE V C EFFICIENCY (%) Effi ciency vs Input V S = 3.3V I LED = 35mA f OSC = 1MHz DUTY CYCLE = 1% 15 LEDs (5 SERIES x 3 CHANNELS) LUXEON I (WHITE) V S (V) 3478 TA3b 3478 TA3a 16W, 12 LEDs at 35mA, Buck-Boost Mode LED Driver plus LT33 V S 12V TO 16V 5V C1 4.7µF 25V C3 3.3µF 1V L1: CDRH15R-8R2 D1: PDS56 D2: 7.5V ZENER LEDs: LUXEON I (WHITE) R1 24k R2 1k 3.3V V 1Hz V S L SW SHDN V REF CTRL1 OVPSET LT OUT LED SS V C R T C SS 1µF DIMMING RATIO = 2:1 C4 1µF L1 8.2µH V C C C.1µF D1 1.5A R T 69.8k V OUT C2 1µF 5V f OSC = 5kHz D2 V OUT LED1 LED2 LED3 V MAX LT33 OT1 OT2 GND V EE V C EFFICIENCY (%) Effi ciency vs Input V S = 5V I LED = 35mA f OSC = 5kHz DUTY CYCLE = 1% LEDs (4 SERIES x 3 CHANNELS) 5 LUXEON I (WHITE) V S (V) 3478 TA4b 3478 TA4a 2

21 TYPICAL APPLICATIO S U 4W, 1 LED at 1A, Buck-Boost Mode LED Driver 3.8V TO 6.5V NiMH 4 ON OFF C1 1µF 1V L1 6.8µH V S L SW SHDN OUT D1 C2 4.7µF 16V 8 75 Effi ciency vs I LED = 1A f OSC = 5kHz DUTY CYCLE = 1% 1kHz R1 1k R2 L1: CDRH15R-6R8 34k D1: B32 Q1: Si232ADS Q2: Si2315BDS LED: LUXEON III (WHITE) 3.3V V V REF CTRL1 OVPSET LT LED SS V C R T C SS 1µF DIMMING RATIO = 2:1 C C.1µF f OSC = 5kHz R T 69.8k 1A R3 1k R4 51Ω R5 51Ω Q1 Q2 EFFICIENCY (%) SINGLE LED 5 LUXEON III (WHITE) (V) 3478 TA6b 3478 TA6a 21

22 TYPICAL APPLICATIO S U 24W, 4 LEDs at 1.5A, Buck Mode LED Driver P 32V C1 3.3µF 5V R SENSE.68Ω 1.5A 4 LEDs TYPICAL EFFICIENCY = 9% FOR CONDITIONS/COMPONENTS SHOWN ( DUTY CYCLE = 1%, T A =25 C) Q2 R4 365Ω C3 1µF 25V L1 1µH R5 51Ω 3.3V L1: CDRH15R-1 D1: PDS56 Q1: 2N72 Q2: Si2319DS LEDs: LXK2 (WHITE) C2 4.7µF 1V R1 24k R2 1k V S L OUT LED SW SHDN V REF CTRL1 OVPSET C SS 1µF LT3478 SS V C R T C C.1µF DIMMING RATIO = 3:1 3.3V V R T 69.8k D1 R3 1k 1Hz Q1 f OSC = 5kHz 3478 TA7a 22

23 PACKAGE DESCRIPTIO U FE Package 16-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # ) Exposed Pad Variation BC 3.58 (.141) * ( ) 3.58 (.141) ± ± (.116) SEE NOTE 4.45 ± ± (.116) 6.4 (.252) BSC.65 BSC RECOMMENDED SOLDER PAD LAYOUT * ( ).25 REF (.433) MAX.9.2 (.35.79).5.75 (.2.3) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE.65 (.256) BSC ( ) TYP 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED.15mm (.6") PER SIDE.5.15 (.2.6) FE16 (BC) TSSOP 24 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23

24 TYPICAL APPLICATIO U 6W, 6 LEDs at 25mA, Boost LED Driver V S 8V TO 16V 3.3V C3 3.3µF 1V L1: CDRH6D28 D1: ZLLS1 Q1: Si2318DS LEDs: LUXEON I (WHITE) 3.3V C1 4.7µF 25V R1 8.25k R2 1k L1 1µH V S L SW SHDN V REF OUT CTRL1 LT3478 LED OVPSET SS V C R T C SS 1µF C C.1µF f OSC = 2MHz D1 R T 1k R SENSE.42Ω 25mA C2 3.3µF 25V EFFICIENCY (%) Effi ciency vs Input V S = 3.3V I LED = 25mA f OSC = 2MHz DUTY CYCLE = 1% 1 6 LEDs = LUXEON I (WHITE) V S (V) 3478 TA5b V 1Hz DIMMING RATIO = 1:1 R3 1k Q TA5a RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1618 Constant Current, 1.4MHz, 1.5A Boost Converter with Analog/ : 5V to 18V, V OUT(MAX) = 36V, I SD <1µA, MS1 Package Dimming LT33 Three Channel LED Ballaster with 3,:1 True Color Dimming : 3V to 48V, I SD <5µA, MSOP1 Package LT V, 1A (I LED ), 2MHz,Step-Down LED Driver with 4:1 True Color : 4V to 36V, V OUT(MAX) = 13.5V, I SD <1µA, TSSOP16E Package Dimming LT3475 Dual 1.5A(I LED ), 36V, 2MHz,Step-Down LED Driver 3,:1 True : 4V to 36V, V OUT(MAX) = 13.5V, I SD <1µA, TSSOP2E Package Color Dimming LT3476 Quad Output 1.5A, 2MHz High Current LED Driver with 1,:1 True Color Dimming : 2.8V to 16V, V OUT(MAX) = 36V, I SD <1µA, 5mm 7mm QFN Package LT3477 LT3479 LT V, 3A, 3.5MHz Boost, Buck-Boost, Buck LED Driver with Analog/ Dimming 3A, 3.5MHz Full Featured DC/DC Converter with Soft-Start and Inrush Current Protection and Analog/ Dimming Dual 1.3A, 2MHz High Current LED Driver with 1,:1 True Color Dimming : 2.5V to 25V, V OUT(MAX) = 4V, I SD <1µA, QFN, TSSOP2E Packages : 2.5V to 24V, V OUT(MAX) = 4V, I SD <1µA, 4mm 3mm DFN, TSSOP16E Packages : 2.5V to 24V, V OUT(MAX) = 36V, I SD <1µA, 5mm 3mm DFN, TSSOP16E Packages LTC3783 High Current LED Controller with 3,:1 True Color Dimming : 3V to 36V, V OUT(MAX) = Ext FET, I SD <2µA, 5mm 4mm DFN, TSSOP16E Packages 24 LT 17 PRINTED IN USA Linear Technology Corporation 163 McCarthy Blvd., Milpitas, CA (48) FAX: (48) LINEAR TECHNOLOGY CORPORATION 27