Power Cardboard Modelling and Cable Model

Transcription

1 3D High Frequency Modelling of Motor Converter and Cables in Propulsion Systems Stephane Yannick Njiomouo Master Thesis Stockholm, Sweden 2014 XR-EE-ETK 2014:012

2 Abstract The use of the power converters in railway traction systems introduces high frequency electromagnetic interference (EMI) in the propulsion system, which causes electromagnetic compatibility (EMC) problems. These high frequency phenomena come from fast variations of current and voltage during the switching operations in the power converter. The high frequency currents generate Electromagnetic (EM) disturbances that could distort the smooth functionality of the electrical drive system. In fact, power and audio frequency emissions could disturb track signaling and the control systems, while high frequency currents injected into cable screens could damage the cables. In order to ensure compatibility to conducted and radiated EMC requirements, and related infrastructure signaling specifications, it is necessary to perform 3D modelling of the drive system to predict the EM emission during the design phase of the propulsion system. CST, an electromagnetic analysis tool, is used to create the 3D model of the converter module and the cables. The model allows for the inclusion of the parasitic characteristics of the IGBTs, the bus-bars, and the motor cables. Influence of different grounding schemes is analyzed. The model predicts the EM field distribution at points inside the converter module and in the vicinity. i

3 3D Högfrekvens modellering av motor omvandlare och kablar i Framdrivnings Systems Användningen av kraftomvandlare i järnvägstraktionssystem introducerar högfrekvens elektromagnetisk interferens (EMI) i framdrivningssystemet, vilket orsakar elektromagnetiska kompatibilitetsproblem (EMC). Dessa högfrekvensfenomen orsakas av snabba variationer i ström och spänning under omkopplingsoperationer i kraftomvandlare. Högfrekvensströmmarna alstrar elektromagnetiska (EM) störningar, som kan påverka funktionaliteten hos det elektriska drivsystemet. Störningar vid kraft- och ljudfrekvenser kan påverka signal- och kontrollsystemen, medan högfrekventa strömmar injiceras i kabelskärmar kan skada kablarna. För att säkerställa kompatibiliteten mellan EMC-kraven, vad gäller ledningsbundna och utsända störningar, och specifikationerna för signalsystemets infrastruktur är det nödvändigt att utföra 3D-modellering av drivsystemet, för att redan under designfasen av framdrivningssystemet kunna förutsäga de elektromagnetiska störningarna. CST, som är ett elektromagnetiskt analysverktyg, används för att skapa 3D-modellen av omriktarmodulen och kablarna. Modellen gör det möjligt att ta med de parasitiska egenskaperna hos IGBT, ledningsmoduler och motorkablar. Inverkan av olika jordningssystemen analyseras. Modellen förutsäger det elektromagnetiska fältet vid olika punkter inuti omriktarmodulen och i dess närhet. ii

4 Acknowledgements I would like to express my appreciation to my masters thesis supervisor: Dr Mathias Enohnyaket who is always available to help me. My gratitude also goes to my college in the company and in particular the EMC group members who eased my integration in Bombardier. I would like to thank also my kth examiner Prof Rajeev Thottappillil and Prof Michele Tartaglia at Polytechnic of Turin for accepting me as master thesis student. Always at kth, a particular thank for my ERASMUS supervisor Magnus Lindqvist for be always available for me. Thank to my friends Ensa, Valerie, Giuseppe, Fabrice, TEK Boris and Nelly who assist me every time. I could not forget my Family with all sacrifices they made for me and to tell you again THANK YOU and I love you. iii

5 Contents Abstract Acknowledgements i iii Contents List of Figures List of Tables iv vi ix 1 Introduction Description Previous work Thesis Outline Cable modeling Introduction Ground plane effects Surface currents The magnetic field S-parameter S Screen current estimation Background Assumptions Simulation results Power Cable Description Simulation results Conclusion Converter Models Introduction Parasitic modeling of the IGBTs Parameter estimation Observations iv

6 Contents v Results Improvement of the damping IGBTs modeled as voltage controlled switches Description Results Conclusion Converter and Cable Coupling Description Voltage and current responses D field distribution EM field 2D plots Conclusion Discussion and Future Work Discussion Future Work Conclusion

7 List of Figures 2.1 The surface current distribution at 10 khz with common ground The surface current distribution at 10 khz with separated ground Zoom of the surface current close to port The surface current distribution with common ground at 200 MHz The surface current distribution at 200 MHz with separated ground Phase field distribution at 10 khz with common ground Magnetic field distribution at 10 khz with separated ground Magnetic field distribution at 200 MHz with common ground Magnetic field distribution at 200 MHz with separated ground Magnitude of S 11 in the frequency domain: in blue when a common ground is used and in green when separated grounds are used Magnitude of S 11 in the frequency domain: in blue when is used a common ground and in green when separated grounds are used Port impedance behaviour in the frequency domain: in blue when a common ground is used and in green when separated grounds are used Theoretical model used for building the equations [1] Screen current ratios computer by CST: I s /I c phase a in blue, I s /I c phase b in green, I s /I c phase c in red Screen current ratios measured Schematic of the reflection test in CST Port voltage in red, line close end voltage in green and far end voltage waveforms for Z L = 1MΩ Port voltage in red, line close end voltage in green and far end voltage waveforms for Z L = 1mΩ The port voltage signal (in blue) and the far end voltage signal (in green) Simplification of the model DC link excitation Load setting in AC side Simultion setting Wire copper connecting terminals of IGBT Lumped element connecting the IGBT terminals Excitation signal pattern vi

8 List of Figures vii 3.8 Schematic of the circuit. The < 110 > state of the inverter is simulated with 100Ω in star connection Voltage responses Current responses Setting of the system with the filter Schematic of the filter Voltage responses Current responses Motor converter module with the IGBT ports IGBT ports Voltage responses using imported IGBTs Voltage responses using VCSW Current responses using imported IGBTs Current responses using VCSW Voltage responses using VCSW with rise time equals 0.4 µs schematic of the system Cable setting Cables coupled with converter IGBT locations Grounding and EMC filter setting schematic of the system load Voltages DC link current and voltage D Electric field at 10 khz D Electric field at 500 MHz Cross section at the middle of the line at 10 khz Cross section at the middle of the line at 500 MHz D Magnetic field at 10 khz D Magnetic field at 500 MHz Probe location in the system Electric field in point A, close to the IGBT in V/m FFT Electric field in point A, close to the IGBT V/m Magnetic field in point A, close to the IGBT in A/m FFT Magnetic field in point A, close to the IGBT in A/m Electric field in point B, left to the MCM in V/m FFT Electric field in point B, left to the MCM in V/m Magnetic field in point B, left to the MCM in A/m FFT Magnetic in point B, field left to the MCM in A/m Electric field in point C, 1m from the cable middle in V/m FFT Electric field in point C, 1m from the cable middle V/m Magnetic field in point C, 1m from the cable middle A/m FFT Magnetic fieldin point C, 1m from the cable middle in A/m Electric field in point D, close to the cables in V/m

9 List of Figures viii 4.29 FFT Electric field in point D, close to the cables in V/m Magnetic field in point D, close to the cables A/m FFT Magnetic in point D, close to the cables in A/m

10 List of Tables 2.1 Cable parameters IGBTs model System parameters with filters Mean values of voltage and current response ix

11 Chapter 1 Introduction 1.1 Description Power converter and cables are very important components of the variable speed drives. The converter delivers switching waveforms of voltage and current. These fast voltage variations (dv/dt) inject into the cables current and voltage transients which are sources of electromagnetic interferences (EMI). Depending of the path, the power propagations, on the motor side, can create over-voltage at the cable terminals which can be a critical issue for the motor insulation material. The fast transients in the cables generate EM radiation which could disturb the smooth functionality of the control and other sensitive systems. In fact, current and voltage sensors installed on the converter module are usually victims of the EM disturbances from the converter and cable. These transients injected back to the grounding line could distort the the functionality of the track signaling systems. To prevent these issues and make sure that the drive system works under the EMI/EMC safety limits, it is necessary to have good and realistic models earlier in the design. CST EM modeling tool, is used to create 3D models of the converter module and the cable interconnections. From CAD model of the converter geometry is imported in CST microwave (MW) studio and the parasitic inductance and capacitance of the bus-bar interconnections are estimated. The IGBT parasitics are represented as lumped parameters. The cable connections are modeled as transmission lines in CST cable (CB) studio. The cable model is complete with the converter module to obtain a complete model of the drive system in CST design studio (DS). The model is excited with current or voltage 1

12 Introduction 2 source, and the current and voltage distribution in the structure are evaluated. From the current distribution, the EM fields are estimated within the converter and the vicinity. 1.2 Previous work In the previous masters thesis [2], the resonant behavior of the power converter including the IGBT parasitic are modeled. It characterizes the IGBTs with lumped elements. It makes the excitation from the DC link bus, observes the results on the AC side.the EM field values at different locations are measured. The software tool CST (Computer Simulation Technology) was used. This thesis is a continuation of the work done in [2] and the new contributions are the following: The off-state capacitance chance to the Connector-Emitter capacitance; The excitation is set between the DC link bus-bars; The dynamic IGBT switching characteristic is introduced; Cables and ground plane are coupled with the converter module in order to complete the drive system. 1.3 Thesis Outline The thesis is arranged as followed: Chapter 2 deals with cable modeling including ground plane effects, screen current estimation, s-parameter estimation and voltage reflections at terminals; Chapter 3 deals with modeling of the converter module, including the represented of the IGBTs, the input lumped filter elements and the converter modulation state; Chapter 4 presents the assembly of the converter module and the cable model with a given motor equivalent circuit representation. Predictions of

13 Introduction 3 the EM fields of the EM field distribution within the converter module and along the cables are represented. Chapter 5 concludes the thesis with some discussions and suggestions for the future work.

14 Chapter 2 Cable modeling 2.1 Introduction In the motor drive system, power is mostly transmitted between components through cables. The performance of the cable can be affected by different factors, for example the ground system, the presence of other cables or the termination nature. The cable deals with problems like heating and breakdown voltage in the insulation. This chapter presents investigations which allow to understand the mechanism of these phenomena. The analysis will focus on the study of the ground plane effect in section 2.2, the screen current behaviour in the frequency domain in section 2.3 and the propagation of the voltage using lossy cable in section Ground plane effects An issue when simulations are made is to make sure that the injected signal will follow the direction and the sense it is supposed to follow. For this aim, the nature and the setting of the ground plane must be studied very well. In this case for instance, a perfect electrical conductor is used as ground material. Two configurations have been considered: A common ground situation: the cable ends are both connected to the same ground through two ports with internal impedance equal to 0.35mΩ. 4

15 Chapter 2. Cable modeling 5 A separated ground situation: the cable ends are connected to different grounds. A 200 cm copper cable length is considered. After the excitations of both ports the surface currents and the magnetic fields are observed at 10 khz and 200 MHz Surface currents The figures 2.1 and 2.3 show the surface current distribution at 10 khz with common ground while figure 2.2 shows the distribution for separated ground respectively. Figures 2.4 and 2.5 show the surface currents at 200 MHz. Observations At 10 khz At low frequency the surface current is reduced using separated grounds. The maximum current amplitudes for 10kHz are 1.026A/m and A/m for the single ground and for the separated grounds respectively. At 200 MHz At high frequency the surface currents are almost the same. The maximum fields amplitudes for 200M Hz are A/m and A/m for single ground and separated ground respectively The magnetic field Using the settings presented in section 2.3 the EM field distribution could be observed. Figures 2.6 and 2.7 show the magnetic field distribution at 10 khz with common and separated ground respectively. Figures 2.8 and 2.9 show the magnetic field distribution at 200 MHz.

16 Chapter 2. Cable modeling 6 Figure 2.1: The surface current distribution at 10 khz with common ground Figure 2.2: The surface current distribution at 10 khz with separated ground. Figure 2.3: Zoom of the surface current close to port 1 Observations: At low frequency the magnetic field is reduced using separated ground. The maximum field amplitudes for 10 khz are A/m and A/m for single ground and separated ground respectively. At high frequency the magnetic fields are almost the same. The maximum field amplitudes for 200 MHz are A/m and A/m for single ground and separated ground respectively. Thus the ground configuration affects the surface current and the magnetic field only in low frequency. In high frequency, currents have almost the same value.

17 Chapter 2. Cable modeling 7 Figure 2.4: The surface current distribution with common ground at 200 MHz Figure 2.5: The surface current distribution at 200 MHz with separated ground S-parameter S11 The scattering parameter S 11 has been computed in the different ground configurations. The S 11 parameter can be derived by the following formula: Γ 0 = S 11 = Z load Z source Z load + Z source (2.1) If Z load = Z source (matchedload), S 11 = 0. Figure 2.10 shows the phase of S 11, figure 2.11 shows the magnitude of S 11 and figure 2.12 shows the port impedance in the frequency domain. As can be observed in figure 2.10, in low frequency the phases overlap each other and we have the same resonance frequency. Then we have some deviation in high frequency. Figure 2.11 shows that in low frequency, the power goes first through the ground since it presents lower impedance than the cable and the port in the case of common ground. The ground impedance is zero, so the S 11 = 1. In the case of distinct grounds, since the two ports have

18 Chapter 2. Cable modeling 8 Figure 2.6: Phase field distribution at 10 khz with common ground Figure 2.7: Magnetic field distribution at 10 khz with separated ground Figure 2.8: Magnetic field distribution at 200 MHz with common ground Figure 2.9: Magnetic field distribution at 200 MHz with separated ground the same impedance and the line impedance is very low, S 11 will be equal to 0. Figure 2.12 shows how much the equivalent port impedance is very high at the resonant frequency and is very low frequency (0.35 Ohm). 2.3 Screen current estimation Background High currents flowing in cable screens could cause heating issues and could even burn the cable in the worst case. In some applications there are requirements that limit the amount of the screen currents. Thus it is essential that the screen cables should be adequately well modelled in the design stage in order to prevent

19 Chapter 2. Cable modeling 9 Figure 2.10: Magnitude of S 11 in the frequency domain: in blue when a common ground is used and in green when separated grounds are used Figure 2.11: Magnitude of S 11 in the frequency domain: in blue when is used a common ground and in green when separated grounds are used

20 Chapter 2. Cable modeling 10 Figure 2.12: Port impedance behaviour in the frequency domain: in blue when a common ground is used and in green when separated grounds are used Elements Materials Diameter [mm] Thickness [mm] Conductor Aluminum Inner insulation PE Screen Copper Outer insulation PVC - 2 Table 2.1: Cable parameters high screen currents. The simulation of the three phase screen motor cables is presented. The characteristics of the cables are given in Table Assumptions The ground is a perfect conductor. The three-phase cables are modelled as shown in figure The distance between conductors is 6 cm.

21 Chapter 2. Cable modeling 11 A solid shield is considered for the calculations. Simulation range is 0 to 350 Hz to compare with the measurements performed in [3]. Three symmetric current sources, with amplitude 1, are used to feed the phase center wires. Symmetric current phases are 0, 2π/3, 2π/3. Figure 2.13 presents the screen cable model with the different current flowing in the system. Figure 2.13: Theoretical model used for building the equations [1] The following equations found in [1] link the phase currents and the screen currents. 0 = R s I a + jω(l a a I a + L a b I b + L a c I c ) + jω(l a ai a + L a bi b + L a ci c ) (2.2) 0 = R s I b + jω(l b a I a + L b b I b + L b c I c ) + jω(l b ai a + L b bi b + L b ci c ) (2.3) 0 = R s I c + jω(l c a I a + L c b I b + L c c I c ) + jω(l c ai a + L c bi b + L c ci c ) (2.4) Where the parameters are:

22 Chapter 2. Cable modeling 12 R s is Resistance per unit length of the screen L i i and L i i are Self inductance of the screen and mutual inductance screen center wire for the same phase L i j and L i j are mutual inductance screen to screen and screen to cable center wire between different phases These values can be calculated using close formulas but in this part the CST results will be used Simulation results According to the previous description, the model has been implemented in CST (Computer simulation technology) cable studio. The screen current over the phase current ratios has been computed for each phase and compared with the measured results. Figure 2.14 presents the simulated results whereas figure 2.15 presents the measured results. The measured results show some differences from the simulated ones. These differences might come out from the following reasons: Missing information about screen (braided shield characteristics); Screen is considered as solid shield in the model; The measured currents are generated by a converter which delivers the fundamental frequency and the harmonics. The probes measure the current RMS values, which include high frequency current components. High frequency current components will increase the measured screen currents; The ground plane in the model is different from the actual ground plane in the project.

23 Chapter 2. Cable modeling Frequency [Hz] Figure 2.14: Screen current ratios computer by CST: I s /I c phase a in blue, I s /I c phase b in green, I s /I c phase c in red. Figure 2.15: Screen current ratios measured

24 Chapter 2. Cable modeling Power Cable Description The voltage waveform delivered by the power converter is a switching waveform between the DC bus voltage polarities with rise and fall time. Reflection phenomena will occur at the cable ends according to the nature of the cable terminal at the motor ends. This can lead to over voltages which can deteriorate the motor insulation materials. As presented in [4] Two coefficients which allow to estimate the voltage at the terminal are: Reflection coefficient Γ L = (Z L Z 0 )/(Z L + Z 0 ) (2.5) Transmission coefficient β L = 1 + Γ L (2.6) Where Z L and Z 0 are the load and line characteristic impedance respectively. The terminal voltage will be: V = V + + V = V + + Γ L V + = β L V + (2.7) Where V + is the incident voltage and V is the reflected voltage. If Z L = (open load terminal), V=1; If Z L = 0 (short circuit load terminal), V=0; Simulation results A 10m length screen cable RG58 is used in this part. The port generates the impulse signal with unity amplitude and with rise time 11.3ns. The figure 2.16 shows the schematic of the system. The port impedance is equal to 50mΩ. Figure 2.16 and figure 2.17 show respectively the results when resistive load are1m Ω and 1mΩ. The following observations can be made: When the load is 1MΩ, over voltage phenomena are observed. In this case we almost reach two times the port voltage.

25 Chapter 2. Cable modeling 15 When the load is 1mΩ, the reflected voltage is very small too. The delay between the close end and the far end voltage depends on the cable length. Figure 2.16: Schematic of the reflection test in CST The travelling wave produces voltage oscillations at the ends of the line as can be seen in figure This figure presents the cable far end voltage and the port voltage when the far end of the line is connected to 1 MOhm resistor. The port signal has the maximum voltage equal to 700 V with rise time equal to 0.2 micro seconds. The damping of these oscillations depends on the cable terminal and the cable losses. The previous formula shows the relationship between the cable length and the frequency of the propagation wave. The frequency decreases when the cable length increases.

26 Chapter 2. Cable modeling 16 Figure 2.17: Port voltage in red, line close end voltage in green and far end voltage waveforms for Z L = 1MΩ Figure 2.18: Port voltage in red, line close end voltage in green and far end voltage waveforms for Z L = 1mΩ

27 Voltage [Volt] Chapter 2. Cable modeling X: 10.6 Y: 1279 Inverter output voltage Motor input voltage Time [ sec)] Figure 2.19: The port voltage signal (in blue) and the far end voltage signal (in green). 2.5 Conclusion The grounding method has been studied in section 2.2, the screen current estimation in section 2.3 and the termination effects of the power cables in section 2.4. From section 2.2, we could observe that the current can flow in the system without difficulties at any frequency when the common ground is used. The separated grounds do not facilitate the power flow at low frequency. The second section shows that the screen current magnitude depends on the frequency. The screen current can increase and reach more than 11 times the center wire cable until at resonant frequency. The last section shows that during the transient period, the terminal voltage amplitude depends on the load impedance. The over voltage occurs when the load impedance is large compared to the cable impedance. Oscillating phenomena are the consequences of the propagation voltage wave and the frequency of these oscillations depends on the cable length.

28 Chapter 3 Converter Models 3.1 Introduction This chapter will focus on the modeling of the internal components of a 1500V AC-DC motor converter module. The components include the IGBTs, The busbar interconnections and the DC link filters. Some simplifications have been made in order to allow good understanding of what happens. Figure 3.1 illustrates how the structure has been simplified for the study. The DC source is discrete port from the DC minus to the DC plus as shown in figure 3.2. Figure 3.3 shows the load in star connection here modeled by the discrete ports. The high frequency responses of the IGBTs are modeled using the following two approaches: Parasitic model of the IGBT; IGBTs modeled as voltage controlled switch. Only the high power conducting part of the converter module has been modeled. This includes the IGBTs, The DC bus-bars, The bus-bars or the AC-side and the input filters. The low voltage components, for example, the gate drive unit and the drive control unit are not modeled. Figure 3.3 shows the setting of the load (connected to the AC side) and the DC signal source. The DC signal source is discrete port (figure 3.2) from the DC minus to the DC plus. 18

29 Chapter 3. Converter Models 19 Figure 3.1: Simplification of the model Figure 3.2: DC link excitation Figure 3.3: Load setting in AC side Figure 3.4: Simultion setting 3.2 Parasitic modeling of the IGBTs This approach consists of replacing the IGBT by lumped elements. For instance, inductance can be used for the on-state and capacitance can be used for the offstate as presented in [2]. The main advantage of this model is that it does not take a big amount of memory. Then, it is fast, easy to check and it gives a very

30 Chapter 3. Converter Models 20 Figure 3.5: Wire copper connecting terminals of IGBT Figure 3.6: Lumped element connecting the IGBT terminals good idea of the distribution of the electromagnetic field in the system in the frequency domain and the EMI characteristic of the system. Furthermore, the instantaneous load power is consistent when compared to simple lumped circuit converter models. The < 110 > state shown with the equivalent circuit given in figure 3.6, results shown in figure 3.7. It gives also an idea of the resonance behavior of the power converter module Parameter estimation Table 3.1 shows the IGBTs parameter used in this simulation and figures 3.5 and 3.6 shows the connection setting in CST MW design studio on phase U. Figure 3.7 shows the excitation signal from the DC bus parameters with the following values: Rise time: 0.2 micro seconds; Fall time: 0.2 micro seconds;

31 Chapter 3. Converter Models 21 Table 3.1: IGBTs model IBGT states Physical connections Lumped element connections On-state Copper wires 10nH inductance Off-state No-connections 235 pf capacitance Figure 3.7: Excitation signal pattern Figure 3.8: Schematic of the circuit. The < 110 > state of the inverter is simulated with 100Ω in star connection Maximum voltage 1500 V. The inverter will be in the configuration < 110 >. This means that in steady state the voltage will distribute itself according to equivalent circuit shown in figure 3.8. Since the system is almost lossless, the system can be analyzed as lossless transmission line. The are two IGBT cases considered in the < 110 > state. This includes: 1. On and Off states represented by physical connections; 2. Lumped elements with values given in Table 3.1.

32 Chapter 3. Converter Models 22 Figure 3.9: Voltage responses Observations The following observations were made: Figure 3.9 and 3.10 show small oscillations can be observed on the responses at about 20 MHz. These oscillations are coming from the resonance behavior including from the IGBT parasitics, the bus parasitic inductances and distributed capacitances. The ripple amplitude is bigger on the DC link signals. That is because the situation is comparable with opening and closing a voltage source on the line terminals. There is increased damping in the case with lumped elements IGBT ripples as shown in figure 3.9 and 3.10.

33 Chapter 3. Converter Models 23 Figure 3.10: Current responses Results The voltage and the current responses have been measured on the DC and the load side in both cases and respective results can be compared. Using the schematic in figure 3.8, the DC link is excited with 1500V dc. In the < 110 > state, the steady state voltages and currents are summarized in Table 3.3. Figure 3.9 and 3.10 show that the steady state model results are consistent. However, due to the parasitic capacitances and inductances, we can observe the presence of the ripples.

34 Chapter 3. Converter Models 24 These ripples depend only of the converter geometry and materials. The frequency of these oscillations is about 20 MHz. Table 3.2: System parameters with filters New IGBT models DC link capacitor On-state: 10nH inductor series Capacitance of with 1 mohm resistor 4mF Off-state: 235 pf 15 Ohm series (charging resistor) EMC filter One resistor: 4.7 Ohm 3 capacitors: 4 micro farads The most invisible effect of the parasitic capacitances and inductances is the DC link. In both models, the ripple amplitudes are very large. The wire model presents higher damping path than the lumped elements model Improvement of the damping The 20 MHz ripple amplitude could be reduce in different ways. For instance, by adding filters in the system or reducing the dv/dt. This section will present the first method. The filter parameter are presented in the table 3.2. Figure 3.11 shows the setting in 3D, while figure 3.12 shows the lumped circuit. The voltage and current responses with the filters installed are shown in figures 3.13 and 3.14 respectively. Results Smaller oscillations can be observed on the responses at 20 MHz. The oscillation are damped faster than in the simplified model. Table 3.3: Mean values of voltage and current response Elements DC-link U V W Voltage [V] Current [A]

35 Chapter 3. Converter Models 25 Figure 3.11: Setting of the system with the filter Figure 3.12: Schematic of the filter The filter capacitors charged during the simulation. The DC link capacitor increases the supply current because it is empty at the beginning of the simulation as.

36 Chapter 3. Converter Models 26 Figure 3.13: Voltage responses 3.3 IGBTs modeled as voltage controlled switches Description Connectors have been created in the IGBTs locations in the converter box module as shown in figure A zoom of the IGBT connectors is shown in figure These terminals have been connected with the switches. Control voltage of the switches are delivered by the port excitation in CST design studio. The drawback of this approach is the increase in simulation time, and it requires more memory.

37 Chapter 3. Converter Models 27 Figure 3.14: Current responses However, it provides a means of controlling the IGBTs to simulate different modulation patterns and observe the impact on the EM environment. IGBT models could be imported from different softwares or libraries. Regarding the DC link setting, a charged 4mF capacitor is used. It imposed 1500 Vdc between these converter terminals. It is large enough to maintain constant voltage during the simulation. Also, a connecting 30Ω resistor is used for limiting the current when the capacitor makes the contact with the structure. In this chapter, results obtained using Voltage controlled switches (VCSW) and imported IGBT from Pspice are compared. VCSW offers the advantage to control the rise time during the simulation while imported IGBTs from Pspice have the fixe rise time.

38 Chapter 3. Converter Models 28 Figure 3.15: Motor converter module with the IGBT ports Figure 3.16: IGBT ports Results It could be observed that the system reacts at the DC link voltage connection. Concerning the switching pattern, all the three phases are connected in the same time. The ringing strongly depends on the voltage excitation or/and the rise time.the riging strongly depends on the voltage excitation rise time. The voltage responses are presented in figures 3.17 and Figures 3.19 and Reducing the dv/dt, reduces the ringing. Figure figure 3.21 presents the voltage with the rise time increased to 0.4 µs. There is less ripple in figure 3.21 compared to figures 3.17 and Conclusion The resonance behaviour on a motor power converter has been studied in section 3.2 and IGBTs models have been proposed in section 3.3. From section 3.2,

39 Chapter 3. Converter Models 29 Figure 3.17: Voltage responses using imported IGBTs Figure 3.18: Voltage responses using VCSW

40 Chapter 3. Converter Models 30 Figure 3.19: Current responses using imported IGBTs Figure 3.20: Current responses using VCSW

41 Chapter 3. Converter Models 31 Figure 3.21: Voltage responses using VCSW with rise time equals 0.4 µs the parasitic resonance aspects of the converter could be predicted. The introduced ripples could be damped by using filters or reducing the speed the voltage variations (dv/dt) of the source. This approach is fast but it does not allow to change the switching configurations of the MCM without interrupting the simulation. Section 3.3 allows to switch the IGBTs while the simulation is running. The dv/dt could be also modified in order to reduce the ripples after switching. It is more realistic but it takes more memory and uses different physics in the time. This inverter could be coupled with cables and the 2D and 3D results could be obtained as done in chapter 4

42 Chapter 4 Converter and Cable Coupling 4.1 Description This chapter focuses on the creating of the complete model of the motor drive system. The cable model presented in chapter 2 is coupled to the motor converter module (MCM) model presented in chapter 3. The motor load is simply represented as a Y-connected 100Ω resistors. The DC-link capacitor is charged through 30Ω resistor to charge the converter. Figure 4.1 presents a schematic of the complete drive system model. Unshielded cables are used to connect the load, just for study purposes. The cable configuration is presented in figure 4.2. Figure 4.1: schematic of the system 32

43 Chapter 4. Converter and Cable Coupling 33 Figure 4.2: Cable setting Connections have been created in the switch locations in MW studio in order to connect the terminals with the voltage controlled switch as illustrated in figure 4.4. The simulation parameters are the following: DC link steady state voltage: 1500 V; The load: 3 resistors of 100 Ohms in star connection; Commutation from state 110 and 010; The used cables are 5m unshielded cables with parameters given Table 2.1.The MCM is constructed in MW studio, while the cable model is constructed in CS studio. The MCM is coupled with cables model in MS design studio. Outside the MCM, the DC minus is grounded directly, while in the converter the DC minus is grounded through the EMI filters. Figure 4.5 illustrates the DC minus grounding through EMI filter. The meshing should be fine enough for allowing the connection between cables and metallic structure. The possible ways of doing that is maintaining the frequency

44 Chapter 4. Converter and Cable Coupling 34 Figure 4.3: Cables coupled with converter and increase the wavelength ratio or maintaining the wavelength ratio and increase the frequency. By changing these parameters it is possible to find the solution which minimizes the time of simulation. The schematic of the complete system is shown in Figure??. The complete system in 3D view is shown in Figure 4.3, the 3D model is in-closed in the white box in the center. Only the external (hidden) lumped connections are visible on the schematic, for example the voltage control switches and the excitation ports. 4.2 Voltage and current responses Figure 4.7 presents the load voltages while figure 4.8 dc link voltage and current and and the ground current. On the DC link, a first transient could be observed that corresponds to the contact between structure and dc link capacitor. When the current reaches zero amperes, the charging resistor is short-circuited and the voltage remains at 1500V. Between 0.5µs and 1µs, the converter switches to < 110 >, between 2µs and 3µs, it commutate from < 110 > to < 010 > and

45 Chapter 4. Converter and Cable Coupling 35 Figure 4.4: IGBT locations Figure 4.5: Grounding and EMC filter setting

46 Chapter 4. Converter and Cable Coupling 36 Figure 4.6: schematic of the system

47 Voltage V Voltage V Voltage V Chapter 4. Converter and Cable Coupling Phase U load voltage time s 1500 Phase V load voltage time s 500 Phase W load voltage time s Figure 4.7: load Voltages

48 Current in Ampere Voltage Volt Chapter 4. Converter and Cable Coupling DC link voltage time s DC link current and ground curent DC link current Ground current time s Figure 4.8: DC link current and voltage

49 Chapter 4. Converter and Cable Coupling 39 Figure 4.9: 3D Electric field at 10 khz Figure 4.10: 3D Electric field at 500 MHz Figure 4.11: Cross section at the middle of the line at 10 khz Figure 4.12: Cross section at the middle of the line at 500 MHz at around 4µs the switches on the first leg are both opened. Figure 4.7 and figure 4.8 show the voltage response and the current response respectively D field distribution Once the current and voltage distribution in the model are obtained, the EM fields can be computed. Figure 4.9 shows the electric field at 10kHz, while figure 4.10 shows the E-field at 500MHz. Figures 4.11 and 4.12 shows the E-fields in a plane traversing the cables. The magnetic fields are distributed at 10kHz is shown in figure 4.13 while the H-field at 500MHz is shown in figure 4.14.

50 Chapter 4. Converter and Cable Coupling 40 Figure 4.13: 3D Magnetic field at 10 khz Figure 4.14: 3D Magnetic field at 500 MHz 4.4 EM field 2D plots Figure 4.15: Probe location in the system Some probes have been put on different locations in the electric drive vicinity in order to measure the EM fields. Figure 4.15 shows the probe locations. The following probe results are shown for these locations: A Probe measuring the E-field and The H-field to an IGBT inside the MCM; B Probe measuring the E-field and The H-field 1 m left to the MCM; C Probe measuring the E-field and The H-field 1 m above the mean length of the cables;

51 Chapter 4. Converter and Cable Coupling 41 Figure 4.16: Electric field in point A, close to the IGBT in V/m Figure 4.17: FFT Electric field in point A, close to the IGBT V/m D Probe measuring the E-field and The H-field close to the mean length of the cables;

52 Chapter 4. Converter and Cable Coupling 42 Figure 4.18: Magnetic field in point A, close to the IGBT in A/m Figure 4.19: FFT Magnetic field in point A, close to the IGBT in A/m

53 Chapter 4. Converter and Cable Coupling 43 Figure 4.20: Electric field in point B, left to the MCM in V/m Figure 4.21: FFT Electric field in point B, left to the MCM in V/m

54 Chapter 4. Converter and Cable Coupling 44 Figure 4.22: Magnetic field in point B, left to the MCM in A/m Figure 4.23: FFT Magnetic in point B, field left to the MCM in A/m

55 Chapter 4. Converter and Cable Coupling 45 Figure 4.24: Electric field in point C, 1m from the cable middle in V/m Figure 4.25: FFT Electric field in point C, 1m from the cable middle V/m

56 Chapter 4. Converter and Cable Coupling 46 Figure 4.26: Magnetic field in point C, 1m from the cable middle A/m Figure 4.27: FFT Magnetic fieldin point C, 1m from the cable middle in A/m

57 Chapter 4. Converter and Cable Coupling 47 Figure 4.28: Electric field in point D, close to the cables in V/m Figure 4.29: FFT Electric field in point D, close to the cables in V/m

58 Chapter 4. Converter and Cable Coupling 48 Figure 4.30: Magnetic field in point D, close to the cables A/m Figure 4.31: FFT Magnetic in point D, close to the cables in A/m

59 Chapter 4. Converter and Cable Coupling Conclusion Chapter4 describes the 3D assembly of motor converter module and unshielded cables. The steady state of the voltage and current responses present the same results that could be obtained from a 2D simulator and including ripples due to the parasitic resonant characteristic of the system. A resistive load has been connected to the system. This load allows high di/dt and dv/dt in the system. 3D results in section4.3 show how the electric field is perpendicular to the ground in low frequencies even though it is not the same in high frequencies. The magnetic field has an uniform distribution in low frequency unlike the high frequency field distribution. Section4.4 shows the 2D measurements of the electromagnetic field in the vicinity of the electric drive system. The converter shields the electromagnetic radiations produced by the switches. Further investigations could be done in order to determine the more critical frequency components of the fields and thus prevent the unwanted interferences with the signalling system.

60 Chapter 5 Discussion and Future Work 5.1 Discussion CST-computer simulation technology is a very suitable tool for the 3D modelling of the drive system. It allows for a suitable integral of the 3D models, the cable transmission line models and the lumped circuit models. Specific IGBT models can be imported. Different load cycles or modulation schemes can be simulated from the lumped circuit interface in CST MS and DS, to obtain most case field distributions. The the aim of the thesis, multiple physics have been used and this used leading to obtained results. However, some results cannot be used directly from CST. For example, the frequency domain (Fourier transform) should be checked carefully. It is more prudent taking the time domain results and making the frequency domain analysis using a different tool like Matlab. Also, from [2], the impedance value (complex form) also have to be reviewed for avoiding mistakes in the conclusion. 5.2 Future Work The thesis only investigate the converter and cable used in a propulsion system. The use of resistive load allowing the model stability and robustness. Fast di/dt and dv/dt could study and they give an idea of the most critical case. Next step, could be the 3D motor coupling to be able to study the winding effect and completing the common current circuit. Furthermore, the achievement of the 50

61 Chapter 5. Discussion and Future Work 51 control system (signal cables, IGBT drivers and other sensors) could ease the prediction of major issues in the drive system. 5.3 Conclusion Electromagnetic phenomena are present in electrical drive systems. They can come out from different sources but this thesis work focused on converter feeding high frequency electrical components into the cables. In this thesis CST has been used to create 3D models of a motor drive system. The 3D models has been integrated with lumped circuit interface to simulate the different load cycles. The current and the voltage distribution is also obtained. The created model with some modifications could be used in projects to predict characteristics early in the design.

62 Bibliography [1] Prof Michele Tartaglia. Tensioni indutte nelle guaine dei cavi di potenza. Tensioni indutte nelle guaine dei cavi di Potenza, 20: , [2] Ensa Sinyan. Modeling of resonances in a converter module, including characterization of igbt parasitics [3] Fabian Streiff. Sbb fv dosto: Currents in motor cable screens - test repory. [4] Electrotechnical modeling and design. KTH,