Ready-to-copy solutions reduce development costs May 2006 Excellent references The time window from the start of product development up to market launch is critical for the commercial success of every product. In order to keep the time-to-market as short as possible and to assure perfect operation from the outset, semiconductor manufacturers offer suitable reference designs also known as ready-to-copy solutions. A typical example is the circuit of an electronic ballast for fluorescent lamps that was produced in cooperation by Infineon and EPCOS. Infineon has developed the ICB1FL01G, a new integrated circuit designed to control the electronic ballasts of fluorescent lamps of types T4 and T5. The IC offers many different options, including multi-lamp operation. Infineon simultaneously reduced the external circuitry required to operate the IC (Fig. 1). The new fluorescent lamps of types T4 and T5 are optimized to high luminous efficiency. To ensure a long operating life, however, they require a specific start process and more intensive monitoring of their operating behavior than the older lamp types T8 and T12. The ICB1FL01G ballast controller operates a boost converter as an active PFC harmonic filter. An innovative coreless transformer technology allows the MOSFETs of the converter in the half-bridge circuit to be driven by the level-shift procedure. The converter supplies the fluorescent lamp via an oscillating circuit so that a change in operating frequency controls the pre-heating, ignition and nominal operation statuses in sequence. The module detects both a change of lamp and the dangerous rectifier effect at the end of a lamp s service life and does so in configurations comprising between one and four lamps. The IC processes both analog and digital control signals so that process routines and evaluation criteria can be precisely controlled via adjustable and fixed time windows. The time and frequency-determining parameters are set simply with the aid of resistors. The ICB1FL01G ballast controller is manufactured in 20-V BICMOS technology with a structure width of 0.6 µm and three-layer metalization. FIGURE 1: LAMP BALLAST Circuit diagram of the new lamp ballast. The specification and ordering numbers of the components are included in the materials list. The components highlighted in red are from EPCOS. EPCOS AG 2011 All rights reserved www.epcos.com 1 / 10
Function principle of the circuit After switching on the power, filter capacitor C2 and smoothing capacitor C10 are charged up to the peak value of the line voltage. These filter capacitors protect the circuit from high- and low-frequency ripple currents, high surge voltages and brief voltage fades. EPCOS has developed the B43858, B43888, B43866 and B43867 series of capacitors especially for these applications: their properties and dimensions predestine them for use in lamp ballasts. The capacitance of these aluminum electrolytic capacitors designed for a high ripple current capability ranges from 2.2 to 330 µf, whereas their permissible operating voltages extend to 450 V and their maximum dimensions are 22 x 40 mm 2. Low-profile versions with a maximum insertion height or diameter of 14 mm are also available. In this reference design, C10 is a capacitor of the B43888 series with key data 10 µf/450 V (single-ended, diameter 12.5 mm). For this purpose, EPCOS offers high-temperature versions (105 C) in a low-profile package that does not exceed the maximum insertion height of 14 mm. Capacitors C12 and C13 that support the supply voltage Vcc of the ICB1FL01G are charged via starting resistors R11 and R12. In this operating phase, the ICB1FL01G draws less than 100 µa until the supply voltage Vcc has reached a value of 10 V. A current source with a typical value of 20 µa is then activated at the RES pin, thus detecting the presence of the low-level heating coil. As long as the voltage level at the RES pin is below 1.6 V, the IC assumes that the coil is undamaged. Resistor R36 lies in the path of the measuring current. It matches the voltage drop and, in conjunction with capacitor C19, extracts the AC voltage occurring across the heating coil during operation. In addition, zener diode D10 is placed between the RES pin and ground in order to protect the terminal from any voltage surges that may occur during operation when the lamp is changed. A current is led via resistors R34 and R35 through the heating coil that is under high voltage and then via resistors R31, R32 and R33 to pin LVS1. Wherever the current exceeds 15 µa, the controller detects a heating coil. A second measuring input LVS2 is available for multi-lamp operation. It can be deactivated if required by a short circuit to ground, exactly like LVS1. The current fed at the LVS pin is diverted to the supply voltage Vcc and in this way supports the start. If the measuring current at the LVS pin is too low, the controller signals this disturbance by a higher current level with a typical value of 41 µa at the RES pin, so that the voltage drop at R36 exceeds the 1.6-V threshold and prevents a controller start. When the presence of the heating coil is detected in this way and the voltage at pin PFCVS is at least 0.375 V, which is interpreted as a closed control loop for the boost converter, the ICB1FL01G can activate its driver outputs in the event that the supply voltage Vcc has exceeded the turn-on threshold of 14 V. Switching losses of the converter in nominal operation The first control signal turns on the low-level MOSFET Q3 in the half-bridge so that the floating capacitor C14 which supplies the high-level control logic is charged from capacitor C13 via R30 and D6. R30 prevents a response of the surge current switch-off circuit at the LSCS pin. This means that the high-level MOSFET Q2 can already be driven at the following half-clock. Capacitor C16 is connected to the output of the half-bridge converter: it acts as a charge pump together with diodes D7 and D8. The continuous recharging of C16 at the converter frequency shifts portions of energy to C13 for the Vcc supply of the IC. Excess energy is diverted away by zener diode D9. C16 is also used to limit the voltage rise as well as to implement the zero-voltage switching operation. During nominal operation, the inductively driven current of the load circuit recharges capacitor C16 in the key gap of MOSFETs Q2 and Q3 without loss, so that the subsequent turn-on process at the MOSFET takes place at zero voltage. At the turn-off process, C16 limits the voltage rise so that the MOSFET channel is already blocked before the drain-source voltage increases appreciably. As the consequence, the switching losses of the converter are negligible in nominal operation. The load circuit of the converter consists of a series oscillating circuit comprising resonance choke L2 and resonance capacitor C20. To ensure that the inductor fits into a standard housing for T5 lamps without problems, the EFD25 type is used here with an insertion height of only 14 mm. Pin PFCCS of the ICB1FL01G is a surge current limiter for the boost converter and the pin PFCVS is an input for regulating the intermediate circuit voltage to detect overvoltages, undervoltages or an open control loop. EPCOS AG 2011 All rights reserved www.epcos.com 2 / 10
FIGURE 2: TRANSFORMER FOR THE LAMP CONTROL The transformer for lamp control of the EFD25 type has an insertion height of only 14 mm and is thus eminently suited for the design of the new lamp ballasts. The fluorescent lamp is connected in parallel to the transformer. In this example, the lamp is pre-heated with voltage control so that another two windings are applied to resonance choke L2 (Fig. 2): each of them supplies the heating coils with power via a bandpass filter consisting of L21/C21 or L22/C22. If lamps with a lower ignition voltage are used, the two coils can be wound directly onto L2. For ignition voltages with effective values of over 1 kv, in contrast, additional insulation as well as vacuum impregnation of the coils is required. As an alternative to impregnation, a multichannel coil former may be used with placement of the heating coils in the upper and lower parts of a total of four chambers. In the case of vacuum impregnation, gas that can be ionized is replaced by impregnating resin, so that the operating life of the resonance choke cannot be shortened by a partial discharge. Modern lamp chokes from EPCOS can withstand several hundred thousand ignition processes unscathed even at a nominal operating temperature of 100 C. The bandpass filter comprising L21 and C21 or L22 and C22 ensures that current is applied to the heating coils only in the pre-heat phase at an adjustable pre-heat frequency. In nominal operation, in contrast, the heating current is largely prevented by suitable modification of the operating frequency (Fig. 3). The load circuit also contains a decoupling capacitor C17 that charges to half the intermediate circuit voltage in nominal operation and thus operates the lamp symmetrically to the ground potential of the rectified line voltage. FIGURE 3: CHANGE OF OPERATING FREQUENCY Start phase of the inverter with lamp voltage (red), voltage at the output of the inverter (black) and heating current (blue). It is switched off by the bandpass filter after preheating (1000 ms) and ignition of the lamp. Integrated PFC Simultaneously with the converter, the IC drives MOSFET Q1 of the PFC boost converter. This circuit part consists of choke L1, diode D5, MOSFET Q1 and storage capacitor C10. This kind of boost converter can continuously transform an input voltage to a higher output voltage. With suitable driving, this converter acts as an active harmonic filter and also corrects the power factor, so that a sinusoidal current is drawn from the supplying AC voltage network and a regulated DC voltage is made available at intermediate circuit capacitor C10 (Fig. 4). EPCOS AG 2011 All rights reserved www.epcos.com 3 / 10
Here, the PFC boost converter operates for a controlled turn-on duration without recording the input voltage so that it remains switched off after a turn-on duration specified by the controller until the current in the choke, and thus also in the boost converter diode, has decayed to zero. This point in time is detected from the level at the detector winding and notified to the controller via R13 at pin PFCZCD. A continuous triangular current (critical conduction mode) then flows in choke L1 for a control range of the turn-on duration between 2.3 µs and a maximum of 23 µs. As the energy flow is reduced further, the turn-on duration is shortened increasingly down to 0.4 µs, whereas the turn-off duration is simultaneously lengthened beyond the point of zero-current detection. A triangular current with gaps then flows (discontinuous conduction mode). This control procedure allows the boost converter to operate in stable mode over a broad range of input voltages and loads. FIGURE 4: REGULATED DC VOLTAGE DC voltage (410 V, black) at the output of the PFC boost converter, line AC voltage (210 V, red) and drawn line current (280 ma, blue) For L1 too, the developers selected ferrite core type EFD25 (Fig. 2). In this case, the inductor has a broadband design with a chamber construction. When a low voltage is applied to the ballast, the temperature in the main winding of the choke can rise to over 100 C. The ferrite material must also permit rapid magnetic reversal at these temperatures and exhibit low losses. As the core losses of ferrite materials N87 and N97 from EPCOS reach a minimum at temperatures of around 100 C, they are particularly well suited for use in these applications. The material selection is critical for the reliability of both PFC choke L1 (EFD 25) and resonance choke L2 (EFD 25) over their entire operating life. Pre-heating the lamps The converter starts with an operating frequency of 120 khz. Within 10 ms, this frequency changes in 16 steps to the pre-heating frequency that can be set with resistor R22. The operating frequency of the converter remains at this level for the duration of the pre-heating time that can be set between zero and 2000 ms with resistor R23. It then changes within 40 ms in 128 steps from the pre-heating frequency to the running frequency f RUN that can be set with resistor R21. The ballast circuit must be designed so that in the pre-heating phase the voltage at the lamp is minimized and a high heating current simultaneously flows through the heating coils. High voltage in the ignition phase In the ignition phase that follows immediately after the pre-heating period, the frequency change of the converter should be either above or at least close to the resonant frequency of the oscillating circuit so that the voltage at the lamp is sufficiently high for ignition. After ignition, when the nominal frequency is reached and after the transient phase, the lamp current should reach its nominal value and the heating current of the coils be at a minimum. A high voltage and a high resonant circuit current occur at the lamp in the ignition phase due to the resonant circuit (Fig. 5). This current is monitored by resistors R24 and R25. As soon as the voltage at the LSCS pin exceeds a level of 0.8 V, the converter operating frequency is increased by several steps in order to prevent a further rise of the current and thus also the voltage at the lamp. As the level of 0.8 V is no longer attained at the LSCS pin, the converter operating frequency drops with the step width typical for the ignition phase to its nominal value. As a result of this measure, the ignition phase is extended from 40 ms to up to 235 ms at a lamp with sluggish ignition, whereas the EPCOS AG 2011 All rights reserved www.epcos.com 4 / 10
voltage at the lamp remains at the level of the ignition voltage with a certain ripple. EPCOS AG 2011 All rights reserved www.epcos.com 5 / 10
FIGURE 5: IGNITION PHASE Start phase of the inverter with lamp voltage (red), voltage at the converter output (black) and lamp current (blue) that sets in after the pre-heating (1000 ms) and ignition of the lamp Protection functions and error statuses If the nominal operating frequency is not reached within 235 ms of the end of the pre-heating period, the ICB1FL01G changes over to error status. The gate drivers are then blocked, the power consumption is reduced to a typical value of 150 µa and the heating coil detection is activated. It restarts either when the lamp is changed or after another turn off and on of the line voltage. Numerous protection functions supplement the functional scope of the ICB1FL01G. As soon as the level at the LSCS pin exceeds the threshold of 1.6 V for longer than 400 ns, this is recognized as a dangerous operating status due to the lamp being removed during operation or to a line voltage surge, and the ICB1FL01G changes over to error status. If the converter deviates from its typical zero-voltage switching in nominal operation so that peak currents occur when the MOSFET is turned on, for example due to the sudden recharging of charge pump capacitor C16, the ICB1FL01G interprets this operational event as a capacitive load. Such an operational condition may occur when the lamp is removed. The ICB1FL01G distinguishes two different types of capacitive load. In a first case, partial recharging occurs of charge pump capacitor C16. This case is less critical so changeover to error status occurs only when this error lasts at least 500 ms. In a second case, charge pump capacitor C16 is completely recharged by the MOSFET. The inverse diode of the MOSFET may also be commuted by the current flow. In such a critical case, the ICB1FL01G already changes over to error status after 605 µs. Finally, dangerous operating conditions occur when the fluorescent lamp reaches the end of its operating life or becomes thermally instable due to impermissible operating conditions. This leads to unsymmetrical or excessive lamp currents and voltages. In this case, too, the ICB1FL01G changes over to error mode. Electromagnetic compatibility It is in the nature of electronic ballasts to generate strong conducted interference due to the circuit principle. At the same time, the insertion space is greatly limited, especially upwards. To suppress this kind of interference, the input filter is provided with C1, C3, C4, C5 (Fig. 6) and L101 (frame core choke). The new X2 film capacitors from EPCOS offer compact dimensions of only 4 x 9 x 13 mm 3 in the smallest version. Different can variants for each capacitance allow highly compact and very flat circuit layouts to be implemented. The series can be operated up to a voltage of 310VAC and a maximum permissible continuous temperature of 125 C. EPCOS AG 2011 All rights reserved www.epcos.com 6 / 10
FIGURE 6: X2 FILM CAPACITORS The new X2 series offers impressively compact dimensions. EPCOS has further developed the FC17 (Flat-frame Core) and FC23 series especially for use in TV sets, switch-mode power supplies and electronic ballasts. In the reference design, L101 (Fig. 7) has been implemented with the FC23 type. Thanks to their high leakage inductance, both FC17 and FC23 are distinguished by good interference suppression and simultaneously exhibit the lowest electrical resistance in the industry. The sector winding results in a high resonant frequency and the compact design was developed specially for use in lamp ballasts. To achieve a maximum inductance with a simultaneously low cross-sectional area of the core, the frame core chokes consist of a closed core. These cores must therefore be wound by ring-core winding machines. Frame cores used to be initially inserted into the coil former and subsequently wound. However, such insertion and the formation of air gaps and creepage distances led to unnecessarily long winding wires. EPCOS uses a simple and effective way of minimizing the lengths of the windings: instead of a winding form, an insulated core is wound directly. This procedure had previously been used only for toroidal cores. FIGURE 7: FRAME CORE CHOKE The new frame core chokes offer high leakage inductances with simultaneously low resistances. VARIOUS CHOKE SERIES Frame core choke RK17 (competition) D-core choke EPCOS B82732 E-core choke B82731-T Rated current 400 ma 700 ma 550 ma DC resistance 2550 mω 2000 mω 2800 mω Leakage inductance 740 µh 330 µh 800 µh Table 1: Comparison of various choke types EPCOS AG 2011 All rights reserved www.epcos.com 7 / 10
All electronic modules such as switch-mode power supplies or electronic ballasts must satisfy the relevant safety requirements. For this purpose, the leakage currents must be kept below a maximum value so that the Y-capacitors need not be too largely dimensioned. Developers prefer asymmetrical chokes with a high inductance for these applications. On the other hand, modules of this type continuously produce both common and differential mode interference. Currents flowing to ground via parasitic capacitors between a cable and ground then generate electromagnetic common mode interference. In Fig. 8, these currents are designated I COM. The common mode current I COM is the sum of the partial currents that flow in the same direction in both leads: I COM = I COM1 + I COM2. This total common mode current flows via the parasitic capacitors C P1 and C P2 to ground and thus generates magnetic fields of the same amplitude and polarity that do not mutually cancel out. The common mode current can therefore generate an electromagnetic field outside the cable, which acts like an antenna. FIGURE 8: COMMON MODE INTERFERENCE Representation of common mode interference Another source of common mode interference is the distributed transformer capacitance in DC/DC converters (Fig. 9). The dv/dt generated by the power transistor causes a current to flow from the input to the output circuit through the distributed capacitances (C p1 to C px ) of the transformer. Every time the power transistor switches, a large dv/dt is passed via the parasitic transformer capacitances to the input stage, and this in turn results in a current flowing from the input to the output through the transformer capacitance. This current flows twice in each switching cycle and does so simultaneously to all output pins and to all terminals of the transformer. FIGURE 9: CIRCUIT DIAGRAM OF A POWER SUPPLY Formation of common mode interference by parasitic transformer capacitances Combating interference signals EPCOS AG 2011 All rights reserved www.epcos.com 8 / 10
Figure 10 shows a conventional interference filter. L1 and L2 are always optimized to relatively high inductances of up to 100 mh so that a high common mode impedance exists. However, this results in a mismatch with the low common mode impedance of the power supply line. L1 and L2 are wound around a single core, the two windings being as identical as possible so that the magnetic fluxes resulting from differential mode currents in the core compensate each other mutually to avoid saturation of the actual core. Although the magnetic fluxes resulting from the common mode currents add up, they do not saturate the core. Because this impedance is inductive, it is particularly effective at low frequencies. FIGURE 10: CONVENTIONAL INTERFERENCE FILTER The use of the new frame core chokes means that L3 and L4 are obviated. This results in much lower impedance and lower losses. Y-class capacitors C y1 and C y2 are placed between phase (or zero conductor) and ground in order to operate at high frequencies. However, they cannot be too largely dimensioned, as national regulation authorities have specified upper limits for the leakage currents of the equipment. If it turns out that these capacitors do not operate as efficiently as desired, an inductor L 5 may be required. The resistor R is used to discharge C x. An X-class capacitor is inserted between the phase and zero conductor (C x ) in order to attenuate differential (symmetrical) interferences. Here, parasitic leakage inductances of L 1 and L 2 have additional positive effects on the attenuation of the differential mode EMI. Strong electromagnetic interferences normally require the use of additional non-compensated iron powder chokes (L 3 and L 4 ) to improve the differential-mode attenuation. The new FC17 and FC23 chokes from EPCOS obviate these additional chokes thanks to their high leakage inductance. For this reason, the FC17 and FC23 chokes combine two or three components in one. On the electronic ballast reference board, the FC17 type with 2 x 68 mh/0.65 A and ordering number B82732F2651A001 from EPCOS performs these functions. EMC-compliant testing and certification EPCOS also offers an extensive range of measuring options in its own accredited laboratories: everything can be tested, from small modules via motors and converters up to measurement objects the size of an automobile. In addition to the interference suppression of completed modules such as this reference design board, EPCOS may also be involved at an early stage of the development process. EPCOS can test modules to all the usual standards, and if required modify and certify them. An extra large hall is also available for free-field measurements. EPCOS AG 2011 All rights reserved www.epcos.com 9 / 10
BILL OF MATERIAL FOR BALLAST RFERENCE DESIGN Table 2: Complete materials list for a lamp ballast ( Table 2 as PDF) Authors: Wolfgang Dreipelcher (Sales Reference Design Accounts, EPCOS), Bernhard Roellgen (Product Development Inductors, EPCOS), Michael Herfurth (Technical Marketing, Infineon) EPCOS AG 2011 All rights reserved www.epcos.com 10 / 10