Experimental Analysis of Common Mode Currents on Fibre Channel Cable Shields due to Skew Imbalance of Differential Signals Operating at 1.

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1 Experimental Analysis of Common Mode Currents on Fibre Channel Cable Shields due to Skew Imbalance of Differential Signals Operating at 1.06 Gb/s James L. Knighten Joseph T. (Ted) DiBene II Norman W. Smith Convergence Design NCR Corporation 9948 Hibert St., Suite Via de1 Campo San Diego, CA San Diego, CA Lothar 0. Hoeft Consultant, Electromagnetic Effects 9013 Haines Ave., NE Albuquerque, NM Abstract: The spectral nature of common mode currents induced on high speed differential cables operating at 1.06 Gb/s was investigated using specially constructed shielded test boards. The source test board provided a source with a selectable amount of delay skew. The load test board provided a simple 150 Ohm differential load. The two boards were placed in separate shielded enclosures with a one meter Fibre Channel cable connecting them. Common-mode cable shield current and radiated emissions at 3 meters were measured as a function of delay skew. At the fundamental frequency of MHz, common-mode current and radiated emissions increased at a rate of approximately 9 db/decade of skew. At skew values much lower than the rise time of the signal, common-mode current increased nearly linearly with skew. The second harmonic was present on the cable shield due to duty cycle distortion and rise and fall time differences inherent in the driver transceiver. INTRODUCTION Digital computing systems sometimes employ high speed serial interconnects to transfer data. A massively parallel processing (MPP) system is designed for scalable data warehousing applications and employs a high speed serial data interconnect based on the Fibre Channel protocol at 1.06 Gb/s [l]. Such a system may employ a large number of cables carrying high speed serial data. Common-mode currents, rather than normal operating currents, are often primary generators of EM1 [2]. Common-mode currents are usually created by asymmetries, both mechanical and electrical and are often not apparent from examinations of electrical schematic diagrams. High speed digital signaling is often implemented via dzj kential signaling on two-conductor transmission line structures. (This is not the differential-mode current described in [2], but is a transmission line structure of two signals, propagating in the odd-mode, that are equal and opposite with a virtual ground reference.) In the ideal case, differential signaling produces no net current down the transmission line. When an imbalance in the differential signal pair occurs, the resulting modes of propagation on the transmission line are both odd-mode and even-mode. The even-mode, where the two wires carry equal signals, represents the common-mode signal on the transmission line. In a shielded transmission line, the common-mode current returns to the transmitter via the inner surface of the shield. This current becomes an EM1 issue as it egresses the shield by means of the common-mode surface transfer impedances of the cable and connectors. High speed transmission lines used in digital circuitry are usually designed to minimize imbalance as the characteristic impedances and path lengths of both sides of the differential pair are designed to be the same. While manufacturers of high speed digital differential devices take care to minimize asymmetries, imbalance may be produced by asymmetries in semiconductor drivers as voltage swings, rise and fall times may not be equal. In addition, variabilities in seemingly identical passive components may also contribute to imbalance. Skew is a term used to describe one of the common types of differential imbalance. Delay skew denotes effective path length differences between the sides of a differential pair. Previous work, along with experience, indicate that cable radiation due to skew imbalance of the differential digital signals in the cable can be a dominant EM1 mechanism (vs. common-mode board noise internal to the chassis exiting via the cable shields) [3],[4]. This can be the case when high speed boards are effectively designed to minimize the generation of conventional common-mode currents and placed in effective shielded enclosures. Calculations from analytical models of differential amplitudes, waveshapes, rise and fall times, and repetition rates show that the amplitude of common-mode currents generated on the shields of cables, at the fundamental frequency, are primarily dependent on delay skew of the differential signals within the cable. Dependence on the risetime of the signals is of secondary importance [3]. This paper describes experimental investigations conducted to corroborate analytical conclusions of [3]. Figure 1, taken from [3], illustrates signals on each line of the differential pair, the differential signal, and the resulting common-mode signal for a specific example of risetime and delay skew. The example in Figure 1 employs idealized signals with exponential rise and decay and no amplitude imbalance. 195

2 Line 2 Line 1 I between them at a height of approximately one meter above the floor. The entire apparatus was placed in a fully anechoic shielded chamber. The 20-to-80% risetime of the launched differential voltage signal was on the order of 0 ps. Figure 2 illustrates both the common-mode current and radiated field measurement setup. When radiated field measurements were performed, the current probe and the spectrum analyzer were removed from the shielded chamber containing the test setup. L -at,,,,,, 1 O.OEiOO 5.OE-10 1.OE-O9 1.5E-09 2.OE E-09 Time, Seconds Figure 1: Typical line-to-ground differential and common mode time domain waveforms (170 ps lo-90% risetime, 100 ps delay skew [3] EXPERIMENTALSETUP Two test boards were prepared, one with a battery-powered HP HDMP-1536A Fibre Channel transceiver source and the other with a 150 Ohm differential load with no connection to signal ground (pure differential termination). The source board was designed with selectable amounts of deliberately induced delay skew imbalance (from 0 to 754 ps in six increments by means of soldered jumpers). The bit pattern produced by the transceiver was jumper selectable and set to a continuous repetitive pattern. The fundamental frequency of this bit pattern is MHz. Both boards were outfitted with DB-9 board-mounted connectors and each was installed in a shielded enclosure with a one meter, 150 Ohm differential cable, suitable for Fibre Channel use at 1.06 Gb/s connecting the enclosed boards. The shielded enclosures were (23 cm H x 34 cm W x 75 cm D) arranged so that the one meter cable was stretched taut Figure 3 illustrates the method employed to achieve deliberately induced delay skew on the printed circuit board. Soldered jumpers were used on both sides of the differential line, to equalize skew effects from the soldered jumpers themselves. One side of the differential transmission line had a straight through connection and served as a reference for the other side of the line. The skewed side exhibited path length differences with respect to the straight through side that ranged from 0 to 4.23 inches. The printed circuit board was constructed from FR-4 material with a dielectric constant of approximately 4.0. The differential traces were formed in a microstrip configuration, e.g., with FR-4 material below the traces and air above the traces. Trace width was specified as 5 mils of I/ oz. copper. The intent was to achieve differential lines of 150 Ohm differential impedance, 75 Ohms to ground. The test cable used had two untwisted signal pair, with an internal foil shield around each pair and an outer braid over the entire bundle. The cable was intended for use in Fibre Channel-like environments, i.e., Gbis data transmission and low values of cable skew and loss. The cable was terminated on both ends with a DB-9 connectors and fully enclosed one piece metal can backshells. For each value of induced delay skew, the common-mode current on the outside of the cable shield was measured, using a Fischer F-2000 current probe, a Hewlett Packard 8593EM Spectrum Analyzer, and an HP 84471) Pre-Amplifier. In the sequence of measurements made, some were above the 1.3 GHz calibration limit of the pre-amplifier. Skewed Line, Selectable Amounts Antenna -T 3m Source Current Probe Load I Straight Through Line / Soldered Jumpers Figure 2: Experimental setup for measurement of commonmode shield currents and radiated fields from a 1 meter length cable carrying a 531 MHz repetitive lolo... signal Figure 3: Deliberately-induced delay skew was accomplished by means of soldered jumpers used to increase transmission line length on one side of a differential pair. Jumpers were utilized on both sides of the line. 196

3 Table 1: Peak common-mode current measured along the length of one-meter cable versus induced delay skew for three harmonic frequencies Table 2: Radiated electric field at a three-meter frequencies distance from a one-meter cable versus induced delay skew for three harmonic For these measurements, a calibration measurement was performed to determine the gain at these frequencies to assure accurate measurements. Measurements of common-mode current on the cable shield were performed inside a fully enclosed anechoic shielded chamber so that external ambient noise signals would be minimized. Shield current measurements were made both at the source and load ends of the cable and along the cable length to observe the maximum current. (All settings of skew and all frequencies measured produced standing waves on the cable shield. The maximum current may not have been observed at the source end.) All currents reported in this paper are peak values of current observed along the cable s length. Radiated electric fields were measured at three meters by placing the test configuration in a fully anechoic chamber and measuring the field with an EMCO 3143 BiConilog antenna, along with the spectrum analyzer and pre-amplifier from the previous test configuration. For both current and field measurements, the frequencies of 531. MHz, MHz, and MHz were chosen, representing the fundamental frequency and its second and third harmonics. Figure 5 provides an illustration of the skew behavior for values of delay skew significantly less than the risetime of the differential signal. For these values of delay skew, the peak common-mode current at the fundamental frequency, 531. MHz, is nearly proportional to delay skew, rising at approximately 0.04 db/ps of skew and radiated electric field rising at approximately db/ps of delay skew. For larger values of delay skew, the common-mode current increases at a slower rate. Neglecting the largest value of skew, a regression line 1 --C i%ield Q 3 m + Common-Modes1 RESULTS Tables 1 and 2 provide the peak current measured along the cable length and radiated electric field at a three meter distance, along with noise floor measurements for each frequency and value of delay skew. Fundamental Frequency MHz Figure 4 shows both the peak value of the fundamental frequency component of the common-mode current and the radiated electric field at a three meter distance versus delay skew. An examination of the figure shows that common-mode current and electric field increase monotonically with delay skew. 0, *II, Delay Skew (ps) Figure 4: Radiated electric field and peak commonmode current at 531. MHz on a one meter cable for various values of induced delay skew 197

4 3 m +Cawwn-Mode Current 10 5 o,t~l,,~,~t,t,t Delay Skew (ps) Figure 5: Radiated electric field and peak common-mode current at 531. MHz for values of skew less than the risetime of the differential signal through the common-mode current data points rises at nearly 9 db/decade of skew. The radiated field rises at nearly the same rate. Notice in Figure 5 that there is measurable common-mode current with no deliberately induced delay skew. This is an indication of an 40 5 inherent imbalance in the interconnect path and source circuit. Subsequent investigations of the printed circuit board showed that the individual traces in the differential pair did exhibit some voltage level imbalance, which itself can create a common-mode current. In addition, the cable is an undoubted source of skew imbalance, even though it was a cable designed for transmission of high speed differential signals, with a low skew value. Figure 6 shows the ratio of radiated electric field to peak measured common-mode shield current at 531. MHz for all values of skew examined. In linear units, for all values of delay skew, the average ratio of radiated field at 3 meters to cable shield current was approximately 3.6 pv/m per pa at 531. MHz. [5] postulates a ratio of 10 between radiated electric field in mv/m at 10 meters to current in ma. A 3 meter distance represents approximately 5.3 wavelengths at this frequency, so the antenna may be considered to be approximately in the far field. Multiplying the factor of 3.6 by 3/10 to compare with ratio in [S] yields a factor of 1.1. [51 s ratio appears to be the result of a worst-case calculation for a resonant dipole. This data represents radiation from a longer than resonant dipole (and may not represent true maximum radiation as neither the antenna nor the EUT were moved exhaustively in search of the peak). While the comparison with [5] is not close, for this type of measurement at this high frequency, this is not bad comparison. This transfer function is interesting in that it allows estimation of the maximum level of commonmode current permissible to stay within regulatory limits of radiated emissions. Higher Harmonics Frequencies other than 531. MHz were examined. Figure 7 shows the first three harmonics of the fundamental frequency commonmode currents that were measured on the cable. It should be noted that the existence of the second harmonic, 1.06 GHz, was not predicted in the idealized analytical model used in [3]. The reason for this is the apparent asymmetry in the common-mode current pulse. To avoid even ordered harmonics in a repetitive pulse, half wave symmetry must be observed, i.e., f(t) = - f(t+t/2) where, t= time, and T = period. (1) Common-Mode Shield Current, dbba Figure 1 shows clearly the presence of half-wave symmetry in the common-mode waveform. This is seen by inspection. Hence the spectrum of the waveform exhibits a lack of even numbered harmonics. However, in less ideal circumstances half-wave symmetry can easily be lost. Non-ideal semiconductor devices seldom provide the symmetry required to avoid even-ordered harmonics. As an example of this behavior, [6] provides an oscilloscope trace of a clock pulse on the parallel I/O side of the same HDMP-1536 Fibre Channel transceiver which exhibits some typical duty cycle distortion, along with an instrnment-calculated Fourier Transform spectrum that very clearly shows the presence of even-ordered harmonics. Note that device asymmetries that produce duty cycle distortion are common in these devices. Figure 6: Radiated electric field at a 3 meter distance versus peak common-mode current at 531. MHz for a one meter cable In a non-ideal high speed circuit, lack of symmetry (balance) can be due to many factors. The risetime of the signal pulse may be different than the falltime, and the duty cycle may be less than 50% (one 198

5 -531. MHz...m MHz MHz a lo 2 iii IO 100 Delay Skew (ps) 1000 m RIsetime. 264 current p5 Falltime 284 ps Duty cycle(l) 47.6% u p-p(1) "" Figure 8: Launched differential waveform showing duty cycle distortion and differing rise and fall times (0 ps induced skew. Figure 7: Peak common-mode currents at three lowest harmonic frequencies on a one meter cable half of the pulse is slightly shorter in duration than the second half of the pulse). The launched waveforms and spectra from the test board without cable attached, shown in Figures 8 and 9, illustrate this phenomenon. Asymmetries may be due to behavior of the semiconductor transceiver, the printed circuit board, and/or the cable. The measurements in Figures 8 and 9 were made with an HP 54720D digital oscilloscope. (Some risetime degradation is observed due to bandwidth limitations of the oscilloscope probes.) Clearly, in this example, neither rise and fall waveshapes are identical, nor is the duty cycle 50%. The second harmonic component of the current, MHz, exhibits a behavior that is much different than the monotonic behavior of the fundamental frequency. As discussed previously, the second harmonic is made possible by duty cycle distortion and other asymmetries. The study of the process of differencing of two asymmetric waveforms is complex and is beyond the scope of the present effort. However, the third harmonic, MHz begins to decrease after the skew reaches some intermediate value. This was puzzling until the characteristic impedance of the printed circuit board was examined. Indeed, the board was fabricated with an etch process that was not closely controlled. Under-etching produced traces with a singleended characteristic impedance very close to 50 Ohms, instead of the desired 75 Ohms (the traces were wider than 5 mils). The third harmonic behavior may be attributed to an increase in reflection coefficient as a function of transmission line electrical length. Due to layout disparities in the impedance of the board level transmission lines and the cable transmission lines, an impedance mismatch is present. For very long wavelengths and short transmission lines the reflection coefficient will approach zero and no backward traveling wave is created. As the line length and/or frequency of the signal increases, the phasor relationship of the reflection coefficient +(FT)= 8.95.dBra 531 iihz X(FT)= dbn LiHZ b = da 531 IiHZ l/ax = "5 Figure 9: Fourier spectrum of the launched differential waveform, as computed by an instrument-generated FFT begins to become important, increasing the reflected wave. This increase in the reflected wave decreases the transmitted signal onto the cable transmission line (assumed to be matched at its opposite end). At MHz, the maximum skewed trace length, the 4.73 inch trace length represents approximately The next lowest skewed trace length of 1.91 inches represents 0.26 a. Clearly, the skewed side of the differential pair is long enough to exhibit mismatched transmission line behavior, whereas the 0.5 inch straight through side probably does not, due to its short length. Hence, larger skew values produce larger common-mode currents and simultaneously larger reflections cause net common-mode current to decrease. Thus, for higher harmonics, an increase in delay skew can cause the common-mode current to decrease due to the loss of transmitted signal onto the cable. This is illustrated in the data of Figure 2 for the 31d harmonic. This is the behavior that the MHz current exhibits in Figure 7. The transmission line behavior described in the preceding paragraph is probably evident, but to a lesser degree, at the second harmonic, MHz. The 4.73 inch maximum length skewed side of the 199

6 MHz -a MHz MHz CONCLUSIONS 1. Imbalance in differential signaling produces common-mode signals on cable shields. 2. The second harmonic was present due to duty cycle distortion and rise and fall time differences inherent in the driver transceiver. 3. At the fundamental frequency of 531. MHz, common-mode current and radiated emissions increased at a rate of approximately 9 dbldecade of skew Delay Skew (ps) 1000 Figure 10: Radiated electric fields at a 3 meter distance from a one meter cable carrying differential signals with induced delay skew trace represents 0.43 a, and the 1.91 inch skewed trace represents 0.17 h. The impedance mismatch reflection, combined with the complex generation mechanism of the second harmonic creates a situation which is difficult to analyze. In retrospect, due to the unforeseen low characteristic impedance of the printed circuit board traces, common-mode current data at the harmonics above the fundamental are difficult to draw conclusions from, with the exception of the existence of the second harmonic. Figure 10 shows radiated electric fields at a three meter distance for each of the harmonic frequencies. It should be noted that no adjustment was required on antenna height to peak the reading, i.e., raising or lowering the antenna had little effect on field strength. The fundamental and third harmonics behave in a manner similar to the peak current seen in Figure 7. The second harmonic radiated field, however, does exhibit a monotonically increasing behavior, contrary to the non-monotonic behavior of the cable current. An examination of the data shown in Tables 1 and 2 shows that the measurements of radiated electric field at MHz were very close to the noise floor, whereas the current measurements were many db above the noise. Therefore, it is possible that some of the radiated field measurements, particularly the readings for the three smallest values of delay skew, may be contaminated with noise. 4. At skew values much lower than the rise time of the signal, common-mode current increased nearly linearly with delay skew. 5. For all values of delay skew examined, the average ratio of radiated electric field (at 3 meters) to cable shield current was approximately 3.6 pv/m per pa at MHz. In terms of linear units, this represents a ratio (electric field in pv/m to pa) of 3.6 at a three meter distance and a ratio of 1.1 at a ten meter distance. ACKNOWLEDGEMENTS The authors express their appreciation to their employers, colleagues, and families for the encouragement and support received during the pursuit of this work. In addition, the authors wish to thank Molex, Inc., which contributed to the success of this work by supplying the cable assembly used. REFERENCES 1. McMillan, R.J., The BYNET ~2.0 Interconnection Network, Hot Interconnects 6, August 1998, pp Paul, C.R., A Comparison of the Contributions of Common- Mode and Differential-Mode Currents in Radiated Emissions, IEEE Transactions on Electromagnetic Comuatibility, Vol. 31, No. 2, May 1989, pp Hoeft, L.O., J.L. Knighten, J.T. DiBene II, M.W. Fogg, Spectral Analysis of Common Mode Currents on Fibre Channel Ca- ble Shields due to Skew Imbalance of Differential Signals Operating at 1.06 Gb/s, IEEE International Svmnosium on Electromagnetic Comoatibilitp, Denver, CO, August 1998, pp Xu, M., S. Radu, J. Knighten, J. DiBene II, J. Drewniak, T. Hubing, and L. Hoeft, Signal Induced EM1 in Fibre Channel Cable-Connector Assemblies, IEEE International Svmoosium on Electromagnetic Comnatibilitv, Seattle, WA, August Goedbloed, Jasper, Electromagnetic Compatibility, Prentice Hall, New York, DiBene, J.T. II and J.L. Knighten, Effects of Device Variations on the EM1 Potential of High Speed Digital Integrated Circuits, IEEE International Symposium on Electromagnetic Comoatibili&, Austin, TX, August 1997, pp