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1 2224 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 9, SEPTEMBER 2009 Discriminators for Instantaneous Frequency Measurement Subsystem Based on Open-Loop Resonators Marcio F. A. de Souza, Fabio R. L. e Silva, Marcos T. de Melo, and Lauro R. G. S. L. Novo Abstract The authors have studied a new way to design an instantaneous frequency measurement subsystem (IFMS). The method is based on bandstop filters. The device uses rectangular open-loop resonators instead of delay lines. The resonance frequency is adjusted by a resonator perimeter that must be approximately a half-wavelength long, and also by coupling distance between the resonators and the main transmission line. Simulated and experimental results are presented for a 5-bit IFMS. Index Terms Bandstop filters, microstrip filters, radio spectrum management, resonator filters. I. INTRODUCTION I NSTANTANEOUS frequency measurement subsystems (IFMSs) are widely used in electronic warfare and electronic intelligence systems for the determination of unknown signals over a broad frequency band [1] [4]. The discriminators provide different time delay intervals, where the delay signals are compared with the original ones for measuring the instantaneous frequency. The frequency resolution of the instantaneous frequency measurement (IFM) depends on the length of the delays, which become, in practice, the main elements of the subsystem. The delay lines are often designed with a large number of bends in order to achieve the desired resolution. However, these bends increase the multiple reflections through the delays. A new way proposed to implement this device uses bandstop filters to obtain a signal similar to that one supplied by the interferometer. The development of the bandstop filters with wide passband and narrow rejection band [5] allows to project this subsystem with good resolution in a broadband frequency response. For comparison, the use of loop resonators instead of delay lines and power splitters, to design the discriminators, decreases the simulating time of the whole structure, as there are no more bends or sloping strips. In addition, one has more control over the resolution, as one can couple the resonators one by one and create the rejection bands. In this process, it is possible to associate the loop resonators and design a multibandstop filter. The great idea now, not found elsewhere, is not to design good filters Manuscript received February 26, 2009; revised May 28, First published August 25, 2009; current version published September 04, The authors are with the Departamento de Eletronica e Sistemas, Universidade Federal de Pernambuco (UFPE), Recife PE CEP , Brazil ( marciofbi@gmail.com; fabio.rlsilva81@gmail.com; marcos@ufpe.br; lauronovo@yahoo.com.br). Digital Object Identifier /TMTT Fig. 1. Interferometer used in IFMS. to make new interferometers. The new idea is to use simple filters to control both the resolution and the smoothness to make the digital words in a very easy way, improving the unknown signal detection and increasing the probability of interception. It is definitely very important when one needs to improve resolution for big application with many bits like satellites, radio astronomy, etc. The multibandstop filter looks promising to substitute interferometer in the IFMS. II. BASIC CONCEPTS OF THE IFMS The system is based on frequency mapping, going from an analogical signal into digital words. Any frequency value in the operating band of the system corresponds to a unique digital word. In the process, there is no need to adjust or tune any device. The signal is identified instantaneously. The frequency resolution depends on the longest delay and the number of discriminators. Let us see how the IFMS maps the incoming signal into digital words. First of all, consider a sinusoidal signal split into two parts, as shown in Fig. 1. The signals and are then described as and Due to different delays and, one has and are the signals after passing the delay and, respectively. The output is then given by the addition of (2) and (3), and after some trigonometric manipulations, that sum can be written as (1) (2) (3) (4) /$ IEEE

2 DE SOUZA et al.: DISCRIMINATORS FOR IFMS BASED ON OPEN-LOOP RESONATORS 2225 Fig. 2. Architecture of a traditional IFMS. Fig. 3. Architecture of an IFMS using bandstop filters. From (4), one can see that the frequency interval between two consecutive maxima or minima of are given by where is the delay difference between the two branches of the interferometer. Still, from (5), it is noticed that from, one gets and vice-versa. As in [1], the frequency resolution is given by A binary code can be generated if and this way, the resolution of an -bit subsystem can be rewritten as (5) (6) (7) (8) Fig. 2 shows the architecture of a traditional IFMS, where delay lines are used to implement five interferometers as discrimination channels. Each discriminator provides one bit of the output binary word that is assigned to a certain sub-band of frequency [1]. Wilkinson power dividers are used at the input and output of each interferometer [3]. The output of each discriminator is connected to a detector. The 1-bit A/D converter receives the signal from the amplifier, and attributes 0 or 1 to the output to form the digital word for each frequency sub-band. These values depend on the power level of the received signal. A limiting amplifier is used in the IFM input to control the signal gain, to increase sensitivity, and clean up the signal within the band of interest [1], [7]. The proposed IFMS based on a bandstop filter is shown in Fig. 3. The advantage of using the new architecture is that one has in each channel only multibandstop filters instead of delay lines and power splitter, as one finds in the classical IFMS. Each word is assigned to only one frequency sub-band to generate a one-step binary code. The response of each multiband- j. (b) A/D con- Fig. 4. Responses for the IFMS from Fig. 3. (a) Desired js verters output. (c) Generated code. stop filter should be like the one shown in Fig. 4(a) with discriminators 0, 1, 2, 3, and 4. Discriminator 0 provides the least significant bit (LSB) and discriminator 4 provides the most significant bit (MSB). The form of these responses is suitable to implement the 1-bit A/D converters. Here, let us attribute value 1 if the insertion loss response for the multibandstop filter is greater than 5 db, and value 0 for the opposite case. Fig. 4(b) shows the wave form of each 1-bit A/D converter output. According to this example, the waveforms at the 1-bit A/D converter outputs are shown in Fig. 4(c).

3 2226 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 9, SEPTEMBER 2009 Fig. 5. (a) Physical structure of a resonator with resonance frequency at GHz. (b) Frequency response of the resonator over a wideband. Fig. 6. Open-loop resonator. As seen in Fig. 4, this subsystem has its operating band from 2 to 4 GHz, which was divided into 32 sub-bands. Therefore, the resolution obtained was MHz. III. LSB MULTIBANDSTOP FILTER DESIGN Rectangular microstrip open-loop resonators were chosen to design every discriminator of a 5-bit IFMS. The frequency response of those resonators presents a narrow rejection band and wide passband [5] with first spurious response out of the working band. Fig. 5(a) shows the top view of a resonator with resonance frequency at GHz. One can see in Fig. 5(b) that the first spurious response occurs at GHz. Still in this section, it will be shown how this response makes possible the fabrication of a wideband discriminator. That resonator is placed near to a 50- microstrip transmission line, which was designed with aid of quasi-static analysis and quasi-tem approximation [8], [9]. Fig. 6 shows the resonance frequency adjusted by the length of the resonator, which must be approximately a half-wavelength long [8]. Additionally, there is a coupling gap given by. Moreover, the coupling distance between the resonator and the main transmission line affects this resonance frequency. This distance also affects the bandwidth of the resonator [8]. Despite the narrowband of the isolated resonators, wide rejection bands are created from coupled arrays. Fig. 7(a) presents three sketches of one, two, and three resonators, whose resonant frequencies are 2.02, 2.07, and 2.12 GHz, respectively. The linewidth for the resonators is fixed to be 0.5 mm along this paper. The ideal coupling distance between resonators is obtained varying using an electromagnetic (EM) full-wave software. Fig. 7(b) shows the frequency response obtained at an ideal coupling distance between them. These distances are chosen to Fig. 7. (a) Open-loop resonator arrays. The scale has been enhanced for a better comprehension of the devices. (b) Frequency response of one, two, and three resonators. Fig. 8. Lumped-element model for the coupling of adjacent resonators. obtain the insertion loss greater than 10 db over the rejection band and also to get this band as large as required. One notices that the coupling between nonadjacent resonators is almost zero. This happens because their resonance frequencies are not very close and the distance between them is large enough. Therefore, the insertion of a new resonator does not change the position of the others already inserted.

4 DE SOUZA et al.: DISCRIMINATORS FOR IFMS BASED ON OPEN-LOOP RESONATORS 2227 TABLE I DISTANCES Fig. 9. Comparison between lumped-element model and simulated numerical model for the coupling of resonators 1 and 2 (from Fig. 7). Fig. 10. (a) Layout of discriminator 1. (b) Frequency response of discriminator 1 and the output of the 1-bit A/D converter; 250 MHz for each rejected band. A model of two coupled resonators has been developed by the authors. A parameter sweep was defined, based on that model, in the simulation tool in order to achieve the optimum distance between adjacent resonators. Fig. 8 shows the model for the coupling of resonators 1 and 2, as shown in Fig. 7(a). The resonators coupled to the main line can be modeled by the shunt tank circuits. On the other hand, the coupling between two resonators can be modeled by coupling capacitance and inductance, and the value K denotes the mutual coupling of the inductors in the middle of the model. The minus value indicates that the current induced in the resonators have opposite directions. A comparison between the results of a simulated numerical model and a lumped-element model is shown in Fig. 9. Due to the narrowband nature of the open-loop resonators, it is not possible to define a unique value for the lumped elements of the model for the entire bandwidth. Corrections of that should be performed as the band moves up or down in the frequency spectrum. As one increases the number of resonators, the lumped-element model becomes difficult to be implemented. A qualitative analysis of the bandstop filters used here may be carried out Fig. 11. Bandstop filters for implementation of the: (a) discriminator 4 (MSB), (b) discriminator 3, (c) discriminator 2, (d) discriminator 1, and (e) discriminator 0 (LSB). comparing it with a unified formulation for an -coupled resonator filter, regardless of whether the coupling are magnetic or electric or even the combination of both. The coupling matrix, external quality factor, and scattering parameters (functions for the filter design) are related by the following equations [8]: (9) (10) (11) (12) where is an matrix with all entries zero, except for and, is the unit or identity matrix, and is the so-called general coupling matrix, which is an reciprocal matrix. denotes the scaled external quality factors. In a filter design, after establishing

5 2228 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 9, SEPTEMBER 2009 Fig. 12. Frequency response of the: (a) discriminator 4 (MSB), (b) discriminator 3, (c) discriminator 2, (d) discriminator 1, and (e) discriminator 0 (LSB). TABLE II LENGTH IN MILLIMETERS OF DISCRIMINATORS 4, 3, AND 2 the functions and, one can find the coupling matrix and the external quality factors using an appropriated synthesis procedure. The physical dimensions of the filter for fabrication are found determining the required coupling matrix and then establishing the relationship between the value of every required coupling coefficient and the physical structure of coupled resonators. For a long structure like that, the modeling is not so simple; it is a necessary considerable amount of time for adjusting suitable transfer functions with quality factors and coupling coefficients, and obtaining large matrices for computational calculations, which is out of the scope of this work. Using the previous concepts, one desires to explain the design of discriminator 1.

6 DE SOUZA et al.: DISCRIMINATORS FOR IFMS BASED ON OPEN-LOOP RESONATORS 2229 As the desired insertion loss of discriminator 1 is shown in Fig. 4(a), there must be four rejection bands, where the first one is from to GHz, regarding the chosen operating band. The resonators are arranged one by one. Fig. 10(a) shows this discriminator with its numbered resonators. The device is designed on an RT substrate of relative dielectric constant and thickness mm. The 50- transmission linewidth is 1.2 mm. The gap of every resonator and the distance between the main transmission line and the resonators are kept 0.1 mm for whole structure. Table I shows the coupling distances between the resonators for this device. In Fig. 10(a), one still sees four groups of resonators, whose frequency responses and A/D converter outputs are shown in Fig. 10(b). Looking carefully at their correlation, Group 1 gives the rejection band over 2 GHz, Group 2 gives the rejection band over 2.5 GHz, and so on. Fig. 10(b) presents the simulated results of discriminator 1, which agree with the results shown in Fig. 4. One can see the insertion-loss level is greater than 10 db over all rejection bands, and is less than 5 db over the passbands. The output A/D converter should generate level 0 for db and level 1 for db. Concerning all the involved, the dimensions of this discriminator are 3-cm wide and 15-cm long. Following the same procedure, the others discriminators are projected, where new resonators configurations will give new desired rejection bands. IV. RESULTS AND DISCUSSION Fig. 11(a) (e) presents all the projected IFMS discriminators from Fig. 3 having from 23 to 25 resonators. The number of resonators depends on the desired rejection bands. Following the same principle, each group gives only one rejection band so that discriminators with eight groups have eight rejection bands, as shown in Fig. 11(e). The others, without any specified group, have only one, as shown in Fig. 11(a) and (b). All discriminators were designed, simulated, fabricated, and measured (all the dimensions of the resonators are outlined in the Appendix). Initially, they were projected to operate in the range of 2 4 GHz. However, from the measured results was seen a frequency shift, moving the range to lower values, so that, in practice, one had a final range of GHz. Fig. 12 shows that the simulated and measured results of the five discriminators are in reasonable agreement with each other. One can notice that the filters of Fig. 11(a) (e) have the behavior desired in Fig. 4(a) for an IFMS of 5 bits. In Fig. 12(e) (the LSB discriminator), there is a little difference between the simulated and measured results over the end of the band. For a symmetrical doubly loaded resonator, one has (as in [8]) (13) TABLE III LENGTH IN MILLIMETERS OF DISCRIMINATORS 1 AND 0 where is the external quality factor for a single loaded resonator, and should be seen as the external conductance attached to the lossless LC resonator. Comparing the simulation with the experimental results, a qualitative analysis may be carried out using the above equation. In practice, with cable connections, may change, causing a frequency shift. The losses in all the discriminators are far lower than the ones presented in [3], which concerns at the level of 9dB. Despite this fact, excessive losses in each discriminator can be compensated by inserting an amplifier at the input of the detector. Furthermore, the loss also tends to decrease if the discriminators are fabricated in a superconductive version, offering high bandwidth with low losses [10]. V. CONCLUSION Multibandstop filter were designed, simulated, and measured over a frequency range of 2 GHz. The results show that the use of loop resonators to design the discriminators, instead of delay lines and power splitters, make the simulation and fabrication easier, as there are no more bends or sloping strips. In addition, one has more control over the resolution, as one can couple the resonators one by one and create the rejection bands. In this process, the association of loop resonators was used to design multibandstop filters. In light of the above, the use of multibandstop looks promising as far as planar IFMS applications are concerned.

7 2230 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 9, SEPTEMBER 2009 TABLE IV DISTANCE BETWEEN RESONATORS i AND j IN MILLIMETERS See Tables II IV. APPENDIX REFERENCES [1] G. C. Liang, C. F. Shien, R. S. Withers, B. F. Cole, M. A. Johansson, and L. P. Suppan, Superconductive digital instantaneous frequency measurement subsystem, IEEE Trans. Microw. Theory Tech., vol. 41, no. 12, pp , Dec [2] H. Gruchala and M. Czyzewski, The instantaneous frequency measurement receiver in the complex electromagnetic environment, in 15th Int. Microw. Radar Wireless Commun. Conf., May 2004, vol. 1, 17 19, pp [3] M. Biehl et al., A 4 bit instantaneous frequency meter at 10 GHz with coplanar YBCO delay line, IEEE Trans. Appl. Superconduct., vol. 4, no. 2, pp , Jun [4] M. T. de Melo, M. J. Lancaster, and J. S. Hong, Coplanar strips interdigital delay line for instantaneous frequency measurement systems, in IEE Dig., Nov. 1996, pp. 1/1 1/4, Art. ID. 1996/226. [5] A. Görür, Bandstop filter with a wider upper passband using microstrip open-loop resonator, in Proc. Asia Pacific Microw. Conf., Taipei, Taiwan, 2001, pp [6] R. Bauman, Digital instantaneous frequency measurement for EW receivers, Microw. J., pp , Feb [7] G. C. Liang, C. F. Shien, R. S. Withers, B. F. Cole, and M. A. Johansson, Space-qualified superconducting digital instantaneous frequency-measurement subsystem, IEEE Trans. Microw. Theory Tech., vol. 44, no. 7, pp , Jul [8] J. S. Hong and M. J. Lancaster, Microstrip Filters for RF/Microwave Applications. New York: Wiley, [9] K. C. Gupta, G. Ramesh, and I. J. Bahal, Microstrip Lines and Slotlines, 2nd ed. Norwood, MA: Artech House, [10] G. C. Liang et al., Superconductive digital instantaneous frequency measurement subsystem, in IEEE MTT-S Int. Microw. Symp. Dig., 1993, pp Marcio F. A. de Souza was born in Recife, Brazil, in He received the B.S. and M.Sc. degrees in electrical engineering from the Universidade Federal de Pernambuco (UFPE), Recife, PE, Brazil, in 2006 and 2008, respectively. His research interests include design and fabrication of microstrip structures in microwave frequencies such as resonators and power dividers and mainly filter and IFM discriminators. Fabio R. L. e Silva was born in Recife, Brazil, in He received the B.S. degree in electrical engineering from the Universidade Federal de Pernambuco (UFPE), Recife, PE, Brazil, in 2007, and is currently working toward the M.Sc. degree in electrical engineering at UFPE. His research areas are focused on planar microwave devices such as antennas for RF identification (RFID) systems, filters, power dividers, etc. Mr. e Silva has been a reviewer for the IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS since 2008.

8 DE SOUZA et al.: DISCRIMINATORS FOR IFMS BASED ON OPEN-LOOP RESONATORS 2231 Marcos T. de Melo received the B.S. and M.Sc. degrees in physics from the Universidade Federal de Pernambuco (UFPE), Recife, PE, Brazil, in 1983 and 1992, respectively, and the Ph.D. degree from Birmingham University, Edgbaston, Birmingham, U.K., in His masters thesis concerned microwave absorption on superconducting samples. His doctoral thesis concerned high-temperature superconducting devices. In 1999, he joined the Photonics Group, Department of Electronics and Systems, UFPE. His current research interests include design and fabrication of coplanar structures in microwave frequencies such as resonators, power divider, filters, delay lines, IFM systems, superconducting transmission lines and also the measurement of dielectric properties of novel materials for microwave applications. Lauro R. G. S. L. Novo was born in Jaboatao dos Guararapes, Brazil, in He received the B.S. degree in electrical engineering from the Universidade Federal de Pernambuco (UFPE), Recife, PE, Brazil in 2007, and is currently working toward the M.Sc. degree in electrical engineering at UFPE. He is also involved with parametric characterization of power transmission lines. His research areas are focused on microwave devices, especially planar structures such as filters, power dividers, instantaneous frequency measurement systems (IFMSs), etc.