UNIVERSITY OF CALGARY. In-home PLC to DSL Interference Characterization and Mitigation. Khaled Ali A THESIS

Transcription

1 UNIVERSITY OF CALGARY In-home PLC to DSL Interference Characterization and Mitigation by Khaled Ali A THESIS SUBMITTED TO THE FACULTY OF GRADUATE STUDIES IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE DEGREE OF DOCTOR OF PHILOSOPHY DEPARTMENT OF ELECTRICAL AND COMPUTER ENGINEERING CALGARY, ALBERTA January, 2015 c Khaled Ali 2015

2 Abstract It is often advocated that a solution to the problem of mutual interference between digital subscriber line (DSL) networks and in-home power line communications (PLC) networks is to prevent the PLC networks from utilizing the DSL spectrum. However, this solution will render PLC networks inoperable with the introduction of wide-band DSL technologies like G.fast. As an alternative solution, this thesis proposes to use the common mode (CM) signal, which contains information about the electromagnetic interference (EMI), to estimate and subtract the differential mode (DM) PLC-to-DSL interference from the DM DSL signal. Since the PLC-to-DSL interference environment within a residential setting has neither been characterized via measurements nor by a model, a measurement campaign is conducted. A set of 480 measurements are collected, within two residential test-sites, to characterize the PLC-to-DSL interference environment for two DSL modem installation scenarios. This campaign shows that the PLC-to-DSL interference channels are frequency selective and vary significantly from room to room within the house. Two interference mitigation solutions are proposed in this thesis. The first solution relies on scheduling the PLC channel access; while, the second solution pre-multiplies the PLC symbol by the inverse of the DM cross-coupling channel before transmission. Both solutions utilize adaptive frequency domain interference cancellers (FDICs) that are insensitive to the non-stationarity of the PLC channel and the frequency selectivity of the coupling channels. The performances of the proposed solutions are evaluated, and their effectiveness in mitigating the PLC-to-DSL wide-band EMI is demonstrated using analysis that incorporates the measured PLC-to-DSL coupling channels. i

3 Acknowledgements First and foremost, I would like to sincerely thank my supervisor, Dr. Geoffrey Messier, for his patience, guidance, and support. Dr. Messier taught me how to conduct high calibre research that has practical value. Dr. Messier s commitment to excellence and his ability to advice, guide, and encourage his students to reach their potential is admirable. Working under his supervision made me a better researcher and significantly improved my communication skills. Also, I would like to thank my co-supervisor, Dr. Stephen Lai, for his advice and support. I have been fortunate to have a co-supervisor who cared about my work and gave me valuable feedback. I also thank the Lai and Messier families for welcoming me into their homes to perform channel measurements. I would like to thank my thesis committee, Dr. John Nielsen, Dr. Abu Sesay, Dr. Kyle O Keefe, and Dr. Michael McGuire for their valuable time and advice. Thanks to Lincoln Zhao, Mohamed Gaafar, Michael Wasson and all my colleagues at the FISH Lab for all their help and support. Many thanks to Mohamed Ammar Al Masri for the various stimulating discussions and coffee runs. Thanks to Ms. Ella Lok and the staff at the ECE department. Many thanks to Dr. Rainer Iraschko and the staff at TRTech. Words cannot express my gratitude to my parents, who have been very supportive through out my life. I would have not been able to make it this far, if it were not for their kindness, patience, forgiveness, and advice. In addition, I would like to thank my brothers, Mohamed and Sameh, for their understanding and continuous support. Finally, special thanks to my wife, Noha, and my father-in-law and mother-in law. This work was financially supported by TRTech and the Natural Sciences and Engineering Research Council (NSERC) of Canada. ii

4 Dedication To my loving parents: Mustafa and Magda. iii

5 Table of Contents Abstract i Acknowledgements ii Dedication iii Table of Contents iv List of Tables vii List of Figures viii List of Symbols x 1 INTRODUCTION Broadband Residential Internet Access Technology HFC Broadband Access DSL Broadband Access PLC Broadband Access PLC In-Home Networking DSL and PLC Coexistence Related Work on DSL and PLC Coexistence DM PLC-to-DSL Channel Measurements DSL Electromagnetic Interference Mitigation Solutions Narrow-band EMI Mitigation Solutions Wide-band EMI Mitigation Solutions Thesis Contribution Cross-Coupling Channel Characterization Interference Cancellation Thesis Outline THE DSL and PLC TECHNOLOGIES DSL Technology The Physical Network Fiber to the Exchange Fiber to the Cabinet Fiber to the Premise The DSL Families Basic Rate Interface The High Bit-rate DSL Family The Asymmetric DSL Family The Very High Speed DSL Family Fast Access to Subscriber Terminal DSL Families Comparison DSL Signalling and Frame Structure Discrete Multi-Tone Modulation DSL Signalling DSL Frame Structure The DSL Interference Environment Intrinsic Interference iv

6 Extrinsic Interference PLC Technology PLC In-home Network PLC Signalling and Channel Access PLC Signalling and Modulation MAC for Broadband PLC PLC Noise Environment CROSS-COUPLING CHANNEL MEASUREMENTS DSL Modem Installation Scenarios DSL and PLC Interference Environment Measurement Methodology Measurement Hardware Calibration Measurement Setup Measurement Campaign Test-Sites Case Study A Case Study B Results Cross-Coupling Channel Frequency Responses Stationarity of Cross-Coupling Channels Effect of Spatial Separation INTERFERENCE SYSTEM MODEL Current System Model Effect of Mutual DSL and PLC Interference on Bit Rates Effect of PLC Interference on DSL Bit Rates Effect of DSL Interference on PLC Bit Rates INTERFERENCE MITIGATION SOLUTIONS Modified System Model Proposed System Model Variations in the DM to CM Estimated Ratio Integration of the FDIC Interference Mitigation Block Diagram Scheduling-Based Interference Mitigation Solution Medium Access Cancellation Algorithm C-DSL Training C-PLC Training Performance Evaluation Mean Square Error Analysis Improvement in Bit Rate Pre-Distortion-Based Interference Mitigation Solution PLC Symbol Pre-distortion Cancellation Algorithm C-DSL Training v

7 5.4.3 Performance Analysis Training Phase Transmission Phase Comparison with Spectral Management Solutions CONCLUSION Measurement Campaign Findings Interference Mitigation Solutions Recommendation for Future Research Interference Channel Characterization PLC-to-DSL Cross-Coupling Channels DM and CM DSL Direct Channels Interference Cancellation Bibliography vi

8 List of Tables 2.1 Comparison of DSL Families Average coherence bandwidth in khz Latency: Scheduling-based solution Training phase available bit rates Transmission phase available bit rates Latency: Scheduling-based versus pre-distortion-based solutions vii

9 List of Figures and Illustrations 2.1 Conventional DSL network Sinc functions at various sub-carrier indexes DMT transmitter block diagram DMT receiver block diagram DM and CM signal at the receiver DSL super-frame structure Near end crosstalk among twisted pairs Far end crosstalk among twisted pairs Electrical wiring in north America [1] In-home PLC network Simplified PLC coupling circuit PLC frame structure for a 3-user PLC network Desk Modem Scenario Entry Point Scenario Co-located DSL and PLC networks interference environment North Hills 0320BF Balun Northern Microdesign PLC Coupler Insertion loss of the balun and the PLC coupler before calibration Modified Balun Modified PLC Coupler Right to Left: Open, Short, and Load (100 Ω) Through calibration Setup for PLC-to-DSL coupling GPIB controlled measurement setup Case Study A: test-site floor plan Case Study B: test-site floor plan Coupling in Differential Mode Coupling in Common Mode Common mode to differential mode transfer function (C2DTF) Variation in DM cross-coupling channels from one power outlet to another Entry Point Scenario coupling gain probability density function Desk Modem Scenario coupling gain probability density function Mean DM and CM PLC-to-DSL coupling Desk Modem Scenario.: DM PLC-to-DSL coupling for Room A Plug 1 over measurement interval Entry Point Scenario.: DM PLC-to-DSL coupling for Room A Plug 1 over measurement interval Desk Modem Scenario: Room A Plug 1 DM vs. CM PLC-to-DSL coupling Entry Point Scenario: Room A Plug 1 DM vs. CM PLC-to-DSL coupling. 71 viii

10 3.26 Effect of relative distance between DSL and PLC modems on PLC-to- DSL coupling Effect of relative distance between DSL and PLC modems on crosscoupling channels coherence bandwidth Current system model Received DSL Signal versus PLC interference US and DS frequencies for band plan 998E30 [2] Available DSL bit rates Degradation in DSL bit rates Received PLC Signal versus DS DSL interference Received PLC Signal versus US DSL interference Available PLC bit rates Degradation in PLC bit rates Proposed system model Interference coupling on twisted-pair Integration of the FDIC into a DMT transceiver Interference cancelling scheme block diagram Desk Modem Scenario: MSE of the proposed scheme versus the relative distance between the DSL and PLC modems Entry Point Scenario: MSE of the proposed scheme versus the relative distance between the DSL and PLC modems Desk Modem Scenario: Achieved improvement in bit rates versus the Euclidean distance between the DSL and PLC modems Entry Point Scenario: Achieved improvement in bit rates versus the Euclidean distance between the DSL and PLC modems Achieved total DSL bit rates for DS transmission, for various DSL cable run length Desk Modem Scenario: Achieved bit rates for DSL-DS Entry Point Scenario: Achieved bit rates for DSL-DS Desk Modem Scenario: Achieved bit rates for PLC Entry Point Scenario: Achieved bit rates for PLC Desk Modem Scenario: Achieved bit rates for DSL-US Entry Point Scenario: Achieved bit rates for DSL-US ix

11 List of Symbols and Abbreviations η c,dsl DSL CM AWGN matrix η c,plc PLC CM AWGN matrix η d,dsl DSL DM AWGN matrix η d,plc DSL DM AWGN matrix γ 0 SNR in presence of AWGN only γ F SINR after utilizing C-DSL γ W SINR after utilizing Weiner filter γ W/O SINR before interference mitigation φ MSE of FDIC φ F MSE of C-DSL φ Tr MSE of C-DSL during training period of the pre-distortion solution φ Tx MSE of C-DSL during transmission period of the pre-distortion solution φ W MSE of Weiner filter f Sub-carrier spacing f DSL DSL sub-carrier spacing f PLC PLC sub-carrier spacing Γ SINR gap x

12 γ(i) λ SINR for frequency bin i Normalization factor v d,dsl Estimated of DM PLC interference matrix v d,plc Estimated of DM DSL interference matrix C DSL DSL FDIC coefficient matrix C W Wiener filter coefficient matrix C PLC PLC FDIC coefficient matrix H DSL Direct DSL channel frequency response matrix H PLC Direct PLC channel frequency response matrix H f DSL c,dtx PRx CM interference channel frequency response matrix between the DSL receiver and a PLC transmitter, sampled at integer multiples of f DSL H f DSL c,ptx DRx CM interference channel frequency response matrix between a PLC transmitter and the DSL receiver, sampled at integer multiples of f DSL H f DSL d,dtx PRx DM interference channel frequency response matrix between the DSL receiver and a PLC transmitter, sampled at integer multiples of f DSL H f DSL d,ptx DRx DM interference channel frequency response matrix between a PLC transmitter and the DSL receiver, sampled at integer multiples of f DSL H f PLC d,ptx DRx DM interference channel frequency response matrix between a PLC transmitter and the DSL receiver, sampled at integer multiples of f PLC xi

13 q Transmitted PLC signal matrix r c,dsl Received CM DSL signal matrix r c,plc Received CM PLC signal matrix r d,dsl Received DM DSL signal matrix r d,plc Received DM PLC signal matrix u Generated PLC symbol v c,dsl CM DSL interference matrix v c,plc CM PLC interference matrix v d,dsl DM DSL interference matrix v d,plc DM PLC interference matrix x y z Transmitted DSL signal matrix Desired DSL signal matrix Desired PLC signal matrix (...) Average of a variable ε D DSL maximum transmission power ε P PLC maximum transmission power ŷ ẑ (...) Estimate of desired DSL signal matrix Estimate of desired PLC signal matrix Estimate of a variable xii

14 b(i) Number of bits for frequency bin i b max Maximum number of bits allowed per frequency bin M N Number of DSL sub-carriers Number of PLC sub-carriers N 0,c PSD of CM AWGN N 0,d PSD of DM AWGN N o PSD of AWGN R Total number of bits R 0 Total bit rates in presence of AWGN only R Bound Total number of bits in presence of background noise R F Total bit rates after utilizing C-DSL R Interference Total number of bits in presence of interference plus background noise R W/O Total bit rates before interference cancellation R W Total bit rates after utilizing Weiner filter t i Start time of Slot i T P PLC frame duration T SF DSL super-frame duration T S DSL frame duration (...) Complex conjugate of a variable xiii

15 E[...] Loss % a H(i, j) x(i) A/D ADSL Expectation of a variable Bit rates loss percentage Normalization factor i th row and j th column of the matrix H i th row of the vector x Digital to analog Asymmetric DSL ADSL2 Asymmetric DSL 2 ADSL2+ AM AMR AWGN BNC BRI C-DSL C-PLC C2DTF CB CM Asymmetric DSL 2 plus Amplitude modulation Automatic meter reading Additive white Gaussian noise Baby N Connector Basic rate interface DSL FDIC PLC FDIC CM to DM transfer function Coheren Bandwidth Common mode xiv

16 D/A DFT DM DMT DOCSIS DS DSL EMI FDD FDIC FFT FT FTTB FTTC FTTCab FTTDp FTTEx FTTH FTTP Digital to analog Discrete Fourier Transform Differential mode Discrete multi-tone Data over cable service Interface specification Downstream Digital subscriber line Electromagnetic interference Frequency division duplexing Frequency domain interference canceller Fast Fourier transform Fourier transform Fiber to the building Fiber to the curb Fiber to the cabinet Fiber to the distribution point Fiber to the exchange Fiber to the home Fiber to the premise xv

17 G.fast GPIB HDSL HFC HFTP IDFT IFFT ISP ITU-T Fast access to subscriber terminals General purpose interface bus High bit-rate DSL Hybrid fibre-coaxial cable Hybrid fiber twisted pair Inverse discrete Fourier Transform Inverse fast Fourier Transform Internet service provider International telecommunication union-telecommunication standardization sector LAN LPF OFDM ON PAM PDF PLC PLC-DC PSD PSTN Local area network Low pass filter Orthogonal frequency division multiplexing Optical node Pulse amplitude modulation Probability density function Power line communications PLC domain controller Power spectral density Public service telephone network xvi

18 QAM QAM RJ SHDSL SMA TB TDD US VDSL Quadrature amplitude modulated Quadrature amplitude modulation Registered jack Single-pair high-speed DSL Sub-miniature version A Terminal block Time division duplexing Upstream Very high speed DSL VDSL2 Very high speed DSL 2 VNA xtalk Vector network analyzer Crosstalk xvii

19 Chapter 1 INTRODUCTION Recent advances in power line communications (PLC) have made it popular for inhome networking. This makes PLC an increasingly relevant source of interference for digital subscriber line (DSL) networks within the home environment. This thesis presents two measurement case studies that characterize the PLC-to-DSL coupling channels, within a residential setting. In addition, this thesis proposes two interference mitigation solutions that enhance the coexistence of in-home PLC and DSL networks. The rest of this chapter is organized as follows. Broadband access technology for residential Internet is introduced in Section 1.1, while home networking solutions via PLC technology is discussed in Section 1.2. The coexistence environment between in-home PLC networks and DSL broadband access networks are discussed in Section 1.3, while background information on topics relevant to the PLC and DSL interference characterization and mitigation are discussed in Section 1.4. The thesis contributions are highlighted in Section 1.5, while the outline of the thesis is presented in Section Broadband Residential Internet Access Technology Broadband access technology is an integral part of the world wide communication network. The term broadband refers to any access technology that is always-on and can support multiple services at high data rates. Residential broadband access has become a pillar of the world culture. Throughout the developed world, and in some developing countries, access to the Internet is considered one of the household essentials due to the various services that can be provided through it, along with the 1

20 reduction in cost of having it. The ultimate in broadband access technology, in terms of data rates and reliability, is optical fiber. However, due to the cost of installation and replacement of existing access technologies, creating a world-wide communications network via optical fiber is not feasible. Currently, hybrid optical fiber connections, where the link to any particular subscriber runs partly on optical fiber cables and then partly on copperbased wires, are utilized to form hybrid networks that serve as the backbone of the world wide communications network. Residential broadband access technologies can be classified into two categories: wired (or fixed line) and wireless broadband access technologies. Fixed line broadband access technologies utilize a physical connection between the service provider and the customer; thus, a physical wired network has to be in place before the nodes in the network can communicate. Wireless broadband access, on the other hand, utilize air as the transmission medium. Wireless broadband solutions provide, along with freedom of mobility, instantaneous wide area coverage that is especially beneficial for remote areas where the infrastructure for fixed line technologies do not reach. However, wired broadband access technologies have several advantages over the wireless technologies such as higher data rates and better reliability. The most commonly used copper-based wired broadband access technologies are digital subscriber lines (DSL) and hybrid fiber-coaxial cable (HFC). DSL utilizes the copper twisted-pairs of the public service telephone network (PSTN) as the transmission medium, while HFC utilizes the coaxial cables used by the digital cable television (CATV) network as the transmission medium. Both DSL and cable modems technologies incorporate hybrid fibre optic connections, where portions of the twisted-pairs or the coaxial cables within the networks are replaced with fiber optic cables. Another copper-based fixed line broadband access technology is power line communications 2

21 (PLC). PLC utilizes the power lines that are used to carry electricity to subscriber houses as a transmission medium. The aforementioned copper-based fixed line broadband access technologies are discussed in Sections to HFC Broadband Access The cable television (CATV) network was created for broadband unidirectional transmission. Various optical nodes (ONs) are connected to the head unit via fiber cables, creating HFC connections. Out of each ON, branches of shared coaxial copper cable connect various customers. Thus a tree is formed, where the root of the tree is the head-end [3]. Note that up to 500 customers can be serviced by a single ON. To enable broadband access to the Internet, the CATV network is upgraded to support bidirectional traffic. HFC relies on dividing the bandwidth of the shared coaxial cable into non overlapping channels, where one or more of these channels are dedicated for upstream transmission while the rest of the channels are reserved for downstream transmission and cable TV. The most recent version of the cable modems protocol Data Over Cable Service Interface Specification (DOCSIS), which is the most widely used protocol for HFC broadband access, provides data rates of up to 400 Mbits/s [3]. However, the major drawback of HFC broadband access is that the cable TV network is a shared tree network, and the bandwidth per customer is limited. The available bandwidth per customer is dependent on the number of active users and poses a significant challenge for upstream data rates when a large number of users are active. In addition, due to the structure of the tree, security and scalability are always a challenge DSL Broadband Access DSL technology utilizes dedicated link for each customer, where data is transmitted over the public service telephone network (PSTN). A PSTN utilizes twisted-pairs 3

22 of copper wires that were originally designed to serve as a medium for transmitting speech signals. Since human speech is in the range of 300 Hz to 3400 Hz, higher frequencies can be used for transmitting data over existing PSTNs [4]. The ubiquity of the PSTN has motivated new DSL technologies that increase throughput using a combination of advanced communications techniques and replacing the twisted-pairs in portions of the PSTN by fibre optic cables [5]. These hybrid connections shorten the distance the DSL signal has to travel over passive copper twisted-pairs between the central office and the end users, and results in a wider bandwidth channel. An advanced standardized DSL technology is the very high bit rate digital subscriber line 2 (VDSL2). VDSL2 utilizes up to 30 MHz bandwidth [2] and vectoring [6] to achieve transmission rates up to or exceeding 100 Mbits/s. Currently, a standard is being developed for a new DSL technology, called G.fast (fast access to subscriber terminal) [7], that has the potential of achieving data rates of 1 Gbits/s, over a spectrum that spans 106 or 212 MHz [8]. These fast data rates being dedicated to each customer, unlike HFC, have made DSL a popular choice among end users for residential broadband access. The DSL technology is discussed in more detail in Section PLC Broadband Access Extensive research on utilizing PLC networks to deliver Internet to customer houses has been performed. However, one of the main obstacles that prevented utilizing power lines by Internet service providers (ISPs) is the need for a repeater at each transformer, since data signal cannot pass through the transformers [3]. Installing repeaters at each transformer, especially in North America where each transformer serves only few houses, was proved costly. This high cost and the presence of other telecommunication infrastructure prevented the concept of providing Internet over power lines to materialize. 4

23 1.2 PLC In-Home Networking Utilizing legacy wires, such as power lines, telephone cables, and coaxial cables, to distribute data within the home environment has no rewiring requirement. Various standards have been developed for the three aforementioned legacy wires to be used for residential data distribution. Among these standards, the international telecommunication union-telecommunication standardization sector (ITU-T) home networking standard G.hn [9] has specifications for each of the three legacy wires networks, along with multi-domain specifications. For the multi-domain specifications, two or more of the legacy wires networks can be utilized simultaneously to deliver data. Among the three legacy wires, power lines are the most extensively used in houses. This coupled with the recent advances in the area of networking within the home using legacy wires make PLC networks increasingly common within the home. The power lines can be utilized to form a network that transform the house into a smart home, where appliances within the house are connected. Creating an in-home local area network (LAN) via PLC technology has many advantages over other in-home networking solutions such as Ethernet and wireless. The first advantage is the low implementation cost which is due to the existence of a pre-installed power line network in each house. Thus, no new wiring or physical installation is required, which is not the case with Ethernet LANs. A second advantage that in-home PLC LANs has over Ethernet LANs is the presence of multiple access nodes (in form of electrical outlets), which are spread throughout the house. Currently, the average cost of a PLC network card is the same as the cost of a wireless network card; however, as in-home networking via PLC gains popularity with end users, mass production of PLC networks cards will cost about half the cost of the wireless network card because PLC network cards do not require an RF component [10]. Additionally, data rates over PLC networks can reach up to 200 Mbits/s, with 5

24 the ability of integrating multiple functions over the same network. For instance, automatic meter reading (AMR), home automation, and triple play services (Internet, television, and telephone services) can be integrated over an in-home PLC network simultaneously [11]. All the aforementioned advantages make in-home PLC networks a suitable choice for in-home networking [3] and, in cases where one medium is not sufficient, is a cost-effective complement to other in-home technologies [12]. Further detail on the PLC technology is presented in Section DSL and PLC Coexistence In-home PLC networks operate over the same spectrum as DSL networks. This increases the likelihood of crosstalk between PLC and DSL communications systems. For instance, two home networks that operate at the same frequency range, one over copper twisted-pairs (138 khz - 30 MHz [2]) and the other over power lines (1.8 MHz - 30 MHz [9]), would interfere with each other. The DSL and PLC interference environment is discussed in more detail in Section 3.2. Communication standards such as ITU G.hn [9], have been developed with mechanisms that prevent any interference between various systems within the home environment. However, the interference from PLC-to-DSL is usually prevented by forcing the PLC network to notch frequencies that affect the DSL signal. While this is a viable solution, denying PLC networks access to the DSL spectrum will render in-home PLC networks inoperable especially if VDSL2 or G.fast technology is employed. Further detail on mitigating the PLC-to-DSL interference via spectral notching is discussed in Section Other PLC-to-DSL interference reduction solutions, such as reducing the PLC transmit power (which is known as spectral management) or increasing the distance 6

25 between the DSL and PLC modems (which is referred to as spatial separation), have been proposed in the literature. As will be discussed in Section , spectral management degrades the performance of the PLC network, and with the increase in the usable DSL bandwidth, spectral management solutions will render the PLC networks inoperable. From the measurement case studies presented in this thesis and in [13, 14], it is shown that the interference levels between the PLC and DSL modems do not depend on the distance separating the modems. Thus, spatial separation, which is discussed in more detail in Section , does not mitigate the PLC-to- DSL interference. The goal of this thesis is to enhance the co-existence environment between DSL and PLC networks in a residential environment. The objective is to mitigate the PLCto-DSL interference without hindering the performance of the in-home PLC network. Ultimately, this will lead to the increase of data transmission rates in both DSL and PLC networks. Complementary signals are inserted onto each of the wires of the twisted-pair by the transmitter. At the receiver, the difference between the complementary signals is the differential mode (DM) signal, while the common mode (CM) signal is the arithmetic mean of the complementary signals. The DM signal contains the desired signal, while the CM signal is mainly composed of the interference. Further detail on the DM and CM signalling in DSL systems is discussed in Section The main hypothesis of this thesis is that the common mode signal contains information about the differential mode interference. By utilizing the common mode signal, the differential mode interference can be estimated and subtracted from the differential mode signal. Removing the differential mode interference increases transmission rates in DSL networks without limiting the capacity of the in-home PLC network. The common mode and differential mode signalling are discussed in more 7

26 detail in Section 2.1.3, while estimating the DM interference from the CM signal is discussed in Chapter 5. A thorough literature survey indicates that the PLC-to-DSL cross-coupling channel measurements in a residential setting that included both DM and CM reception methods have never been performed [13]. Only DM PLC-to-DSL channel measurements in lab settings have been performed in the literature; further detail on work done to date with regards to PLC-to-DSL cross-coupling channel measurements can be found in Section Characterizing the PLC-to-DSL cross-coupling channels is central to the main hypothesis and the interference mitigation solutions proposed in this thesis. Thus, a measurement campaign that studies the cross-coupling channels between DSL and in-home PLC networks has to be performed. The focus of this thesis is divided into two areas. The first focus area is PLC-to- DSL cross-coupling channel characterization in a residential setting. A measurement campaign that studies the PLC-to-DSL interference environment within residential test sites is performed. The second focus area is interference cancellation. Two interference mitigation solutions based on adaptive filter theory are proposed. The effectiveness of the proposed solutions in mitigating wide-band EMI from an in-home PLC network on to the DSL system, based on field measurements, is evaluated. 1.4 Related Work on DSL and PLC Coexistence In this section, work related to the two focus areas of the thesis is discussed. Related work to the PLC-to-DSL coupling channel measurements is presented in Section 1.4.1, while work related to the PLC-to-DSL interference mitigation is given in Section

27 1.4.1 DM PLC-to-DSL Channel Measurements Measurement case studies that investigate the differential mode PLC-to-DSL coupling channel in a laboratory environment were performed in [15, 16, 17, 18]. These studies investigated the effects of various factors, such as the distance separating the PLC and DSL cables and the shared length between the PLC and DSL cables, on the PLCto-DSL coupling over the VDSL2 spectrum. The main finding of these studies is the potential for significant crosstalk between PLC and DSL systems. However, these studies do not characterize the PLC-to-DSL coupling channel in an actual residence, nor do they consider the CM PLC-to-DSL coupling channel. The coupling between PLC and DSL systems over the frequency range of 138 khz to MHz (band plan 998ADE17 ) was measured in [19]. The objective of this study was to investigate the quality of service for Internet protocol television (IPTV). Two case studies, [20] and [21], investigated the effect of PLC on the throughput of a VDSL modem. The DM PLC-to-DSL coupling channels were neither measured in [20] nor in [21]. Rather the effect of the PLC system on the VDSL modem was simulated by inducing a PLC signal into a co-axial cable with variable attenuators. The main finding of [20] and [21] is that the VDSL throughput experiences degradation when the ratio of the received VDSL signal power to the PLC signal power is less than 20 db. In other words, as long as the DSL cable run length is less than 300 m, the VDSL throughput will not be hindered. Note that in [19] the DM PLC-to-DSL coupling channels up to only MHz was measured, while in [20] and [21] no channel measurements were performed. In addition, the CM PLC-to-DSL coupling channel was not studied in [19, 20, 21]. It is important to note that the studies discussed above did not measure the PLC-to-DSL coupling channels, within a residential setting. Knowledge about the coupling channels between in-home PLC and DSL networks is essential in determining 9

28 an interference mitigation solution. Since there is neither a model nor previously conducted field measurements that characterize the interference environment between in-home PLC networks and a DSL modem, further field measurements of the DM and CM PLC-to-DSL coupling channels are required DSL Electromagnetic Interference Mitigation Solutions The copper twisted-pairs, utilized as the transmission medium by the DSL networks, were not initially designed to transmit broadband signals over a wide spectrum. As with all copper based wires, the twisted-pairs turn into an antenna at high frequencies [22]. This antenna radiates a portion of the transmitted signal as electromagnetic waves which cause interference to other systems operating at same frequencies. This radiated electromagnetic interference (EMI) is known as EMI egress. The copper twisted-pairs also pick up the radiated electromagnetic waves from other systems, and this causes what is known as EMI ingress. EMI to DSL can be classified as narrow-band EMI or wide-band EMI interference. Narrow-band EMI, as the name suggests, typically affects few of the DSL sub-channels. The most common narrow-band EMI to DSL systems are from amplitude modulation (AM) radio and amateur radio. Wide-band EMI, however, causes interference to a large portion of the DSL spectrum, which is usually from co-located systems that operate over the same spectrum as DSL. The most common wide-band EMI to DSL systems is from a co-located in-home PLC network [12, 23]. Techniques used to mitigate narrow-band and wide-band EMI to DSL systems are discussed in Sections and respectively Narrow-band EMI Mitigation Solutions Narrow-band EMI mitigation can be performed at the analog front of the DSL modem or within the digital structure. No optimal solution exists, but rather the choice 10

29 depends on each situation [22]. Digital narrow-band EMI mitigation solutions are usually preferred to analog solutions, since digital solutions are less expensive and more flexible than analog ones. However, in certain scenarios an analog solution would be necessary. For instance, if the EMI levels are high enough to saturate the analog to digital converter, EMI mitigation has to be performed at the analog front of the DSL modem. Note that a mixture of analog and digital narrow-band EMI mitigation solutions are utilized in most scenarios [22]. Various techniques are utilized in the literature to mitigate the narrow-band EMI effect on DSL signal. These techniques are either active or passive. Note that both analog and digital narrow-band EMI mitigation solutions can utilize either active or passive techniques. Passive techniques rely on information about the frequency of the interference and utilize filters (usually, notch filters) to eliminate the narrow-band EMI egress. Active techniques, on the other hand, rely on finding a reference signal or on a priori knowledge of the EMI to mitigate the differential mode narrow-band EMI egress. Usually, a reference signal is obtained via satellite or antenna in wireless systems; however, for DSL this is not required since the common mode signal can be utilized. Studies that take advantage of the common mode signal to mitigate the effect of narrow-band EMI were performed in [24, 25]. In [24], the authors mitigate the effect of impulsive noise on the DSL signal by utilizing the CM signal. Similarly, in [25], the CM signal was used to reduce the impact of EMI on the DM DSL signal Wide-band EMI Mitigation Solutions Wide-band EMI mitigation solutions fall into one of the following four categories: spectral notching, spectral management, interference cancellation, and spatial separation. In Section to , the work done to date in each of the four aforementioned wide-band EMI mitigation solution categories is discussed. 11

30 Spectral Notching Home networking standards, such as ITU G.hn [9], ensure that home networks that operate over the same spectrum as any of the DSL technologies do not interfere with the functionality of the DSL system. This is achieved by preventing home networks from utilizing frequencies occupied by the DSL system. Even though preventing home networks from operating over the DSL spectrum ensures successful operation of the DSL system, it denies home networks from benefiting from a wide range of frequencies. In addition, with the emerging of new DSL technologies that utilize wider bandwidth, such as G.fast (up to 212 MHz), denying PLC networks access to the spectrum that overlaps with DSL networks will render the PLC network inoperable [26] Spectral Management As an alternative to spectral notching, spectral management is proposed in [27] and [28] to alleviate the effects of the PLC interference by reducing the transmit power for PLC sub-carriers that cause interference to the received DSL signal. In [27], the authors assume a flat PLC-to-DSL cross-coupling channel and study the achieved DSL bit rates at various PLC transmit power levels. In [28], the authors propose reducing the transmit power for PLC sub-carriers that interfere with the downstream frequencies of VDSL2. As will be shown in Chapter 5, the proposed solutions achieve higher DSL bit rates than both solutions proposed in [27] and [28]. In addition, it will be shown that, mitigating the effects of PLC interference through spectral management negatively affects the PLC bit rates, while the proposed solution does not hinder the PLC bit rates Cross-Coupling Channel Equalization An interference cancellation solution was proposed in [28], which entails utilizing adaptive filters to estimate and mitigate the effects of the DM PLC interference on the DM DSL signal. In [28], the authors propose connecting the PLC network and 12

31 the DSL network, by utilizing a coupler that carries the PLC signal transmitted on the power lines to an adaptive filter that is connected to the DSL line. A fundamental assumption in [28] is that the coupling channel between the PLC transmitter and DSL receiver is the same as between the DSL transmitter and PLC receiver. As shown by the measurement campaign presented in Chapter 3, the PLCto-DSL coupling channel varies significantly from one outlet to the other. And the coupling channel experienced at the PLC receiver outlet will be very different than the coupling channel at the PLC transmitter outlet. In addition, the authors in [28] assume that the retraining of the adaptive filter is only required to accommodate the time-variation of the PLC channel, and they do not account for the outlet-to-outlet variation experienced by different users. Another wide-band EMI mitigation solution that utilizes an adaptive filter to equalize the cross-coupling channel was proposed in [29]. In this study, a time domain adaptive filter utilizes the CM signal to estimate the DM EMI over the asymmetric DSL (ADSL) spectrum ( MHz [30]). Because of the nature of the wide-band interference, this filter requires a long training time [29]. Note that this time domain filter requires a long training time for a DSL network operating over the ADSL spectrum, which is much smaller than the spectrum utilized by VDSL, VDSL2, or G.fast technologies. Thus, utilizing a time domain adaptive filter is not feasible for DSL technologies that operate over a spectrum that span tens of MHz. In addition, a core assumption made in [29] is that the relationship between the differential and common mode cross-coupling channels is a smooth function of frequency. As will be shown in Chapter 3, the ratio of the PLC-to-DSL DM to CM cross-coupling channels varies significantly from one sub-channel to the other Spatial Separation The authors in [31] propose that changing the location of the PLC transmitter or 13

32 the DSL receiver might be utilized to mitigate the interference between PLC and DSL; since the coupling levels vary from one power outlet to the other, the locations with the lowest coupling levels should be chosen. However, as will be shown in Chapter 3, the spatial separation between the DSL and the PLC modems does not have a significant impact on the coupling channels captured in the measurements presented in this paper. The average coupling levels are relatively the same for all outlets. 1.5 Thesis Contribution The goal of this thesis is to mitigate the mutual wide-band interference between co-located DSL and PLC networks, within the home environment. To achieve this goal, the focus of the thesis is on two areas: cross-coupling channel characterization and interference cancellation. The thesis contribution to the area of cross-coupling channel characterization is summarized in Section 1.5.1, while the thesis contribution to the area of interference cancellation is presented in Section Cross-Coupling Channel Characterization Cross-coupling channel characterization is essential to understand the various interference sources affecting the DSL networks. In this thesis, the cross-coupling channels between DSL and in-home PLC networks are studied. Residential field measurements of PLC-to-DSL cross-coupling channels have never been performed. A measurement campaign, which is discussed in detail in Chapter 3, has been designed and conducted to characterize the PLC-to-DSL cross-coupling channels. Both the DM and CM PLC-to-DSL cross-coupling channels are studied within two residential houses. For each of the two houses, the DM and CM PLC-to- DSL cross-coupling channels are measured for various rooms. A set of 480 DM and CM cross-coupling channel measurements between the DSL 14

33 and PLC networks has been collected for the two DSL modem installation scenarios discussed in Chapter 2. The main findings of the measurement campaign are: A strong DM PLC-to-DSL cross-coupling channel exists, with cross-coupling channel coupling gains ranging between 40 db and 80 db. The PLC-to-DSL cross-coupling channels are frequency selective and independent of the proximity between the PLC modem and the DSL modem. The PLC-to-DSL cross-coupling channels vary significantly from one outlet to the other. The CM to DM PLC-to-DSL interference ratio is not a smooth function of frequency, and varies from one outlet to the other Interference Cancellation The interference cancellation portion of this thesis is concerned with how to mitigate the mutual interference between a DSL modem and a co-located in-home PLC network, without negatively impacting the performance of the PLC network. Two PLC-to-DSL interference mitigation solutions are presented in Chapter 5. Both solutions utilize adaptive filters to estimate the differential mode interference on a tone by tone basis using the common mode signal. Only adaptive time domain cancellers that attempt to mitigate the wide-band EMI suffered by the DM DSL signal have been researched [29]. Due to the frequency selective nature of the PLC interference, these time domain adaptive filters cannot mitigate DM PLC interference. The first interference mitigation solution uses an adaptive frequency domain canceller that utilizes the CM DSL signal to extract information about the DM PLC interference and relies on restricting channel access to one PLC user per DSL super-frame. This scheduling is performed to combat the variation in cross-coupling channels from 15

34 one power outlet to the other. While this will increase the latency experienced by the PLC users, it will be shown that this additional delay is acceptable for multi-media applications. The second interference mitigation solution eliminates the need to restrict channel access to one PLC user per DSL super-frame by pre-multiplying the PLC symbol with the inverse of the cross-coupling channel. This eliminates the latency caused by the first solution but will be more expensive to implement due to the additional signal processing required. Similar to the first solution, the second solution also utilizes an adaptive frequency domain canceller that estimates the DM PLC-to-DSL interference from the CM DSL signal. As will be shown in Chapter 5, both proposed solutions successfully mitigate the PLC-interference on the DSL network, without degrading the performance of the PLC network. This is one of the main advantages that the proposed interference mitigation solutions have over the interference mitigation solutions discussed in Section Thesis Outline This thesis is organized as follows. In Chapter 2, a brief background on the DSL and PLC technologies is provided, along with description of the PLC-to-DSL interference environment within a residential setting. In Chapter 3, a measurement campaign that characterizes the PLC-to-DSL crosscoupling channels within two test sites is presented. Both the differential and the common mode cross-coupling channels were measured in two test sites. In Chapter 4, the current system model is presented. Additionally, the effects of the mutual wide-band interference on the achieved bit rates for both DSL and PLC systems are discussed. In Chapter 5, two interference cancelling solutions are presented. While both 16

35 solutions utilize adaptive filters to extract an estimate of the DM interference from the usually ignored CM signal, each solution has its advantages and disadvantages. The first solution while having the advantage of simplicity, adds restriction on the PLC channel access per DSL super-frame. This restriction introduces latency, albeit within the acceptable range for multimedia services. The second solution removes the restriction on PLC channel access per DSL super-frame; thus, eliminating the added latency introduced by the first solution. However, the second solution encompasses pre-shaping the PLC symbols before transmission. This adds to the complexity of the PLC transmitter, which is translated in the implementation cost. Finally, in Chapter 6, the thesis is concluded where the main findings of the thesis are summarized and future research direction is introduced. 17

36 Chapter 2 THE DSL and PLC TECHNOLOGIES The objective of this thesis is to enhance the coexistence between DSL and PLC networks in residential settings. DSL, despite its maturity, is still an active area of research. Recent advancement in DSL research lead to the development of standards that promise to provide data rates of up to 1 Gbits/s, which makes DSL the leading technology in copper-based wired broadband access. Similarly, PLC, because of its existence within every structure, has generated a lot of research advancement which has made it a popular choice for in-home networks. DSL and PLC utilize overlapping spectra; as a result, interference between both systems occurs. In order to mitigate the interference between co-located PLC and DSL networks, a better understanding of both networks is required. In this chapter, the DSL and PLC technologies are briefly introduced in Sections 2.1 and 2.2 respectively. 2.1 DSL Technology In this section, we introduce DSL technology. The layout of the DSL physical network is presented in Section The various DSL families are discussed in Section 2.1.2, and the DSL modulation, frame structure, and signal transmission techniques are presented in Section Finally, the DSL interference environment is presented in Section

37 From Core Network Central Office (Local Exchange) FTTEx FTTCab FTTB FTTH Street Cabinet (Pedestal) Street Cabinet (Pedestal) Building (remote DSLAM) USER USER USER Figure 2.1: Conventional DSL network. Twisted-pairs Fiber The Physical Network In a DSL network, as shown in Fig. 2.1, cables run from the core network to central offices (COs) (or local exchanges or access nodes (ANs)). Historically, the cables between the core network and the COs were twisted-pairs. In each of these cables, there is about 1000 twisted-pairs. From each CO, up to 100 twisted-pairs are connected to pedestals that service a number of houses. Finally, each house is connected to its respective pedestal by two twisted-pairs. However, only one of these twisted-pairs is used in conventional DSL networks. Thus, a house is connected by one passive copper twisted-pair to the CO. As a result, the DSL signal travels a very long distance which considerably reduces the transmission bit rate. Presently, however, sections of the twisted-pairs are being replaced by fiber optic cables. This connection is referred to as a hybrid fiber twisted pair (HFTP) connec- 19

38 tion [5]. These hybrid connections shorten the distance the DSL signal has to travel over passive copper twisted-pairs. HFTP connections have many deployment scenarios, depending on the connection from the CO to the end user. The main HFTP deployment scenarios are discussed in Sections to [32, 33] Fiber to the Exchange In the fiber to the exchange (FTTEx) deployment, copper twisted-pairs connect the CO to cabinets (or pedestals) that service a number of houses. Each house is connected to its respective cabinet by two twisted-pairs. However, only one of these twisted-pairs is usually active. Thus, in an FTTEx deployment, a house is connected by one passive copper twisted-pair to the CO. As a result, the DSL signal travels a very long distance which considerably reduces the transmission rate Fiber to the Cabinet In fiber to the cabinet (FTTCab) deployment, fiber cable is utilized to connect the CO to a remote optical unit (ONU) within the cabinet. Thus, DSL transmission over copper twisted pairs occurs only between the ONU and the customer premise. The FTTCab deployment has many variants, such as fiber to the curb (FTTC) or fiber to the distribution point (FTTDp). The main difference among these variants is the reach of the fiber connection within the DSL network Fiber to the Premise Fiber to the premise (FTTP) deployment has two variants. First is fiber to the building (FTTB), where a fiber cable connects the CO with a multi-dwelling building. Once in the building, a remote DSL access multiplexer (DSLAM) manages the DSL transmission over very short runs of twisted-pairs. The second variant is fiber to the home (FTTH), where the CO, and consequently the core network, is connected to the user by a fiber optic cable. FTTH is considered ideal and future proof [5], since a need 20

39 for data rates higher than the rates achieved by this deployment cannot be envisioned. However, installation of FTTH is both time consuming and very expensive, which is why it is not widely deployed The DSL Families DSL technologies have a very broad range, and as a result, they can be grouped by modulation scheme, data rate transmission direction, or by any other common factor between the technologies. In this section, the DSL technologies are classified into families, where each family share similar modulation schemes and data rate transmission directions. In Sections to , the major DSL families are introduced, while in Section the spectrum, data rate, and maximum reach of the various DSL technologies are compared Basic Rate Interface The basic rate interface (BRI), which is considered as the original DSL, is based on the integrated service digital network (ISDN) [34] technology. BRI provides the ability to transmit both data and voice signals over the PSTN, with data rates up to 160 kbits/s over a single twisted-pair. The maximum length of a single run of the twistedpair cannot exceed 18 kft; however, with the use of repeaters, operation over longer distances can be achieved. BRI is offered as a replacement for plain old telephone service (POTS). However, where the demand for POTS exists, BRI and POTS can coexist. Based on the geographical region of deployment, various modulation schemes for BRI are used. In North America, a four level pulse amplitude (4-PAM) modulation is used [35]. BRI provides a symmetric transmission rate, i.e., the entire bandwidth is utilized for both upstream (US) and downstream (DS) transmission. This is achieved by the use of echo cancellation. 21

40 The High Bit-rate DSL Family The high bit-rate DSL (HDSL) family of DSL technology is the most mature DSL technology. The HDSL family has witnessed a number of technologies, all of which have not been standardized. The common feature among theses variants is the ability to deliver symmetric high data rates over long distances, which is suitable for business customers. The two most common technologies in this family are the high bit rate DSL (HDSL) and the single-pair high-speed DSL (SHDSL), which are standardized by the international telecommunication union-telecommunication standardization sector (ITU-T) in G [36] and G [37] respectively. HDSL offers symmetric data rates of Mbits/s over two twisted-pairs. Each of the twisted-pairs carry a 784 kbits/s, over a single run of 12 kft. To increase the reach of a single run, more twisted-pairs can be used, in addition to utilizing repeaters. Similar to BRI, HDSL utilizes a 4-PAM modulation with echo cancellation. HDSL utilizes low frequencies to increase its reach; thus, HDSL cannot coexist with either POTS or BRI. SHDSL, on the other hand, provides data rates that range from 192 kbits/s to 2.3 Mbits/s, in increments of 8 kbits/s, over a single twisted pair. However, there is an option to utilize a second pair to improve the reach. SHDSL provides a multi-rate transmission, and relies on coding gain to achieve its high data rates. The modulation utilized is a 16-PAM modulation with trellis coding (TC) and echo cancellation. Various regional operational conditions are specified in G.991.2, with an annex dedicated to achieving data rates up to Mbits/s. Similar to HDSL, SHDSL is not compatible with either POTS or BRI The Asymmetric DSL Family Residential customers, unlike business customers, require higher data rates in the DS transmission direction. As a result, the asymmetric DSL (ADSL) technology emerged, 22

41 which is compatible with both POTS and BRI and deliver high data rates over a long reach. The three main technologies in the ADSL family are the asymmetric DSL (ADSL), the asymmetric DSL 2 (ADSL2), and the asymmetric DSL 2 plus (ADSL2+), which are standardized by the ITU-T in G [30], G [38], and G [39], respectively. ADSL offers asymmetric data rates up to 8 Mbits/s in the DS direction (from the CO to the customer) and up to 896 kbits/s in the US direction. The maximum reach of an ADSL line is 18 kft; however, at this reach the available DS data rate is Mbits/s. ADSL was the first of the DSL technologies to utilize DMT modulation. The frequency spectrum of ADSL starts from 25 khz up to 1.1 MHz; however, as an option, frequencies up to 80 khz can be avoided in presence of BRI. Frequency division duplexing (FDD) of US and DS transmission is achieved in ADSL by utilizing the frequencies from 25 khz to 138 khz for US transmission, while the frequencies from 138 khz to 1.1 MHz are reserved for DS transmission. ADSL2 is an extension of ADSL. The same frequency spectrum is utilized in both; however, ADSL2 offers asymmetric rates of up to 12 Mbits/s (DS transmission) and 1 Mbits/s (US transmission). The enhancement in the data rates is achieved by utilizing trellis coding, which was optional in ADSL, and the ability to use one-bit constellations, such as binary PAM [4]. In addition, an all-digital mode is present in ADSL2, where the entire bandwidth can be utilized, in absence of POTS and BRI. ADSL2+ is an enhanced version of ADSL2, where the DS bandwidth is extended to 2.2 MHz. As a result, the achieved DS data rates are increased to 20 Mbits/s, especially for customers close to the CO. For various regional operational conditions, the DS transmit PSD can be shaped to meet specific requirements. 23

42 The Very High Speed DSL Family The main feature of the very high speed DSL (VDSL) family is that it provides both symmetric and asymmetric transmission, which is suitable for both business and residential customers respectively. While the ADSL family successfully offer high bit rate over long distances, the demand for higher rates kept on rising. The VDSL family of DSL technologies offers the highest data rates achieved over copper twistedpairs. However, these high data rates are only available over short distances. The two DSL technologies within the VDSL family are the very high speed DSL (VDSL) and the very high speed DSL 2 (VDSL2), which are standardized by the ITU-T in G [40] and G [2] respectively. VDSL offers asymmetric data rates that can reach 52 Mbits/s for DS transmission and 1.5 Mbits/s for US transmission, for customers within a 1 kft radius from the ONU. Symmetric data rates of 10 Mbits/s can be achieved over distances of 4.5 kft. DMT modulation, along with FDD of US and DS transmission is used in VDSL. VDSL uses frequency ranges from MHz to 12 MHz, with these frequencies divided into two DS and two US band plans. In addition to being compatible with both POTS and BRI, VDSL is compatible with the entire ADSL family. VDSL2 offers data rates that can reach up to 200 Mbits/s (asymmetric) and 100 Mbits/s (symmetric), over short distances. VDSL2 uses a wider bandwidth (from 25 khz up to 30 MHz) and has a longer reach than VDSL. DMT modulation, along with FDD of US and DS transmission is used in VDSL2. G standard specifies four major band plans, with eight profiles, where each of these band plans is suitable for a specific HFTP deployment [41]. Band plan 8 (4 profiles: 8a, 8b, 8c, and 8d) utilizes a bandwidth of 8.6 MHz with data rates up to 50 Mbits/s. Band plan 12 (2 profiles: 12a and 12b) utilize a bandwidth of 12 MHz and has a maximum data rate of 68 Mbits/s. Both band plan 8 and 12 are suitable for FTTEx deployments, 24

43 with an 8 kft maximum length of twisted-pair run. Band plan 17 (1 profile: 17a) utilizes a bandwidth of 17.7 MHz and provides a maximum data rate of 100 Mbits/s. This band plan is suitable FTTCab deployments, where the maximum length of the twisted-pairs is approximately 5 kft. Finally, the Band plan 30 (1 profile: 30a) utilizes the entire VDSL2 bandwidth and offer data rates up to 200 Mbits/s. This band plan is suitable for HFTP deployments where the twisted-pair length does not exceed 1 kft, such as FTTB deployments. Similar to VDSL, VDSL2 is fully compatible with POTS, BRI and the entire ADSL family Fast Access to Subscriber Terminal Fast access to subscriber terminal (G.fast) technology is the most advanced among the DSL technologies. Currently, G.fast is being standardized by ITU-T in G.9700 [7]. G.fast, which has the potential of delivering aggregate data rates of up to 1 Gbits/s, is suitable for FTTP deployment, where the maximum length of the twisted-pairs is approximately 0.82 kft (250 m) [8]. This high data rate is achieved by utilizing a wider bandwidth. G.fast is being studied for two band plans: from 2.2 MHz to 106 MHz and to 212 MHz. Instead of FDD, G.fast is expected to utilize time division duplexing (TDD) of US and DS transmission. It is expected that the US to DS ratio will be flexible, but constant among lines served by the same distribution point. The US to DS ratio has to be constant to mitigate the NEXT effect. Finally, it is expected that G.fast will be compatible with all the aforementioned DSL technologies DSL Families Comparison Table 2.1 compares the utilized bandwidth, the maximum data rates, the maximum reach over twisted-pairs, and the ITU-T standards for the DSL technologies mentioned above. In this thesis, the focus is on profile 30a of the VDSL2 technology. Thus, any 25

44 Table 2.1: Comparison of DSL Families. Family Technology Spectrum Max. Data Rates Reach ITU-T MHz Mbits/s kft Standard BRA ISDN up to G.961 HDSL up to G HDSL on two pairs SHDSL up to G on a single pairs ADSL DS: G US: 0.64 ADSL ADSL DS: G US: 1 ADSL DS: G US: 1 VDSL (net) 1.0 G VDSL 10 (net) 4.5 VDSL (net) 1.0 G (net) 5.0 G.fast G.fast (212) 1000 (net) 0.82 G.9700 referral to the DSL spectrum, indicates frequency ranges up to 30 MHz. Note that the solutions to interference mitigation, introduced in this thesis, are not frequency dependent. As will be shown in Chapter 5, the interference mitigation filters proposed in this proposal can be utilized by any DSL technology that is based on DMT modulation DSL Signalling and Frame Structure Currently, DSL systems utilize discrete multi-tone (DMT) modulation. DMT modulation, which utilizes a discrete Fourier transform (DFT) based block transceiver, is a variant of orthogonal frequency division multiplexing (OFDM). One of the main differences between DMT and OFDM is that the output of the DFT block in DMT is real valued samples, while the output of the DFT block samples in OFDM are complex. The rest of this section is organized as follows. DMT modulation and DFT-based 26

45 transceivers are discussed in Section DSL signalling and frame structure are discussed in Section , while the DSL frame structure is discussed in Section Discrete Multi-Tone Modulation In DMT, the channel bandwidth is partitioned, utilizing the inverse discrete Fourier transform (IDFT), into orthogonal sub-channels (or tones). This division simplifies the channel equalization needed to negate the channel dispersive effect. Moreover, DFT-based transceivers are favoured in multi-channel modulation due to the availability of computationally efficient implementation methods for the DFT, such as the fast Fourier transform (FFT). Fig. 2.2 shows three sub-channels (sub-channels 10, 11, and 12), where the subchannels remain orthogonal at the sub-carrier frequency. Note that, at the sub-carrier indexes, i.e., the center frequencies of the sub-channels, the contribution from other sub-channels is zero. Thus, from the receiver s point of view, sampling the received signal at the sub-carrier frequencies is equivalent to having multiple parallel nonoverlapping sub-channels. The number of sub-channels depends on the frequency separation between subcarriers (i.e., the sub-carrier spacing f) which varies from one system to another. The main factor that affects the number of sub-carriers is the coherence time of the channel. The coherence time is defined as the duration during which there is no variation in the channel impulse response. The number of sub-carriers is usually chosen such that the number of sub-carriers is maximized and the symbol duration is less than the coherence time. Variation in the channel during the DMT symbol duration degrades the orthogonality of the sub-channels. Since the DMT symbol consists of multiple tones, it inherits the robustness of a tone, i.e., a sinusoid signal, to dispersion. This fact is exploited to simplify the channel 27

46 1 0.8 Carrier no. 10 Carrier no. 11 Carrier no Sub carrier index Figure 2.2: Sinc functions at various sub-carrier indexes. equalization for DMT-based systems. The insensitivity of a tone to the dispersion of the channel can be easily seen in the frequency domain, where the Fourier transform (FT) of an infinite duration tone is a delta function. No matter how dispersive the channel is, distortion of the tone by the channel is limited to amplitude and phase changes, which can be negated by a single-tap equalizer. For a signal that is composed of multiple tones at various frequencies, the effect of the dispersive channel will be also limited to changes in the amplitudes and the phases of each of the delta functions corresponding to the various sinusoids. The same concept can also be applied to discrete-time tones, where the DFT of a discrete-time tone is also a delta function in the discrete-frequency domain. However, during the truncated DMT symbol, the received signal is composed of the summation of different replicas each of which is delayed differently due to the dispersive channel. Thus, the received multiple tones are no longer orthogonal during the DMT symbol time. This, along with causing inter-symbol interference (ISI), complicates the equalization of the channel effect. To overcome ISI and simplify the channel equalization process, two things are 28

47 performed. First, a guard interval is utilized between consecutive symbols such that the ISI occurs during this guard interval. Additionally, few samples are taken from the end of the DMT symbol and appended to the beginning of said symbol. By doing so, the received multiple tones are guaranteed to have suffered the same dispersion, and thus, the DFT of these multiple tones are delta functions and the effect of the dispersive channel will be limited to changes in the amplitudes and the phases of each of the delta functions. This is achieved, as will be explained below, via adding a cyclic prefix to the input sequence before applying it to the channel. A block diagram of the DMT transmitter is shown in Fig The bit stream is converted from a serial stream of bits into N parallel sets of bits via a serial to parallel (S/P) converter. The number of parallel sets corresponds to the number of sub-channels, and the number of bits in each of the parallel sets is based on each subchannel signal to noise ratio (SNR). Each set of bits is modulated via a quadrature amplitude modulation (QAM) encoder, such that the output of the QAM encoder is the complex N 1 vector X define by X = [X 0, X 1,..., X N 1 ] T, (2.1) where X i is the modulated QAM symbol for sub-channel i. X 1 x 1 Bit Stream S/P QAM Encoder X 2 x 2 X N 2N-point complex-to- real IFFT P/S x 1, x 2,..., x 2N Add Cyclic Prefix D/A and Filter x 2N Figure 2.3: DMT transmitter block diagram. The IDFT of the output of the QAM encoder is obtained via a 2N-point complex 29

48 to real IFFT operation, which results in the 2N 1 real vector x defined by x = [x 0, x 1,..., x 2N 1 ] T. (2.2) A real output is required because, in DMT, the output of the IFFT is applied directly to the channel after digital to analog conversion. To obtain the real vector x from the complex vector X via the IFFT operation, a 2N 1 vector X H with Hermitian symmetry property has to be formed from the N 1 complex vector X. This can be easily performed by X H = [ Re{X N 1 }X 0, X 1,..., X N 3, X N 2, Im{X N 1 }, X N 2, X N 3,..., X 2, X 1, ] T, (2.3) where Re{X i } and Im{X i } indicate the real and the imaginary parts of X i respectively, and X i is the complex conjugate of X i. The elements of x, i.e., the IDFT of X H are calculated by x n = 1 2N 1 X H,k e j(2π/2n)kn, n [0, 2N 1]. (2.4) 2N k=0 The output of the IFFT block, i.e., the time domain input samples defined by vector x, is passed through a parallel to serial (P/S) converter, which converts the parallel sub-symbols to a series of sub-symbols as shown in Fig Before the inserting the DMT symbol into the channel, the cyclic prefix is added to the beginning of the DMT symbol. The cyclic prefix is a copy of the last v 1 sub-symbols of x 2N, which is appended to the beginning of time domain input sequence as shown in Fig The length of v is chosen such that it equals the length of the channel impulse response. In that manner, the ISI between consecutive symbols will be confined to the cyclic prefix. Note that instead of appending the last v 1 sub-symbols to the DMT symbol before transmitting it over the channel, one could append v 1 zeros, to confine the ISI to the cyclic prefix. However, the cyclic prefix serves another purpose. 30

49 As mentioned earlier, adding the cyclic prefix serves the purpose of simplifying the channel equalization. By appending the last v 1 samples of the time sequence representing the DMT symbol to the beginning of the sequence, the aperiodic input time sequence seems periodic over the length of the convolution. Thus, the linear convolution can be represented by a circular convolution, once the cyclic prefix is removed. Modelling the linear convolution as a circular convolution is important because it allows the usage of DFT. This means that, after removing the cyclic prefix, the DFT of the received sequence equals the product of the DFTs of the transmitted sequence and the channel impulse response. Consequently, the channel effect can be negated using a simple one-tap equalizer for each sub-channel. Finally after adding the cycling prefix to the input sequence, the input sequence is converted to an analog signal via digital to analog (D/A) block as shown in Fig Before the analog signal is applied to the channel, a transmit filter, shown in Fig. 2.4, may be utilized to eliminate any out of band power leakage and to ensure that the transmitted signal power spectral density (PSD) remains in a specific range. Note that the PSD levels can be controlled digitally, but sometimes it is easier to control it via the analog front end. [42] Fig.2.4 shows the DMT receiver block diagram. Once the signal is received, it passes through the receiver filter which minimizes the out of band noise, after which it is converted from analog to digital via the analog to digital (A/D) block. The cyclic prefix is then stripped from the received sequence y. Through a P/S converter the received sequence is converted to 2N sub-symbols. These sub-symbols are fed to an FFT block which performs and 2N-point real-toimaginary conversion. The FFT block takes the DFT of the received sequence and reverses the Hermitian symmetry. Recall, the cyclic prefix is added to the input sequence to force periodicity over the convolution interval, which results in a linear 31

50 y 1 Y 1 ^ X 1 y 2 Y 2 FEQ ^ X 2 QAM Decoder P/S Bit Stream Filter and A/D Add Cyclic Prefix y 1, y 2,..., y 2N S/P N-point complex-to- real IFFT Y N ^ X N y 2N Figure 2.4: DMT receiver block diagram. convolution being equivalent to a circular convolution. Thus, for a noiseless channel, the received sequence after the FFT block is a multiplication of the input sequence and the channel frequency response, which can be written in vector form as Y 0 X 0.H 0 Y 1 X 1.H 1 Y = =, (2.5).. Y N 1 X N 1.H N 1 where H i is the frequency response of the i th sub-carrier defined in (2.6). H k = 1 N 1 h n e j(2π/2n)kn, k [0, N 1]. (2.6) N k=0 To extract the X i from Y i, the received sequence is passed through a single-tap filter per sub-channel called frequency-domain equalizer (FEQ) as shown in Fig The coefficients of the FEQ is the inverse of the channel frequency response. Note that division is not usually preferred from an implementation perspective. Thus, the single-tap FEQ is implemented via a complex operation that involves scaling and rotating Y i to mitigate the effect of channel impairment [22]. In presence of noise, the outputs of the FEQ are not exactly equal to X but rather ˆX, which is an estimate of X. This estimate of X is then passed through a QAM encoder followed by a P/S converter, as shown in Fig. 2.4, to reconstruct the 32

51 transmitted bit stream. Note that in absence of coding, ˆX is passed through a simple symbol by symbol decision process to recover the constellation points before utilizing the QAM decoder. However, if trellis encoding (i.e., convolution codes) is utilized, then a trellis decoder is required to decode the constellation points DSL Signalling DSL signals are transmitted in differential mode (DM) where complementary signals are transmitted over a twisted-pair of wires. DM signalling provides resilience to electromagnetic interference (EMI) as any external interference will couple identically on both of the pairs, and thus, any interference will be eliminated at the receiver. However, in practical situations, external interference affects each wire in a pair differently. The common mode (CM) signal, which is the arithmetic mean of the signals, can be determined at the receiver, with little extra cost. The CM signal contains information about the external EMI; thus, the CM signal can be utilized to estimate the DM EMI. That estimate can then be used to cancel the interference. Balun DM Signal CM Signal Figure 2.5: DM and CM signal at the receiver. Fig. 2.5 shows a balun that is connected to a twisted-pair, and outputs two signals: the DM and the CM signal. A balun is a resistance transformer that converts the balanced DSL signal to a differential mode signal, at the receiver end. From the center tap of the balun, the common mode signal, which is usually ignored since it does not contain relevant information about the desired signal, is obtained. Ideally, a balun will output a DM signal that is only made up of the desired signal and a 33

52 CM signal that only contains noise and electromagnetic interference. However, due to imperfections in the balun, and in the twisted-pairs, the DM portion of the EMI is found in the DM signal DSL Frame Structure The DSL network utilizes a dedicated channel for each user. Data is transmitted over the DSL channel in super-frames. Since each DSL user has its own dedicated channel, all super-frames on the DSL channel belong to a single user. The super-frame for VDSL2 is divided into 257 frames. The last frame within a super-frame is the synchronization (sync) frame [2]. Sync symbols are usually utilized for only synchronization purposes, where bits are modulated via 4-QAM constellation; however, pilot signals can be transmitted during the sync frame for channel state estimation purposes [6]. To facilitate parallel sub-channel estimation, orthogonal pilot sequences are again used [43]. Note that the pilot sequences are vendor specific [44]. DSL Super-frame t Synch T S T SF Data T D Figure 2.6: DSL super-frame structure. A super-frame, shown in Fig. 2.6, has duration of T SF. Note that the value for T SF varies from one standard to the other; for VDSL2 T SF =64.25 ms [2], while for G.fast T SF < 10 ms [8]. Recall, the DSL technology utilized in this thesis is VDSL2. For simplicity, and without loss of generality, let us assume that the first frame in each 34

53 super-frame is a synchronization frame and there are 256 frames per super-frame. Thus, as shown in Fig. 2.6 a DSL super-frame constitutes a synchronization frame followed by 255 data frames, each with a duration of T S =0.25 ms. Thus, in this thesis, the duration of VDSL2 super-frame is assumes to be 64 ms The DSL Interference Environment Interference in DSL can be classified as intrinsic, such as thermal noise and crosstalk, or extrinsic, such as impulsive noise and PLC interference [45]. In this section, the various sources of interference in DSL networks and the interference mitigating techniques utilized to combat them are discussed. The rest of this section is organized as follows. In Section , DSL intrinsic interference sources are discussed, while DSL extrinsic interference sources are presented in Section Intrinsic Interference Crosstalk (xtalk), which occurs when a signal power leaks from one twisted-pair to another, is a wide-band interference and is the main cause of errors in a DSL network [22]. There are two types of crosstalk: near end crosstalk (NEXT) and far end crosstalk (FEXT) [46]. Tx/Rx Signal NEXT Tx/Rx Figure 2.7: Near end crosstalk among twisted pairs. NEXT occurs among twisted-pairs when an interfering signal is transmitted from the same end of the cable as the receiver, as shown in Figure 2.7. FDD is utilized to 35

54 prevent NEXT among DSL lines served by the same distribution point [22]. However, since TDD is used instead of FDD in G.fast, the US to DS ratio should be constant among twisted-pairs served by the same distribution point. By fixing the US to DS ratio, it is ensured that all transceivers at the customers premises (or at the CO) are transmitting at the same time, which eliminates NEXT. Tx/Rx Signal FEXT Tx/Rx Figure 2.8: Far end crosstalk among twisted pairs. FEXT, on the other hand, occurs when an interfering signal is transmitted from the end of the cable that is opposite to the receiver as shown in Figure 2.8. Various mitigation techniques are utilized to mitigate the FEXT, the most prominent of which is vectoring [6]. Vectoring is discussed in more detail Section The concept of interference mitigation is essential to any communication systems. Various techniques over the years have been developed to estimate and cancel DSL intrinsic interference. In Sections to , interference mitigation techniques that are utilized in mitigating DSL intrinsic interference are highlighted Adaptive Filters Mitigating the effects of NEXT by utilizing time domain adaptive filters was proposed in [47] [48]. For a cable containing m+1 twisted pairs, m filters are required to mitigate the NEXT effect on each twisted-pair. Thus, a total of (m+1)m adaptive filters are needed to eliminate NEXT in said cable. In addition, access to the disturbing signals is required for this approach to work. The computational complexity of this 36

55 approach makes it impractical, especially at the customers premises. A variation of this approach was proposed in [49], where the highest n NEXT sources are cancelled using n adaptive filter. However, the same limitations on this variation still exist, along with the task of detecting the highest n disturbers Frequency Division Duplexing Frequency division duplexing (FDD) of US and DS transmission in DSL networks is used to mitigate the effects of NEXT [22]. By utilizing FDD, transceivers at the same end of the cable are transmitting and receiving at different frequency bands. This creates a zipper pattern where the available spectrum is divided into smaller sub-spectra. The sub-spectra are allocated in an alternating manner to US and DS transmission; thus, eliminating NEXT. However, as a result of utilizing FDD, the available VDSL2 bandwidth is restricted to specific non-overlapping frequency ranges, which affects the overall transmission rate. In addition, in presence of multiple DSL systems that employ different duplexing schemes, utilizing FDD is rendered ineffective in mitigating the effects of NEXT [50] Vectored Transmission For a single DSL system, the utilization of FDD results in the elimination of NEXT. Thus, the major source of intrinsic interference for DSL is FEXT. In 2001 a new transmission technique called vectored transmission (vectoring) was proposed in [51], which can be utilized to eliminate FEXT from a DSL network for all transceivers within the same cable binder co-located at the DSLAM. As long as all the twisted-pairs in the cable are connected to the same CO and are managed by the same service provider [52], FEXT among twisted-pairs can be virtually eliminated, which results in transmission rates up to 100 Mbits/s [44]. However, if not all the twisted-pairs within the cable binder are considered in the joint signal processing (i.e, presence of uncontrolled lines), the resulting FEXT will negatively 37

56 impact the vectoring process [53]. In 2010, vectoring was standardized in [6], and is being adopted by a number of service providers as the next-generation broadband technology [54]. While vectoring shows promising improvement in data rates, especially over short lines, the presence of uncontrolled lines significantly deteriorates this improvement. This is a severe limitation of vectoring, especially when local loop unbundling (LLU). In LLU the incumbent local exchange carriers are forced to share their infrastructure with other operates. Since multiple operators share a cable binder, and since channel state information is not shared among the multiple operators, mitigating the effect of FEXT via vectoring will not be possible. In [55], it was shown that data rates dropped from 100 Mbits/s to 70 Mbits/s in presence of a single uncontrolled line. Another limitation of vectoring is its complexity in situations where a relatively large number of twisted-pairs served by the CO [56] Extrinsic Interference Extrinsic interference can be classified as narrow-band or wide-band interference. Radio frequency interference from AM and amateur radio, which usually affects few tones at a time, is considered narrow-band interference. These types of narrow-band extrinsic interference usually are geographically variable [45]. On the other hand, wide-band extrinsic interference is usually from systems co-located with the DSL system, and utilizes the same spectrum as DSL. One of the most prominent of these technologies is power line communications (PLC) [12, 23]. Techniques that mitigate the effect of EMI on DSL, both narrow-band and wide-band EMI, have been discussed in Section Note that this thesis focuses on wide-band extrinsic interference from PLC on DSL in a residential environment. Recall, the objective of the thesis is to propose solutions that mitigate the interference between the DSL modem and a co-located 38

57 PLC network, within a residential setting. 2.2 PLC Technology Although PLC as a technology has been proposed since the beginning of the last century [57], it is only recently that PLC networks are seen as a viable option for home networking. Communications over power lines utilized a narrow-band single carrier at its early stage [58]; currently, broadband PLC utilize a wide spectrum that reaches up to 100 MHz and has the potential of delivering data rates of up to 500 Mbit/s [3]. As discussed in Chapter 1, delivering the Internet to customers over the power lines faced many challenges due to the cost and availability of cheaper alternatives. However, PLC technology saw success in various areas such as vehicular networks, smart grid applications, municipal applications, and local area networks (LANs). Because of the existing extensive power line infrastructure and the continuous advancement in LAN technology, PLC is becoming an attractive solution to create in-home networks among devices that might benefit from access to the Internet. In this section, broadband PLC technology and its application in home networking are introduced. The architecture of a typical in-home PLC network is presented in Section 2.2.1, while broadband PLC signalling and frame structure is discussed in Section PLC In-home Network Electric power is carried to neighbourhoods over high voltage power lines in the range of 6 to 16 kv. For residential usage, this high voltage is reduced via step-down distribution transformers to 240 V [1]. Fig. 2.9, shows a distribution transformer with two leads. The two leads of the transformer are connected to Line 1 and Line 2. 39

58 The potential between Line 1 and Line 2 is 240 V [58]. In north America, the 240 V is split into two phases via a central tap that is connected to the neutral of the house [59]. The potential between the neutral and each of Line 1 and Line 2 is 120 V, as shown in Fig Thus, within the panel, there are two lines, the neutral, and the earth ground. Note that the earth ground connection is not shown in Fig Distribution Transformer Panel Line 1 16 kv Neutral 120 V 240 V Line V Figure 2.9: Electrical wiring in north America [1]. The power lines within the house are terminated at bus bars with the panel (also known as the circuit breaker panel). These power lines are either single-phase or two-phase. The single-phase power lines are connected to outlets within the house to supply light fixtures and small appliances such as TV, PLC couplers, etc, with electricity. Large appliances such electric stoves, water heaters, etc, are connected to the two-phase since they require the usage of the 240 V between Line 1 and Line 2. In-home PLC networks are utilized for many purposes. An in-home PLC LAN can have multiple functions, and support communications on various spectra. There are two categories of PLC networks: narrow-band and broadband. Narrow-band PLC (NB-PLC) is used for AMR and home automation; broadband PLC (BB-PLC) is mainly used for distribution of triple play services within the house. Fig shows a PLC network within the home environment. The unshielded power lines run throughout the house. PLC transceivers are connected via the mains outlets to form 40

59 the PLC network. To these transceivers, devices such as computers and television, are connected. Power Lines Broadband Router PLC Transceiver PLC-DC PLC Transceiver PLC Transceiver Figure 2.10: In-home PLC network. Note that cross-phase coupling, i.e., the ability of the PLC signal to couple from one phase to the other, is an issue for NB-PLC. The low operating frequency used by NB-PLC prevented the signal from coupling across phases because the impedance between phases caused the signal to attenuate significantly. A capacitive coupler is required to enhance the coupling between phases [59]. However, since BB-PLC utilize high frequencies, the bus bars in the panel act as a capacitor which enables the cross-phase coupling. Additionally, BB-PLC signal couples across phases at large appliances, which are connected to both phases. Thus, cross-phase coupling is not an issue for BB-PLC [1]. 41

60 As will be discussed in Section 2.2.2, the shared PLC channel access by PLC transceivers (users) is performed in one of two ways, depending on the type of traffic. Contention based channel access does not require a controlling entity to organize the channel access among users. Contention free channel access, as with other broadband access technologies with shared mediums, requires a domain controller to facilitate access among PLC users based on each user s requirement. Note that this is the case when a minimum quality of service is required. In contention free channel access, one of the PLC transceivers takes over as the PLC domain controller (PLC-DC). The PLC-DC regulates access to the PLC channel and is connected to the Internet via a broadband router, as shown in From Fig In addition to accessing the Internet via a computer, services such as IPTV or voice over Internet protocol (VOIP) are provided via the PLC transceivers PLC Signalling and Channel Access In this section, the aspects of the physical and MAC layer of in-home PLC networks relevant to this thesis are discussed. In Section , PLC signal transmission and modulation are discussed, while the MAC standards for broadband PLC networks are introduced in Section PLC Signalling and Modulation Power Lines PLC Coupler DM Signal CM Signal Figure 2.11: Simplified PLC coupling circuit. 42

61 Similar to DSL, differential mode signalling is utilized in PLC. Fig shows how a capacitive PLC coupling circuit used to transmit broadband signals over inhome power lines [60]. The objective of this coupling circuit is to filter out the 60 Hz high voltage waveform and transmit a broadband signal over the power lines. The PLC modems also have fuses before the capacitors and diodes after the transformer for over voltage protection. The fuses and diodes are not shown in Fig What is relevant to this thesis is the functionality of the transformer. Similar to the balun, the coupler acts as an impedance transformer, where two balanced signals are converted to two unbalanced signals. The DM signal, which is the difference between the two balanced complementary signals, is utilized to carry the desired Signal. The CM signal, is usually ignored, since it mainly contains EMI. PLC utilizes DFT-based transceivers for signal modulation/demodulation [58], which allows the partitioning of the broadband channel into narrower sub-channels. Among the variants of DFT-based modulation techniques, DMT modulation is the most suitable for PLC communications. This is because base-band transmission is utilized in PLC networks. DMT modulation was discussed in detail in Section Note that the PLC sub-carrier spacing f PLC is larger than the DSL sub-carrier spacing, since the PLC channel state varies more rapidly than the DSL channel. Recall, the number of sub-carriers (or sub-channels) is inversely proportion with the coherence time of the channel. Since the DMT symbol duration has to be shorter than the coherence time of the channel to prevent degrading the orthogonality of the sub-channels, and since the coherence time of a PLC channel is shorter than the coherence time of a DSL channel, the DMT symbol duration for a PLC network is shorter than that of the DSL network. And due to the inverse relationship between time and frequency, a shorter PLC symbol duration means a wider PLC sub-carrier spacing. 43

62 MAC for Broadband PLC As discussed in Section , the DSL network utilizes a dedicated channel for each user. The PLC channel, on the other hand, is shared among all the users in the PLC network. As will be discussed in Chapter 3, the cross-coupling channel varies significantly from one PLC user to the other (i.e., from one power outlet to the other). Thus, to propose solutions that overcome the outlet-to-outlet variations in the PLC-to-DSL cross-coupling channels, it is essential to understand the channel access mechanism utilized by in-home broadband PLC networks, for multimedia applications. Various standard bodies have developed standards for the MAC layer of a broadband PLC network. Among the most wide-spread standards are HomePlugAv [61], IEEE P1901 [62], and G.hn [9], which are standardized by the HomePlug Consortium, the IEEE, and the ITU-T standard bodies, respectively. In all the aforementioned three standards, PLC nodes are divided into two classes, PLC domain controllers (PLC-DCs) and PLC users. In [62], the PLC-DC is called the local administrator, while in [61] and [9], the domain controller is referred to as the connection manager and the domain manager, respectively. PLC users are referred to slave stations or PLC nodes. PLC-DCs have various functions such as transmitting beacons to provide info on the contention periods and contention free periods, broadcasting information about channel, and assigning time-slots to PLC users. Usually, the first node that joins the network takes over as a PLC-DC [9]; however, as will be discussed in Chapter 5, the PLC user closest to the DSL modem is the PLC-DC in this thesis. All the above standards employ one of two channel access mechanisms, either carrier sense multiple access with collision avoidance (CSMA/CA) or time division multiple access TDMA [63],[64]. CSMA/CA is reserved for best effort traffic, where channel access is contention-based. Best effort traffic, such as web-based or 44

63 applications, is a classification of traffic that does not have a quality of service (QoS) requirement, such as packet loss, latency, etc. Sensitive traffic, such as VOIP and video conferencing, is traffic that requires its packets to be delivered on time with a certain error threshold. For this type of sensitive traffic, TDMA is used as an access mechanism where the PLC-DC assigns time-slots to the various PLC users according to their QoS requirements. In this thesis, it is assumed that the in-home PLC LAN is utilized for QoS traffic; thus, the PLC network employs TDMA as a channel access mechanism. The TDMA channel access for a three-user PLC network (User A, User B, and User C) is depicted in Fig The vertical axis indicates the names of the users and the combined channel, while the horizontal axis indicates the time, which is divided into time-slots. The user s packets only represent when traffic becomes ready for transmission at each node. Thus, Fig shows how the time-slots are allocated to the various PLC users. PLC t 1 t 2 t 3 t 4 t 5 t 6 t 7 t 8 User A A A A A A User B B B B B B User C C C C C Channel Slot 1 Slot 2 Slot 3 Slot 4 Slot 5 Slot 6 Slot 7 - B A C A C B T P Figure 2.12: PLC frame structure for a 3-user PLC network. t Assuming that all three PLC users have the same QoS requirement, the PLC-DC allocates the next available time-slot to the PLC user with the earliest frame arrival 45

64 time, (i.e., the PLC user whose frame was queued for transmission before all other PLC frames in the network). In Fig. 2.12, the first frame arrival is between t 1 and t 2 from User B followed by User A then User C, where t i is the start time of Slot i. Thus, the PLC-DC assigns the next available time slot, Slot 2, to User B; after which, the PLC-DC allocates Slot 3 and Slot 4 for User A and User C, respectively. Note that Slot 1 is not used because the first frame arrival occurs after the beginning of Slot 1. Note that it is assumed that the PLC transceivers follow the recommendations of the ITU-T G.hn home networking standard. While the interference mitigation solutions discussed in Chapter 5 propose some modification to MAC layer of the PLC network, these modifications comply with the recommendations of the G.hn standard. 2.3 PLC Noise Environment PLC utilizes unshielded power-lines which are more sensitive to EMI than the twistedpairs utilized by DSL. The power lines within the home pick up EMI from various noise sources such as AM and amateur radio, appliances within the house, etc. In the literature, there are three main classes of noise present on the PLC channel. However, in this thesis, a fourth class is introduced. The first class of noise is the coloured background noise, which is the sum of various noise sources and is considered time-invariant since the noise level does not change for various consecutive AC cycles [58]. Typically, the coloured background noise has a PSD of -145 dbm Hz [65]. The second class of noise is the impulsive noise which is characterized with shorter durations and higher amplitudes than the background noise. Impulsive noise within PLC system is either synchronous or asynchronous to the mains frequency, or in form of isolated impulses. Synchronous impulsive noise occurs at the frequency of the AC 46

65 mains and is usually due to silicon controlled rectifiers and from electronic circuits. Asynchronous impulsive noise occurs at frequencies much higher than the AC mains, and a typical source of asynchronous impulsive noise is switching regulators. In the literature, the PSD of impulsive noise is in the range of -105 dbm db [65]. Hz The third class of noise to PLC systems is the narrow-band interference from AM and amateur. As was the case with narrow-band DSL interference, this type of interference affects few sub-channels at a time and is usually mitigated via notch filters. Finally, the fourth class of noise to PLC systems is wide-band interference from co-located DSL systems, which is introduced in this thesis in Chapter 4. This class of noise typically affects a number of sub-channels. For PLC, the dominant wide-band interference source occurs due to the US-DSL transmission. Further detail on the effect of DSL on the performance of a co-located PLC system is presented in Section

66 Chapter 3 CROSS-COUPLING CHANNEL MEASUREMENTS The main goal of this thesis is to mitigate the interference between co-located PLC and DSL networks, within the home environment. The interference mitigation solutions proposed in this thesis utilize adaptive filters to extract an estimate of the DM interference from the CM signal. As will be discussed in Chapter 5, both the DM and CM PLC-to-DSL cross-coupling channel frequency response matrices are used in the training of the adaptive filters, and in the performance evaluation of the proposed interference mitigations solutions. Field measurements of the interference environment between a DSL modem and a co-located in-home PLC network have never been performed. In addition, a model that describes the PLC-to-DSL cross-coupling channels within a residential house does not exist. Thus, a field measurement campaign is required to study the interference environment between a DSL modem and an in-home PLC network. The measurement campaign, introduced in this chapter, characterizes both the DM and CM PLC-to-DSL cross-coupling channels in two test-sites. A vector network analyzer (VNA) based measurement system is developed to measure the complex PLC-to-DSL cross-coupling channels. Measurements were performed in residential houses to study the effect of the co-located DSL and PLC systems on each other. Among other findings, analysis of the measurement data reveals that the PLC-to-DSL cross-coupling channels are frequency selective and the spatial separation between the PLC and DSL modems have no significant impact on the interference levels. The rest of this chapter is organized as follows. In Section 3.1, the DSL modem 48

67 installation scenarios are discussed. The PLC-to-DSL interference environment is discussed in Section 3.2; the methodology used to measure the DM and CM PLCto-DSL cross-coupling channels is discussed in Section 3.3. Finally, the measurement campaign, which is composed of two case studies, is presented in Section DSL Modem Installation Scenarios The DSL modem installation scenarios used in the measurement campaign are presented in this section. Within a house, the DSL copper twisted-pair is terminated by a modem using one of two DSL modem installation scenarios: the Desk Modem Scenario and the Entry Point Scenario. The PLC-to-DSL cross-coupling channels, presented in Section 3.4, are measured for both the Desk Modem Scenario and the Entry point Scenario. Additionally, the performance of the interference mitigations solutions, proposed in Chapter 5, is evaluated for both scenarios. TB LPF Telephone Lines LPF DSL Transceiver Broadband Router Figure 3.1: Desk Modem Scenario. In the Desk Modem Scenario, shown in Fig. 3.1, the DSL modem is located on a desk within the house. The twisted-pair carrying the DSL signal to and from the 49

68 house is connected to a terminal block (TB). The TB is connected to the house s interior telephone wiring, over which the broadband DSL signal is carried through the house to a DSL modem. Low pass filters (LPFs) are installed on each telephone to prevent audio distortion from the DSL signal. TB LPF DSL Transceiver Telephone Lines Broadband Router Figure 3.2: Entry Point Scenario. The Entry Point Scenario, on the other hand, considers the situation where the DSL modem is installed where the telephone cable enters the house, as shown in Fig The signal from the DSL modem is distributed throughout the house either via Wi-Fi, Ethernet, or over the house interior coaxial TV cable to one or more set top boxes. In addition, an in-line LPF is installed after the TB for the household telephone cable so that the DSL signal is blocked from travelling within the house. This makes installing LPFs on each telephone unnecessary. Both scenarios have their own advantages and disadvantages. The Desk Modem Scenario carries the DSL signal deep into the house, which is beneficial if the Internet is carried throughout the house via a WLAN. The Entry Point Scenario, on the other hand, minimizes the EMI suffered by the DSL signal, because the LPF that prevents the DSL signal from travelling within the house over the house s interior telephone 50

69 wiring also blocks the EMI that coupled on the house s interior telephone wiring from affecting the DSL signal. Thus, to reduce the cross-coupling between DSL and in-home PLC networks, the Entry Point scenario is preferred. In addition, installing the DSL transceiver within the house requires the installation of extra LPFs for each telephone outlet to prevent interference to the voice band. However, since most customers utilize wireless routers to carry the Internet throughout the house, terminating the modem within the house, i.e., the Desk Modem Scenario, allows for a better coverage. 3.2 DSL and PLC Interference Environment Power lines are not designed to communicate data at high frequencies; rather, the main purpose of power lines is to supply alternating high voltages at very low frequency. Since power lines are not shielded, portions of the signals transmitted over the power lines will radiate. Fig. 3.3 shows the interference environment between co-located DSL and PLC networks, within the home environment. In this figure, a DSL modem and an inhome PLC-network that operate over the same spectrum co-exist. Note that the DSL modem is installed via the Entry Point Scenario discussed in Section 3.1 and it supplies the house with Internet, while the PLC network forms a LAN that carries the Internet within the house. The power lines within the house form a tree with various branches that connect the various rooms of the house. Any signal transmitted over the power lines travel through all the branches of the power line tree; as the signal travels down the power line tree, portion of it is radiated. Due to the radiation from the various branches of the power line tree, the house is transformed into a large antenna. This radiation is picked up by the copper twisted-pair. As will be confirmed by the measurements 51

70 TB LPF DSL Transceiver Broadband Router Power Lines PLC Transceiver Telephone Lines PLC-DC PLC Transceiver PLC Transceiver Figure 3.3: Co-located DSL and PLC networks interference environment. presented in Section 3.4, this PLC radiation causes significant interference to the DSL network. Similarly, due to imperfections in the twisted-pairs, a portion of the DSL signal is radiated. This radiation is picked up by the power lines; however, since the DSL signal travels from and to the house, the level of DSL-to-PLC interference varies significantly between upstream and downstream DSL transmission. Further detail on the levels of DSL interference on a co-located PLC network is discussed in Section

71 3.3 Measurement Methodology In this section, the methodology behind the channel measurement campaign is described. Recall, the main purpose of the measurements is to characterize the DM and CM PLC-to-DSL cross-coupling channels. Since both DSL and PLC systems utilize multi-carrier FFT based modulation, as discussed in Chapter 2, measuring the crosscoupling channels in the frequency domain is appropriate. The hardware utilized in the measurement campaign is described in Section 3.3.1, while the calibration process is discussed in Section Finally, the measurement setup is introduced in Section Measurement Hardware To study the cross-coupling channels between PLC and DSL networks within the home environment, an Agilent E5071B vector network analyzer (VNA) with an operating range of 300 khz to 8.5 GHz is utilized. One port of the VNA is connected to the twisted-pairs within the house, while the second port of the VNA is connected to the power lines. The scatter matrix between the two ports of the VNA is obtained, which is essentially the cross-coupling channels between the PLC and DSL networks. However, since a VNA has coaxial cable outputs (which are unbalanced with respect to the ground), and both DSL and PLC utilize DM signalling (which is composed of two complementary signals that are balanced with respect to the ground), the unbalanced coaxial ports of the VNA have to be matched to the balanced DSL and PLC networks [14]. To connect the balanced DSL network to the VNA, a balun is used. A balun is essentially a resistance transformer that converts a balanced input to an unbalanced output and vice versa. Fig. 3.4 shows the North Hills 0320BF Balun utilized in the measurement campaign. The 0320BF Balun operates over the range of 10 khz to 30 53

72 Figure 3.4: North Hills 0320BF Balun. MHz. This balun has two 50 Ω unbalanced coaxial ports (labelled J1 and J2 in Fig. 3.4) and two balanced ports with a combined 100 Ω resistance (labelled 1 and 2 in Fig. 3.4). To use the balun in differential mode, it is connected to the VNA via port J1; while for common mode readings, the balun is connected to the VNA via port J2. Note that the twisted-pairs are connected to the Balun via ports 1 and 2, shown in 3.4. Figure 3.5: Northern Microdesign PLC Coupler. Fig. 3.5 shows the Northern Microdesign PLC coupler that operates over the range from 1.8MHz to 39MHz, which is utilized to connect the VNA to the power lines 54

73 within the house. The PLC coupler has two sub-miniature version A (SMA) ports and a three-pin plug to connect to the the power lines via the mains outlets within the house. Both SMA ports are connected to the differential output of the coupler. Thus, to inject a DM signal onto the power lines, the PLC coupler is connected to the VNA via one of the SMA ports. Note that an SMA to BNC (baby N connector) adaptor is required to connect the coupler to the VNA coaxial ports Calibration Balun PLC Coupler 1.5 Gain in db Frequency in MHz Figure 3.6: Insertion loss of the balun and the PLC coupler before calibration. Fig. 3.6 shows the insertion loss due to the balun and the PLC coupler. Thus, before collecting the measurements, the VNA is calibrated to remove the effects of the balun and the PLC coupler. However, before calibration both the Balun and the PLC coupler have to be modified. A registered jack (RJ) is connected to ports 1 and 2 of the balun to facilitate the calibration procedure and promote a more secure connection to the twisted-pairs as shown in Fig.3.7. A female 3-pin socket is soldered to a BNC adaptor to enable the calibration of the PLC coupler as shown in Fig Calibration of the VNA, Balun, and PLC coupler is required to eliminate any 55

74 Figure 3.7: Modified Balun. Figure 3.8: Modified PLC Coupler. effect on the collected measurements. Each port of the VNA is calibrated for four scenarios: open, short, load, and through (between ports). A standard calibration kit provided by the manufacturer is usually utilized in the calibration process; however, since standard calibration kits are designed for the 50 Ω VNA ports, and since both the balun and PLC coupler require a 100 Ω calibration kit, a customized calibration kit is utilized in the calibration process. Three RJ 45 jacks are utilized for the open, short, and load calibration, as shown in Fig The through calibration, shown in Fig. 3.10, is performed via a pair of wires that are connected to a BNC adaptor at one end and to an RJ 45 jack at the other. 56

75 Figure 3.9: Right to Left: Open, Short, and Load (100 Ω). Figure 3.10: Through calibration Measurement Setup After calibration, the scatter matrix between port 1 and port 2 of the VNA is used to determine the PLC-to-DSL cross-coupling channels. One port of the VNA is connected to the power lines of the house through the PLC coupler, while the second port is connected to the telephone wires in the house through the balun, as shown in Fig The VNA is set to output 0 dbm sinusoidal signals and to collect 1500 samples, between frequencies 300 khz and 30 MHz. To measure the DM PLC-to-DSL channel frequency response matrix the balun is connected in the DM mode to the VNA, as shown in Figure 3.11(a). To measure the CM PLC-to-DSL cross-coupling channel, the balun is connected in the CM mode to the VNA, as shown in Figure 3.11(b). 57

76 VNA Port 1 Port 2 PLC adapter Balun Power Network Telephone Network Coupling (a) Setup for measuring DM coupling. VNA Port 1 Port 2 PLC adapter Balun Power Network Telephone Network Coupling (b) Setup for measuring CM coupling. Figure 3.11: Setup for PLC-to-DSL coupling. To study the stationarity of the PCL-to-DSL cross-coupling channels, measurements of the the channels have to be collected over time. An Agilent E5810A general purpose interface bus (GPIB) controller is used to control an Agilent E5071B vector network analyzer (VNA), via MATLAB as shown in Fig Note that the setup shown in Fig is required when consecutive measurements of the channel are needed. If the variation of the channel over time is not a concern, the GPIB is not necessary, and thus, the setup shown in Fig 3.11 is sufficient. In the literature, the background noise on the DM DSL line is usually assumed to be -140 dbm [4]. However, a similar assumption for the CM background noise Hz does not exist. Both the DM and the CM background noise levels are required to study the performance of the adaptive filters utilized by the interference mitigation solutions proposed in this thesis. The background noise present on the CM DSL line was measured with a spectrum analyzer which showed the background noise on the CM DSL channel is approximately -120 dbm Hz. 58

77 GPIB Controller VNA Port 1 Port 2 PLC adapter Balun Power Network Telephone Network Coupling MATLAB Balun Balun Differential Mode Setup Common Mode Setup Figure 3.12: GPIB controlled measurement setup. 3.4 Measurement Campaign The measurement campaign, introduced in this section, is composed of two case studies: Case Study A and Case Study B. The measurements were conducted in two residential test-sites (one site per case study). Since no field measurements have been conducted in a residential house before Case Study A, the case study was designed with two objectives in mind. The first objective is identifying the level of interference between a DSL modem and a colocated PLC network in the home environment. Identifying and determining the relationship between the DM and CM PLC-to-DSL cross-coupling channels within a residential setting is the second objective of Case study A. Note that only the Desk Modem Scenario was considered in Case Study A because the test-site s wiring did not permit measuring the cross-coupling channels for the Entry Point Scenario. From Case Study A, the existence of strong frequency selective cross-coupling channels between the DSL and PLC networks is identified. In addition, it was noticed that the cross-coupling channels were not dependent on the spatial separation between the PLC and DSL modems. However, Case Study A considered only the Desk Modem 59

78 Scenario. In addition, the variation of the cross-coupling channels over time was not studied. Case Study B was a more extensive case study, where a set of 480 DM and CM cross-coupling channel measurements between the DSL and PLC networks is collected, within a residential house. Also, both DSL modem installation scenarios discussed in Section 3.1 are considered in Case Study B. The rest of this section is organized as follows. In Section 3.4.1, the test-sites for both case studies are discussed, while the measurements results are presented in Section Test-Sites The test-sites for the two case studies of the measurement campaign are discussed in this section. In Section , the test-site of Case Study A is presented, while the test-site of Case Study B is discussed in Section Case Study A In this case study, only the Desk Modem Scenario discussed in Section 3.1, is considered. By not terminating the copper twisted-pair at the entrance of the house, the distance over which the twisted-pair DSL line coexists with the PLC power lines increases. This, in-turn, increases the coupling between the two systems. Both DM and CM cross-coupling channels were measured in four rooms, all on the same level, within a 1000 ft 2 residential house. The floor plan of the house is shown in Figure The circle indicates the location of the telephone outlet (i.e., the location of the DSL modem within the house), while the triangles indicates the various locations of the power outlets for which the cross-coupling channels were studied. Note that in Case Study A, the variation of the PLC-to-DSL cross-coupling channels over time is not considered, and thus, the setup shown in Fig 3.11 was utilized. 60

79 40 ft Room D Room B 25 ft Room C Case Study B Room A Telephone Outlet Power Outlet Figure 3.13: Case Study A: test-site floor plan. In Case Study B, a new measurement case study is conducted. The PLC-to-DSL cross-coupling channels within a residential house are measured. Note that the setup shown in Fig was used in this case study. A set of 480 measurements is collected to characterize the DM and CM PLC-to-DSL cross-coupling channels, for both DSL modem installation scenarios discussed in Section 3.1 (i.e., the Desk Modem Scenario and the Entry Point Scenario). The floor plan of the upper and lower floors of the residential single storey bungalow, which is used as a test-site, are shown in Fig. 3.14(a) and Fig. 3.14(b), respectively. The bungalow was built in 1930 and renovated to modern wiring standards in 1992, when the telephony network was professionally installed. Note that the distance between the two floors is approximately 9 ft. Within each room, there are multiple power outlets. Each power outlet where a PLC-to-DSL coupling measurement was collected is labelled. Thus, Room A Plug 1 refers to the PLC-to-DSL cross-coupling channels for Plug 1 in Room A. Note that the location of the DSL modem for the 61

80 Room F Room E Telephone Plug Power Outlet Cable T.V. 48 ft 1 1 Room G 1 Room Room I Room J 1 H DN 92 ft (a) Upper floor. 3 2 Room D 2 Telephone Plug Power Outlet Cable T.V. 42 ft 1 1 Room A Room C Room B UP 72 ft (b) Lower floor. Figure 3.14: Case Study B: test-site floor plan. Desk Modem Scenario and the Entry Point Scenario within the measurement house are in Room J in Fig. 3.14(a) and Room C in Fig. 3.14(b), respectively. For each of the measured power outlets, 10 scatter matrix measurements are collected 0.5 seconds apart. This is performed for both the Desk Modem Scenario and the Entry Point Scenario to study the changes in the CM and DM PLC-to-DSL crosscoupling channels over time. A single set of measurements, which consists of 1500 measurement points, is completed in 0.5 seconds. Both the smoothing (averaging over frequency) and averaging (averaging over time) functions of the VNA were turned off 62

81 during the measurements, and the VNA was not synchronized with the AC cycle. Given that the AC cycle has a duration of 1/60 seconds, over the 10 measurement sets, each measurement point has a different location in the AC cycle. Thus, any variation detected in the PLC-to-DSL cross-coupling channel measurements is attributed to the change in the loads of the power line network Results This section presents the results of the measurement campaign. The frequency responses of the measured DM and CM PLC-to-DSL cross-coupling channels are presented in Section The stationarity of the cross-coupling channels is discussed in Section , while the effect of the spatial separation on the coupling gain and coherent bandwidth of the cross-coupling channels is studied in Section Cross-Coupling Channel Frequency Responses To determine the frequency response matrix for a given channel, the scatter matrix between the ports of the VNA is used. In Section , the cross-coupling channels measured in Case Study A are presented, while the cross-coupling channels measured in Case Study B are discussed in Section Case Study A Fig shows the DM PLC-to-DSL cross-coupling channels, for Rooms A, B, C, and D. From Fig the following is noticed. First, the measurements indicate a very high degree of DM coupling between the PLC and DSL channels. In addition, it also indicates that the DM PLC-to-DSL cross-coupling channels are frequency selective. The figure also shows that the relative distance between the telephone outlet and the power outlet has no significant impact on the PLC-to-DSL coupling. Fig shows the CM PLC-to-DSL cross-coupling channels, respectively, for Rooms A, B, C, and D. Similar to the findings of the DM PLC-to-DSL cross-coupling 63

82 30 40 DM coupling Channel Gain in db Room A 90 Room B Room C Room D Frequency in MHz Figure 3.15: Coupling in Differential Mode. channel measurements, the CM PLC-to-DSL cross-coupling channel measurements show the expected high degree of coupling between the PLC and DSL channels in the common mode. Additionally, Fig indicates that the CM PLC-to-DSL crosscoupling channel is frequency selective and the relative distance between the telephone outlet and the power outlet has no significant impact on the PLC-to-DSL coupling. Finally, by comparing Figs and 3.16, it is noted that the DM coupling is lower than the CM coupling, which is expected since DM transmission is used to mitigate the effects of EMI. The CM to DM transfer function (C2DTF), which is defined as the ratio of the DM to CM cross-coupling channels, is calculated by G(i) = g d(i) g c (i) (3.1) where, G(i) is the C2DTF ratio for frequency bin i, and g d (i) and g c (i) are the DM and CM channel gain vectors for frequency bin i respectively. Fig shows that the C2DTF is quite different for each room. This is due to the tree topology of the power line network within the house. The various branches of the power line network cause either constructive or destructive interference to the DSL network. This tree 64

83 20 30 CM coupling Channel Gain in db Room A 80 Room B Room C Room D Frequency in MHz Figure 3.16: Coupling in Common Mode. also causes the relationship between the DM and CM cross-coupling channels to vary significantly from one power outlet to another. In essence, the topology of the network changes from room to room which causes the variation in the C2DTF Case Study B Fig shows the outlet-to-outlet variation in the DM PLC-to-DSL cross-coupling channels, for the Desk Modem Scenario. Only 5 cross-coupling channels are shown in Fig for the sake of clarity; however, variations among the DM and the CM PLCto-DSL cross-coupling channels for all labelled outlet in Fig have been studied. It was observed that both the DM and CM PLC-to-DSL cross-coupling channels are frequency selective and vary significantly from one power outlet to another, for both the Desk Modem Scenario and the Entry Point Scenario. Note that, as discussed in Section , the frequency selectivity of the cross-coupling channel, and its variation from one outlet to the other, is due to the tree structure of the power lines within the house. Similar to Room A Plug 1, the CM and DM PLC-to-DSL cross-coupling channels for the rest of the labelled power outlets in Fig were measured, for both 65

84 C2DTF Gain in db Room A 40 Room B Room C Room D Frequency in MHz Figure 3.17: Common mode to differential mode transfer function (C2DTF). Desk Modem Scenario and Entry Point Scenario. From these measurements, it was observed that the PLC-to-DSL cross-coupling channels are frequency selective, and vary significantly from one power outlet to another. In addition, the relationship between the DM and the CM cross-coupling channels is not the same for all power outlets. Note that these findings agree with the finding of Case Study A. To study the distribution of the coupling gains of the DM and CM PLC-to-DSL cross-coupling channels, the empirical probability density functions (PDFs) of the DM and CM PLC-to-DSL coupling gains, for both the Desk Modem Scenario and the Entry Point Scenario are generated. To generate the PDF for each PLC-to- DSL cross-coupling channel, a histogram of the gains at all frequency points, for the measured channels, is generated and normalized such that area under the curve tracing the heights of the histogram bins is equal to one. 66

85 30 40 Channel Gain in db Frequency in MHz Figure 3.18: Variation in DM cross-coupling channels from one power outlet to another. Emperical Probability Density Function DM Mean DM CM Mean CM Coupling Gain in db Figure 3.20: Entry Point Scenario coupling gain probability density function. For each scenario, the means of the DM and CM PLC-to-DSL cross-coupling channels, represented by vertical lines in Figs and 3.20, are calculated. Note that, on average, the DM coupling for the Entry Point Scenario is approximately 20 db lower than the DM coupling for the Desk Modem Scenario, which indicates the significant effect of the in-line LPF on the reduction of the DM interference. 67

86 Emperical Probability Density Function DM Mean DM CM Mean CM Coupling Gain in db Figure 3.19: Desk Modem Scenario coupling gain probability density function. However, as will be shown in Chapter 4, this reduction is not sufficient to prevent the PLC system from degrading the performance of the DSL system Coupling Gain in db Desk Modem: Mean DM 90 Desk Modem: Mean CM Entry Point: Mean DM Entry Point: Mean CM Frequency in MHz Figure 3.21: Mean DM and CM PLC-to-DSL coupling. The means of the DM and CM PLC-to-DSL cross-coupling channels for all rooms, for both the Desk Modem Scenario and the Entry Point Scenario, are shown in Fig The mean for each scenario represent the average of the gains for a particular frequency across all the measurement locations shown in Fig. 3.14, i.e., these plots 68

87 show the cross-coupling channels averaged across all outlets used to inject the PLC signal. From Fig. 3.21, it is noted that both the DM and CM PLC-to-DSL crosscoupling channels are frequency selective. Furthermore, it is noted that the in-line LPF present in the Entry Point Scenario reduces the amount of DM coupling relative to the Desk Modem Scenario. However, the CM coupling for both scenarios is not affected by the in-line LPF. In addition, it is evident that the CM coupling is higher than the DM coupling, for both scenarios. This is expected since the DM reception mode partially cancels EMI Stationarity of Cross-Coupling Channels Figure 3.22: Desk Modem Scenario.: DM PLC-to-DSL coupling for Room A Plug 1 over measurement interval. Figs and 3.23 show the changes in the DM PLC-to-DSL coupling for Room A Plug 1 over time for the Desk Modem Scenario and the Entry Point Scenario respectively. Recall, for each power outlet, the PLC-to-DSL cross-coupling channels are measured for 10 consecutive times, with a separation interval of 0.5 seconds, to study the changes in the channel over time. 69

88 Figure 3.23: Entry Point Scenario.: DM PLC-to-DSL coupling for Room A Plug 1 over measurement interval Channel gain in db DM Coupling CM Coupling Frequency in MHz Figure 3.24: Desk Modem Scenario: Room A Plug 1 DM vs. coupling. CM PLC-to-DSL 70

89 Channel gain in db DM Coupling CM Coupling Frequency in MHz Figure 3.25: Entry Point Scenario: Room A Plug 1 DM vs. CM PLC-to-DSL coupling. For each scenario, the average channel gains of the DM and CM PLC-to-DSL cross-coupling channels, for Room A Plug 1, are calculated, and the variations in the channel gains over time is determined by calculating the standard deviation of the channel gains at each frequency point. The average DM and CM PLC-to-DSL crosscoupling channels for Room A Plug 1, for the Desk Modem Scenario and the Entry Point Scenario, are shown in Figs and 3.25 respectively. Also, the variations in the channel gains over time are indicated via error bars in Figs and From Figs. 3.22, 3.23, 3.24, and 3.25 it is noted that the channel gain variations of the PLC-to-DSL cross-coupling channels over time are insignificant. However, over frequency, the variation of the PLC-to-DSL cross-coupling channel gains is quiet significant. Also, it is evident that the CM coupling is higher than the DM coupling, for both scenarios. In addition, it is noted that the DM and CM cross-coupling channels are frequency selective, for both scenarios. Additionally, for each room, the standard deviation for each frequency bin over the 10 measurements is determined; after which, the standard deviation values are 71

90 averaged across all the rooms and then across all the frequency bins. The average standard deviations for the DM and CM PLC-to-DSL cross-coupling channels for the Desk Modem Scenario are 1.06 and 0.98 db respectively, while for the Entry Point Scenario the average standard deviations for the DM and CM PLC-to-DSL coupling are 1.87 and 1.24 db respectively. These numbers indicate that the variation in the channel coupling gains over time is insignificant, and thus, the PLC-to-DSL crosscoupling channel is considered stationary. The measurements were performed in a furnished house, albeit, most of the appliances were not actively drawing power during the measurements. Only the fridge, deep freezer, and coffee maker were on during the measurement campaign. Thus, the test-site was not very electrically active. As will be discussed in Chapter 5, the proposed interference mitigation solutions presented in this thesis are insensitive to changes in the cross-coupling channels due to the non-stationarity of the PLC channel. For a given outlet, any variations in the PLC interference is reflected in both the DM and CM cross-coupling channels simultaneously. Additionally, the proposed interference mitigation solutions utilize adaptive filters that estimate the ratio of the DM to CM PLC-to-DSL cross-coupling channels. Since variations in the PLC interference simultaneously affect both the DM and CM PLC-to-DSL cross-coupling channels, the ratio of the DM to CM PLC-to-DSL cross-coupling channels for a given room remains constant over time Effect of Spatial Separation Fig shows the average DM and CM PLC-to-DSL cross-coupling channel gains versus the relative distance between the DSL and PLC modems for all the measurement locations. Each point represents the cross-coupling channel gain averaged across the measurement frequency range. Note that the Euclidean distance between the three dimensional positions of the DSL modem and the PLC modem is used in 72

91 30 40 Average Coupling Gain in db DM: Desk Modem DM: Entry Point CM: Desk Modem CM: Entry Point Euclidean Distance in ft Figure 3.26: Effect of relative distance between DSL and PLC modems on PLC to-dsl coupling. Fig Information on the actual run length of the cables was unavailable. From this figure, it is noted that the relative distance between the DSL modem and the PLC modem does not have a significant impact on the coupling, since the coupling levels do not decrease as the relative distance between the modems increase. This is likely because the unshielded power lines radiate interference approximately uniformly throughout the home due to the length of the power lines and the relatively small size of the home. Finally, the 90% coherence bandwidth (CB) of both the DM and CM PLC-to-DSL cross-coupling channels versus the Euclidean distance between the PLC and DSL modem is shown in Fig The 90% coherence bandwidth (CB) was determined by calculating the autocorrelation of each channel and determining the width of the bandwidth at which the magnitude of autocorrelation drops to 90% of its maximum value. Table 3.1: Average coherence bandwidth in khz. Scenario DM CM Desk Modem Entry Point

92 Coherence Bandwidth in khz DM: Desk Modem CM: Desk Modem DM: Entry Point CM: Entry Point Euclidean Distance in ft. Figure 3.27: Effect of relative distance between DSL and PLC modems on cross-coupling channels coherence bandwidth. It is noted that, on average, the CB for the DM and CM PLC-to-DSL crosscoupling channels for Desk Modem Scenario is lower than the CB for Entry Point Scenario, as shown in Table 3.1. This due to the fact that by allowing the DSL signal to travel within the house, more branches of the PLC tree contribute (constructively or destructively) to the cross-coupling channels. 74

93 Chapter 4 INTERFERENCE SYSTEM MODEL In Chapter 3, the cross-coupling channels between DSL and PLC networks within a residential setting have been measured. Two case studies have been performed, and among the findings was a strong DM cross-coupling channel between DSL and PLC networks. As a result of this coupling, the mutual interference between two co-located DSL and PLC networks, operating over the same frequency band, will inevitably affect the performance of both systems. Thus, an interference mitigation solution is required. In this chapter, we quantify the effect of the mutual DSL and PLC interference on the data rates of existing DSL and PLC systems. The current system model, which describes the interference environment between PLC and DSL systems as it exists today, is discussed in Section 4.1. The data rates achieved by a DSL modem in presence of an in-home PLC network and the data rates achieved by an in-home PLC network in presence of a DSL modem are studied in Section Current System Model Within a residential house, the DSL and PLC systems utilize the twisted-pairs and the power lines, respectively, as transmission mediums. The twisted-pairs are composed of two twisted copper wires, designed to minimize both EMI egress and ingress. On the other hand, power lines are made of two or three wires. The power lines are neither twisted nor shielded, which result in significant electromagnetic radiation. Fig. 4.1 shows the current system model for the PLC-to-DSL interference environment within a residential setting. In each house, there is a single DSL transceiver 75

94 DSL system Twisted-pair DSL Transceiver Balun DMT Transceiver Wide-band EMI PLC-D.C. Power Lines PLC Transceiver PLC Transceiver PLC Transceiver PLC Coupler DMT Transceiver PLC system PLC Transceiver Figure 4.1: Current system model. while there are multiple PLC transceivers. The measurement campaign presented in Chapter 3 indicates that the cross-coupling channels between the DSL and PLC systems vary significantly from one PLC transceiver to the other. The DSL transceiver is connected to the copper twisted-pairs, over which the DSL signal is carried to and from the house. The DSL transceiver is composed of a balun that outputs a DM signal, which is then passed to a DMT transceiver, as shown in Fig Similarly, the PLC transceivers form a network through the power lines which run through the house. Each PLC transceiver is composed of a PLC coupler that feeds its output to a DMT transceiver, as shown in Fig Note that the 76

95 structure of the DMT transceiver was discussed in Section Effect of Mutual DSL and PLC Interference on Bit Rates The measured cross-coupling channels presented in Chapter 3 are utilized to visualize the effect of the PLC interference on the received DSL signal in Section 4.2.1; while, the degradation of the PLC system performance due to the DSL interference is studied in Section Note that the measured cross-coupling channels are reciprocal; however, as will be shown in the following sections, the effect of the PLC and the DSL networks on one another is not Effect of PLC Interference on DSL Bit Rates To determine the effect of the PLC interference on the received DSL signal, the measured PLC-to-DSL cross-coupling channels are used to calculate the power spectral density (PSD) of the PLC interference. The PSD of the PLC interference is then compared with the PSD of a typical received DSL signal. Note that the direct DSL channel, used to determine the PSD of the received DSL signal, is obtained from the standard two-port model defined in [29]. Similarly, the average of the measured DM PLC-to-DSL cross-coupling channels, presented in Section 3.4 and shown in Fig. 3.21, is utilized to determine the PSD of the PLC interference. The PSDs of the received DSL signal for various DSL cable run lengths versus the PSD of the PLC interference, for both the Desk Modem Scenario and the Entry Point Scenario, are shown in Fig The maximum transmit power of both the DSL and PLC systems, as specified in their respective ITU-T standards, is multiplied by the DSL channel and the mean DM PLC-to-DSL cross-coupling channel to determine the PSD of the signals. For VDSL2, the maximum transmit power is -50 dbm [2]; while Hz for in-home PLC networks, the maximum allowed transmit power is -60 dbm Hz [9]. 77

96 PSD in db m Hz DSL: Recieved at 2kft 200 DSL: Recieved at 4kft PLC: Desk Modem Scen. PLC: Entry Point Scen Frequency in MHz Figure 4.2: Received DSL Signal versus PLC interference. Fig. 4.2 illustrates two points. First is the extremely high level of PLC interference on the DSL line for both the Desk Modem Scenario and the Entry Point Scenario. Although, in the Entry Point Scenario, the in-line LPF reduces the PLC interference levels, this reduction is not sufficient to prevent the DSL system from experiencing low signal to interference plus noise ratio (SINR) over the VDSL2 spectrum. The second observation is, while the PLC interference levels are unaffected by the distance between the central office (or distribution point) and the house, the received DSL signal power decreases as the distance increases. Thus, the PLC interference does not only impact VDSL2 services, but also it poses a significant risk to DSL services that utilize narrower spectrum, such as ADSL2+ (up to 2.2 MHz [39]) and VDSL (up to 12 MHz [40]), as the DSL cable run increases. Bit loading, where the number of bits allocated to each sub-channel is dependent on the sub-channel s SINR, is used in both DSL and PLC [22]. The number of bits b(i) that can be loaded on frequency bin i is calculated by ( ( b(i) = min b max, log γ(i) ) ), (4.1) Γ where γ(i) is the SINR of frequency bin i, Γ is the SNR gap, and b max is the maximum 78

97 number of bits that can be allocated to a frequency bin. After determining the number of bits that can be loaded on each frequency bin, the total bit rate R is then calculated by R = λ f i b(i), (4.2) where λ is the normalizing factor and f is the sub-carrier spacing [27]. Note that the value of b max, λ, and f varies from one system to the other. In presence of AWGN with PSD of N o = -140 dbm, the signal to the interference plus noise power is used Hz to determine the available bits that can be loaded to each of the DSL sub-channels via (4.1). After which, from (4.2), the total number of bits is determined. Note that for VDSL2, b max =15 bits, Γ=9.45 db, λ=0.79, and f DSL =8.6 khz [27]. Recall, DSL utilizes FDD, where the sub-channels are grouped and the groups are allocated to US and DS transmission in an alternating fashion. The number of sub-channels per transmission direction varies from one band plan to the other. Note that band plan 998E30 [2], shown in Fig. 4.3, is utilized in this thesis. Downstream Upstream Figure 4.3: US and DS frequencies for band plan 998E30 [2]. Fig. 4.4 shows the available DSL bit rates vs the length of the DSL cable run lengths. Six plots are shown in Fig. 4.4, where Upstream and Downstream indicate 79

98 the available bit rates for US and DS transmission direction respectively, based on the bandwidth allocation shown in Fig Bound DSL denote the maximum available bit rates in presence of AWGN only, i.e., in absence of PLC interference. Similarly, Desk Modem and Entry Point indicate the available bit rates in presence of both AWGN and PLC interference for both the Desk Modem Scenario and the Entry Point Scenario, respectively. 250 Available Bit rates in Mbit/s Upstream: Bound DSL Downstream: Bound DSL Upstream: Desk Modem Downstream: Desk Modem Upstream: Entry Point Downstream: Entry Point DSL cable run length in ft Figure 4.4: Available DSL bit rates. From Fig. 4.4, it is noted that the PLC interference effect on the DS transmission is higher than the PLC interference effect on the US transmission. This is expected since in DS transmission the DSL signal is attenuated as it travels from the central office to the house over the twisted pairs, and thus, it arrives at low power levels at the house. The data rate loss percentage is calculated to determine the loss percentage in each case. The loss percentage Loss % is calculated by Loss % = R Bound R Interference R Bound 100, (4.3) where R Bound is the total number of bits in presence of background noise only and 80

99 R Interference is the total number of bits in presence of interference plus background noise. Fig. 4.5 shows the degradation in DSL bit rates in terms of data rate loss percentage. From this figure, it is note that DSL suffers up to 83% and 52% loss in available data rates for DS transmission for the Desk Modem Scenario and the Entry Point Scenario respectively. While, on the other hand, the highest loss in available data rates for US transmission are 38% and 3% for the Desk Modem Scenario and the Entry Point Scenario respectively. Note that for US transmission, the receiver is at the central office; thus, the PLC interference is attenuated along with the desired signal as it travels over the twisted pair Bitrate Loss Percentage Upstream: Desk Modem Downstream: Desk Modem Upstream: Entry Point Downstream: Entry Point DSL cable run length in ft Figure 4.5: Degradation in DSL bit rates. Another observation from Fig. 4.5 is that the impact of the PLC interference on the DS DSL available data rates depends significantly on the DSL cable run length. For instance, the reduction in DS DSL bit rates for the Entry Point Scenario is 8% for a DSL cable run length of 200 ft, which increases to 52% for a DSL cable run length of 1600 ft, and then decreases to 31% for a DSL cable run length of 2600 ft. This is explained as follows. For short DSL cable run length, the SINR is very large; thus, the limiting factor on the number of bits that can be loaded onto each sub-channel 81

100 is b max in (4.1). However, as the signal power decreases due to the attenuation from travelling down the twisted-pairs, the PLC effect becomes more prominent, which limits the available bit rates. As the DSL cable run length increases, the signal power decreases significantly, especially at high frequency. This makes the AWGN the limiting factor on the number of bits that can be loaded to each sub-channel. Finally, it is noted that the impact of the PLC interference on the US DSL data rates does not vary significantly with the DSL cable run length. This is because both the DSL signal and the PLC interference originate from the house Effect of DSL Interference on PLC Bit Rates To study the impact of the DSL network on a co-located in-home PLC network, the same approach used in Section is utilized. However, unlike the PLC signal that is confined to the house, the DSL signal travels to and from the house (i.e., DS and US traffic). Thus, the DSL interference impact on the received PLC signal has to be studied for both traffic directions. Note that the US and DS frequencies are based on the band plan 998E30 [2], shown in Fig PSD in dbm Hz PLC Desired Signal DSL: Desk Modem Received at 2kft DSL: Desk Modem Received at 4kft DSL: Entry Point Received at 2kft DSL: Entry Point Received at 4kft Frequency in MHz Figure 4.6: Received PLC Signal versus DS DSL interference. 82

101 Fig. 4.6 shows the PSD of the received PLC signal versus the interference caused by DS traffic for various DSL cable run lengths, for both the Desk Modem Scenario and the Entry Point Scenario. Note that the effective DM DSL-to-PLC cross-coupling channels and the direct PLC channel are utilized in the analysis presented in this section. The effective DM DSL-to-PLC cross-coupling channels are obtained by multiplying the direct DSL channel defined in the two-port model in [29] by the mean DM DSL-to-PLC cross-coupling channels shown Fig Note that the DM crosscoupling channels between the DSL and PLC systems are reciprocal. The direct PLC channel is obtained from the model defined in [66]. From Fig. 4.6 it is noted that the DS DSL interference is below the DM noise floor (i.e., below -140 dbm ) for Entry point scenario; thus, it is negligible. However, Hz for Desk Modem scenario, the DS DSL interference for short DSL cable run length might affect some of the PLC sub-channels. This degradation will be more noticeable if the PLC transmit power is reduced to mitigate the PLC-to-DSL interference, which is the solution proposed by [27, 28]. As will be shown in Chapter 5, the solutions proposed in this thesis do not have this negative impact on the PLC network. Fig. 4.7 shows the PSD of the received PLC signal versus the interference cause by US traffic for both the Desk Modem Scenario and the Entry Point Scenario. Note that US DSL traffic is not affected by the DSL cable run length; thus, the mean DM DSLto-PLC cross-coupling channel is multiplied by the DSL maximum transmit power to simulate the effect of the US DSL interference on the received PLC signal. From Fig. 4.7, it is noted that interference from the US DSL traffic on the received PLC Signal is significantly higher than the DS DSL interference shown inf Fig Similar to what was performed in Section 4.2.1, the ratio of the PLC signal power to the DSL interference power is translated into achievable bit rates to quantify the effect of the DSL interference on the PLC bit rates. From (4.1), the SINR is used 83

102 60 80 PLC Desired Signal DSL: Desk Modem DSL: Entry Point 100 PSD in dbm Hz Frequency in MHz Figure 4.7: Received PLC Signal versus US DSL interference. to determine the available bits that can be loaded to each of the PLC sub-channels. Note that it is assumed that the AWGN on the PLC DM channel has a PSD of N o = -140 dbm. The total number of bits that can be transmitted is then determined Hz from (4.2). For PLC, b max =12 bits, λ=0.75, and f PLC =24.4 khz. Recall, PLC utilizes TDMA, where the channel time is divided into slots and only a single PLC user can transmit at a time. However, since DSL utilizes FDD, the effect of the DSL interference for both US and DS transmission, for band plan 998E30 [2] shown in Fig. 4.3, is considered. Fig. 4.8 shows the available DSL PLC rates versus the length of the DSL cable. Five plots are shown in Fig Upstream and Downstream indicate the available PLC bit rates in presence of US and DS DSL interference respectively. Bound PLC, Desk Modem, and Entry Point denote the available bit rates in presence of AWGN only, in presence of both AWGN and DSL interference for Desk Modem Scenario, and in presence of both AWGN and DSL interference for Entry Point Scenario, respectively. 84

103 Available Bit rates in Mbit/s Bound PLC Upstream: Desk Modem Downstream: Desk Modem Upstream: Entry Point Downstream: Entry Point DSL cable run length in ft Figure 4.8: Available PLC bit rates. From Fig. 4.8, it is noted that the DS DSL interference effect on the PLC signal is lower than the US DSL interference effect on the PLC signal. This is due to the attenuation of the received DSL signal as it travels down the length of the copper twisted-pairs, and thus, it arrives at low power levels at the house Bitrate Loss Percentage Upstream: Desk Modem Downstream: Desk Modem Upstream: Entry Point Downstream: Entry Point DSL cable run length in ft Figure 4.9: Degradation in PLC bit rates. 85

104 This is confirmed by Fig. 4.9, where the DS DSL interference effect on the PLC signal results in a loss in bitrates below 5%, except for short DSL cable run lengths (less than 600 ft). On the other hand, US DSL interference causes a 20% loss in the available PLC data rates for Desk Modem scenario for all DSL cable run lengths. Note that the loss percentage is calculated by (4.3). From the analysis conducted in this section and Section 4.2.1, it is concluded that there is significant potential for PLC networks to degrade DSL performance in a home environment, regardless of whether or not an in-line LPF is used for the home telephone wiring. On the other hand, the interference from the DSL network on the PLC network is not as significant; albeit, potential for PLC network performance degradation due to a co-located DSL network does exit. This motivates the need for the interference cancellation solutions presented in this Chapter 5. 86

105 Chapter 5 INTERFERENCE MITIGATION SOLUTIONS Both DSL and PLC systems are subject to various sources of interference such as radio frequency interference from amateur radios, impulsive noise, etc.; however, only mutual interference between the DSL and the PLC systems, in presence of thermal noise, are considered in this thesis. As discussed in Chapter 3, field measurements indicate the presence of significant cross-coupling levels between DSL and PLC. This level of coupling negatively impacts the performance of both DSL and PLC systems, as was confirmed in Chapter 4. In this chapter, we introduce our proposed interference mitigation schemes. The rest of this chapter is organized as follows. The modified system model for the interference environment between DSL and in-home PLC networks is presented in Section 5.1, and in Section 5.2, the interference mitigation block diagram is introduced. Based on this block diagram, two interference mitigations solutions are presented. A scheduling-based interference mitigation solution is presented in Section 5.3; while, a pre-distortion-based interference mitigation solution is discussed in Section 5.4. Finally, in Section 5.5, the performance of the proposed interference mitigation solutions are compared with the performance of the spectral management solutions proposed in [27] and [28]. 5.1 Modified System Model In Chapter 4, the current system model of the interference environment between a DSL modem and an in-home PLC networks have been discussed. In this section, a modified system model is introduced. Recall, within a residential house, the DSL 87

106 modem (transceiver) is connected to the distribution point, outside the house, via a twisted-pair. PLC transceivers, on the other hand, are connected to each other via the power lines within the house. Traditionally, both networks are not connected; however mutual wide-band EMI interference affects their performances as was shown in Section 4.2. The rest of this Section is organized as follows. The system model, which is the basis of the proposed interference mitigation solutions, is presented in Section Time variations in the PLC interference and its effect on the estimate of the DM to the CM cross-coupling channels ratio is discussed in Section Finally, integration of the FDIC within the structure of the DMT tranceiver is shown in Section Proposed System Model Both DSL and PLC utilize DM transmission, where complementary signals are inserted via the transmitter over the transmission medium, and the difference between the signals is obtained at the receiver as the desired signal. The CM signal, on the other hand, is the arithmetic mean of the complementary signals. If EMI couples identically on each of the wires of either the twisted-pairs or the power lines, the DM signal will only contain the transmitted signal, while the CM signal will be composed of only EMI. Since the CM signal contains no useful information about the signal, it is usually ignored at the receiver. In practical situations, EMI does not couple identically on the wires, and thus, the DM signal is composed of the transmitted signal and interference, while the dominant component of the CM signal is the interference. Since the CM signal contains information about the EMI, we propose utilizing this information to mitigate the effect of the EMI on the DM signal. Fig. 5.1 shows the system model for our proposed scheme. In each house, there is a single DSL transceiver while there are multiple PLC transceivers. One of the PLC transceivers is assigned as a PLC-DC. This thesis proposes that the PLC transceiver 88

107 DSL system Balun DSL Transceiver DM Signal DMT Transceiver Twisted-pair CM Signal FDIC PLC-D.C. Power Lines PLC Transceiver PLC Transceiver PLC Transceiver PLC Coupler DM Signal DMT Transceiver PLC system CM Signal PLC Transceiver FDIC Figure 5.1: Proposed system model. closest to the DSL modem takes over as the PLC-DC. Note that the PLC-DC is connected to both the twisted-pair carrying the DSL signal and to the DSL transceiver FDIC. Further detail on the functionality of these connections are presented in Sections 5.3 and 5.4. Frequency domain interference cancellers (FDICs) are utilized by the interference mitigation solutions presented in Sections 5.3 and 5.4. As shown in Fig. 5.1, there is one FDIC per transceiver. The main role of the FDIC is to extract an estimate 89

108 of the DM interference from the CM signal. Each canceller is composed of multiple single-tap filters. The number of single-tap filters per FDIC equals the number of subcarriers. Thus, the FDIC utilized by the DSL system has more single-tap filters than the FDICs utilized by the PLC system because DSL systems utilize more sub-carriers than a PLC system for a given bandwidth. The tap coefficients of each FDIC are an estimate of ratio of the DM cross-coupling channel to the CM cross-coupling channel. This ratio is utilized to estimate the DM interference affecting the desired signal from the CM signal. This interference signal estimate is the product of the CM signal and the ratio of the DM cross-coupling channel to the CM cross-coupling channel. Thus, the FDIC installed in the DSL transceiver is utilized to estimate and mitigate the PLC interference on the DSL system; while, the FDIC installed in a PLC transceiver estimates and mitigates the DSL interference on the PLC system. Note that the FDICs have to be trained to obtain the ratio of the DM interference to the CM interference. The procedure used to train the tap-coefficients of the FDICs depends on the utilized interference mitigation solution. In this thesis, FDIC training is discussed in Sections 5.3 and 5.4. The modified DSL transceiver is composed of a balun, an FDIC, and a DMT transceiver; while, the modified PLC transceivers are composed of a PLC coupler, an FDIC, and a DMT transceiver, as shown in Fig The balun outputs two signals: DM signal and CM signal. The CM signal is passed through the FDIC to obtain an estimate of the DM interference. This estimate is then subtracted from the DM signal, before it goes through the DMT transceiver to retrieve the desired DSL signal. Similarly, the PLC coupler converts the complementary signals transmitted over the power lines into a DM signal and a CM signal. From the CM signal, an estimate of the DM interference is obtained via the FDIC. This estimate is subtracted from the DM signal; then, the resultant is then demodulated by the DMT transceiver to 90

109 obtain the desired PLC signal Variations in the DM to CM Estimated Ratio After training the FDIC, the weights of its tap-coefficients remain valid until the PLC user generating the interference changes. The measurements discussed in Chapter 3 indicate that the PLC-to-DSL cross-coupling channels are stationary when few appliances were drawing power. In the event that more appliances are actively drawing power from the power lines the PLC-to-DSL cross-coupling channels might vary significantly within a DSL super-frame. However, despite the non-stationarity of the PLC-to-DSL cross-coupling channels, the FDIC will still operate properly because the FDIC tap-coefficients are an estimate of the ratio of the DM interference to the CM interference. k.s Balun DM Interference s.(k-j) s Interference j.s CM Interference s.(k+j)/2 Figure 5.2: Interference coupling on twisted-pair. Fig. 5.2 illustrates how the radiated interference is coupled on a DSL twistedpair, where s is the interference radiated from the power lines, and k and j are the imbalance factors on the twisted-pair wires. Any variation in the PLC cross-coupling channel would manifest as a variation in s, and any variation in s will change both DM and CM interference by the same amount. As a result, the variation would cancel in the ratio of DM to CM interference which, in Fig. 5.2, would be s(k j) divided by s(k + j)/2. Note that the PLC channel is a known for its non-stationarity. In addition to the PLC interference, impulsive noise on the power lines cause by in-home appliances will couple through the cross-coupling channels. As discussed in Section 2.3, the PSD of 91

110 the impulsive noise is -105 dbm, while the transmit power of the PLC signal is -50 Hz dbm. Thus, the dominant interferer to DSL in the home environment is the PLC Hz signal. In this thesis, the PLC noise sources are not considered in the analysis of the PLC-to-DSL interference mitigation solutions Integration of the FDIC DSL Symbols DSL Signal Direct DSL Channel AWGN Filter & Remove CP S/P FFT FEQ QAM Decoder P/S Bit Stream DM PLC-to-DSL Channel + - PLC Symbols AWGN CM PLC-to-DSL Channel PLC Interference Filter & Remove CP S/P FFT FDIC Figure 5.3: Integration of the FDIC into a DMT transceiver. To accommodate the FDIC, a modified version of the DMT transceiver discussed in Section is shown in Fig As shown in this figure, the FDIC can be easily integrated into the current DSL modems. The CM signal is passed through a receiver filter, where the out-of-band noise is minimized and the signal is converted to digital via a digital to analog converter. After that, the cyclic prefix is removed and the serial samples are converted to a set of parallel sub-symbols, which are fed to an FFT block. Up to this point, both the DM and CM signal paths have the same blocks. At the CM signal path, the output of the FFT is fed to the FDIC, whose output is subtracted from the output of the FFT block utilized by the DM signal. The result of the subtraction, i.e., the estimate of the desired DSL signal, is passed through an FEQ. The output of the FEQ is an estimate of the transmitted DSL symbols, which 92

111 are then fed to a QAM decoder to determine the transmitted bits. By comparing Figs. 2.4 and 5.3, it is noted that most of the modification occurs in the CM signal path. 5.2 Interference Mitigation Block Diagram For the rest of this chapter, a boldface lower case letter will indicate a vector, while a boldface upper case letter will denote a matrix. When referring to the DSL system, all vectors are M 1 column vectors and all matrices are M M square diagonal matrices, where M is the number of DSL sub-carriers. Similarly, all vectors and matrices that belong to the PLC system are N 1 column vectors and N N square diagonal matrices respectively, where N is the number of PLC sub-carriers. When needed, a superscript is added to variables to indicate whether variables belongs to the DSL or the PLC system. The notation x(i) indicates the element in the i th row of the column vector x. Similarly, the notation H(i, j) refers to the element in the i th row and the j th column of the square diagonal matrix H. To indicate the complex conjugate of u, the notation u will be used. Finally, the notation ê and e will be utilized to indicate the estimate and the average of e, respectively. DSL and PLC utilize bit loading on orthogonal sub-carriers, based on the subcarrier s signal to interference plus noise ratio (SINR). The frequency separation between PLC sub-carriers f PLC is usually greater than the frequency separation between DSL sub-carriers f DSL [9]. Thus, over a given spectrum, the number of PLC sub-carriers is smaller than the number of DSL sub-carriers. This means the cross-coupling channel is sampled at slightly different frequencies by the DSL and PLC systems. The cross-coupling channels between the DSL and the PLC systems are shown in Fig The DM and CM cross-coupling channels between a PLC transmitter 93

112 x H DSL y η d,dsl r d,dsl ^y v d,dsl ^v d,dsl v c,dsl r c,dsl C DSL Δf H DSL d,ptx-drx Δf H DSL c,ptx-drx η c,dsl Δf H PLC c,dtx-prx Δf H PLC d,dtx-prx DSL system Interference Channels q H PLC z v d,plc PLC system r d,plc z^ v c,plc η d,plc ^ v d,plc η c,plc r c,plc C PLC Figure 5.4: Interference cancelling scheme block diagram. and the DSL receiver, sampled at integer multiples of f DSL, are denoted by the channel frequency response matrices H f DSL d,ptx DRx and H f DSL c,ptx DRx. A signal transmitted through these channels correspond to the electromagnetic waves radiated by a PLC transceiver, which then couples onto the twisted-pair connected to a balun that outputs a DM and a CM signal, as shown Fig Note that since the DM and CM PLC-to-DSL cross-coupling channels are sampled at integer multiples of f DSL, both H f DSL d,ptx DRx and H f DSL c,ptx DRx are M M square diagonal matrices. 94

113 Similarly, the DM cross-coupling channel between the PLC transmitter and the DSL receiver when sampled at integer multiples of f PLC is defined by the channel frequency response matrix H f PLC d,ptx DRx, which is an N N square diagonal matrices. Note that while both H f DSL d,ptx DRx and H f PLC d,ptx DRx define the same channel, these two matrices are not equal due do the difference in the sampling frequencies (i.e., f DSL f PLC ). The channel frequency response matrices H f PLC d,dtx PRx and H f PLC c,dtx PRx define the DM and CM cross-coupling channels between the DSL transmitter and the PLC transceiver, when sampled at integer multiples of f PLC. Note that since the DM and CM DSL-to-PLC cross-coupling channels are sampled at integer multiples of f PLC, both H f PLC d,dtx PRx and H f PLC c,dtx PRx are N N square diagonal matrices. Fig. 5.4 also shows the block diagram for the DSL and the PLC systems. The desired DSL signal at the customer s premise is y, which is given by y = H DSL x, (5.1) where x is the transmitted DSL symbol and H DSL is the channel frequency response matrix for the direct DSL channel. Note y is received at the house via the twisted-pair shown in Fig Similarly, the desired PLC signal is z and is defined by z = H PLC q, (5.2) where H PLC is the channel frequency response matrix for the direct PLC channel and q is the transmitted PLC symbol. Note y is transmitted within the house from the PLC transmitter to the PLC receiver via the power lines shown in Fig The PLC interference is composed of the transmitted PLC symbol q radiated within the house and picked by the twisted-pair. At the DSL receiver, the DM PLC interference v d,dsl can be seen as the signal q transmitted through the differential 95

114 PLC-to-DSL cross-coupling channel H f DSL d,ptx DRx. Similarly, the CM PLC interference v c,dsl is the signal q transmitted through the common mode PLC-to-DSL crosscoupling channel H f DSL c,ptx DRx. Thus, the DM PLC interference v d,dsl and the CM PLC interference v c,dsl are defined by (5.3a) and (5.3b) respectively. The DM and CM PLC interference corresponds to the interference component of the DM and CM DSL signals at the output of the balun shown in Fig v d,dsl =H f DSL d,ptx DRx q v c,dsl =H f DSL c,ptx DRx q (5.3a) (5.3b) Moreover, the DSL interference is composed of the received DSL symbol y radiated within the house and picked up by the power lines that form the in-home PLC network. At the PLC receiver, the DM PLC interference v d,plc (shown in (5.4a)) is the signal y transmitted through the differential DSL-to-PLC cross-coupling channel H f PLC d,dtx PRx, while the CM PLC interference v c,dsl (shown in (5.4a)) is the signal y transmitted through the common mode DSL-to-PLC cross-coupling channel H f PLC c,dtx PRx. Both v d,dsl and v c,dsl are the interference component of the DM and CM PLC signals at the output of the PLC coupler shown in Fig v d,plc =H f PLC d,dtx PRx y v c,plc =H f PLC c,dtx PRx y (5.4a) (5.4b) The received DSL DM signal r d,dsl, shown at the output of the balun in Fig. 5.1, is composed of the desired DSL signal y, the DM PLC interference v d,dsl, and the DM additive white Gaussian noise η d,dsl, as shown in (5.5a). The received DSL CM signal r c,dsl, also shown at the output of the balun in Fig. 5.1, is composed of only 96

115 the CM PLC interference v c,dsl and the CM additive white Gaussian noise η c,dsl, as shown in as shown in (5.5b). r d,dsl =y + v d,dsl + η d,dsl r c,dsl =v c,dsl + η c,dsl (5.5a) (5.5b) Similarly, the received DM PLC signal r d,plc, which is given in (5.6a), is the summation of the desired PLC signal z, the DM DSL interference v d,plc, and the DM additive white Gaussian noise η d,plc. However, the received CM PLC signal r c,plc, shown in (5.6b), is composed of only the CM DSL interference v c,plc and the CM additive white Gaussian noise η c,plc. Both r d,plc and r c,plc are the DM and CM PLC signals at the output of the PLC coupler in Fig r d,plc =z + v d,dsl + η d,dsl r c,plc =v c,dsl + η c,dsl (5.6a) (5.6b) The tap-coefficients matrices of the adaptive frequency domain interference cancellers C-DSL and C-PLC are shown in Fig Note that C-DSL and C-PLC correspond to the FDICs utilized by the DSL and PLC systems, shown in Fig. 5.1, respectively. The tap-coefficients of the canceller C-DSL are defined by the M M square diagonal matrix C DSL, while the tap-coefficients of the canceller C-PLC are defined by the N N square diagonal matrix C PLC. The estimate of the DM PLC interference v d,dsl is extracted from r c,dsl via C DSL as shown in (5.7a), while the estimate of the DM DSL interference v d,dsl is extracted from r c,plc via C PLC as shown in (5.7b). 97

116 v d,dsl =C DSL r c,dsl = C DSL ( vc,dsl + η c,dsl ) v d,plc =C PLC r c,plc = C PLC ( vc,plc + η c,plc ) (5.7a) (5.7b) The canceller C-DSL is utilized to mitigate the effects of v d,dsl on r d,dsl. The signal ŷ in Fig. 5.4, which is the difference between the received DSL signal r d,dsl and the output of the canceller v d,dsl, is defined by (5.8a). Similarly, the canceller C-PLC is utilized to mitigate the effects of v d,plc on r d,plc. The signal ẑ in Fig. 5.4 is the difference between the received PLC signal r d,plc and the output of the canceller v d,plc, as shown in (5.8b). ŷ =r d,dsl C DSL r c,dsl ẑ =r d,plc C PLC r c,plc (5.8a) (5.8b) The manner in which the tap-coefficients of the FDICs are trained, in this thesis, vary according to the mitigation solution in which they are utilized. In Sections 5.3 and 5.4, two interference mitigation solutions are presented. For each solution, the FDICs are trained and their abilities to negate the effect of the cross-coupling channels are evaluated. 5.3 Scheduling-Based Interference Mitigation Solution In this section, the first interference mitigation solution, which is based on scheduling the PLC users access to the PLC channel along with utilizing an adaptive FDIC, is introduced. The medium access techniques used by both the DSL and PLC networks, as well as the proposed modification to the PLC medium access, is discussed in Section The proposed cancellation algorithm utilized to mitigate the PLC-to- 98

117 DSL interference is presented in Section 5.3.2; while, its performance in mitigating the PLC-to-DSL interference is analyzed in Section sub:performanalysisi. Note that this interference cancellation scheme does not require modification to either G [2] or G.hn [9], which are the ITU standards for VDSL2 and home networks respectively. Although, the proposed cancellation scheme would require some changes in the silicon design of existing DSL and PLC customer premise equipment, these changes are limited the receiver section of the DSL modem and the PLC domain controller. The necessary modifications to the DSL and PLC transceivers have been shown in Fig Medium Access The frame structure of both the DSL and the PLC networks are discussed in this section. This section also describes how the PLC-DC schedules the transmissions of PLC users in order to facilitate interference cancellation. Data is transmitted over the DSL channel in super-frames. For VDSL2, a superframe has a duration of T SF = ms [2]. The super-frame is divided into 257 frames, the last of which is the synchronization frame [2]. Since each DSL user utilizes a dedicated physical link, all super-frames on the DSL channel belong to a single user. The PLC channel, on the other hand, is a shared medium divided into time-slots of duration T P [9]. Only a single PLC user can transmit per time-slot. One of the PLC users takes over as a PLC-DC and allocates time-slots for various users, based on the quality of service requirements of each user. In order for the FDIC to successfully mitigate the effects of the PLC interference, the PLC network needs to restrict channel access so only one PLC user is allowed to transmit per DSL super-frame. It was shown in Chapter 3 that the PLC-to-DSL cross-coupling channels vary significantly from one outlet to the next. Thus, the 99

118 trained FDIC tap-weights are only viable for a specific PLC user. As a result, the FDIC has to be retrained to compensate for the change in the cross-coupling channel each time a different user starts to transmit. As will be discussed in the next section, the FDIC can only be trained during the super-frame sync symbols. Thus, it is essential that only the PLC user for which the FDIC was trained transmits during a given super-frame. To prevent more than one PLC user from transmitting during a given super-frame, the PLC-DC is connected to the DSL line, as shown in Fig The PLC-DC does not decode the full super-frame but does synchronize itself to it. It then schedules the PLC users so that only one user transmits per DSL super-frame. The retransmission and acknowledgement protocol of ITU-T G.hn [9] standard specifies two acknowledgement techniques for uni-cast transmission, namely: immediate acknowledgement (Imm-ACK) and delayed acknowledgement (delayed-ack). When Imm-ACK is specified by the transmitter, the receiver is required to transmit an Imm-ACK frame after the successful reception of a given frame; when delayed- ACK is required, the receiver delays the acknowledgement frame until it is assigned a transmission opportunity (TXOP). In this thesis, it is assumed that delay-ack procedure is employed by the PLC network, which would allow a DSL super-frame duration to hold both the current PLC data frame and the ACK frame for a previous transmission. Thus, an entire super-frame duration is not dedicated to a relatively short ACK frame. Given a delayed-ack procedure, the latency experienced by a PLC user in our proposed solution is dependent on the number of PLC users sharing the PLC network and on the duration of the DSL super-frame. Assuming a round-robin style of access, the latency experienced by a user is T LAT,Scheduling = (n 1)T SF, where n is the number of users and T SF =64.25 ms [2]. Using this simple model, it is possible to satisfy the 100

119 latency requirements for the multimedia applications in [67], which is between 100 and 300 ms, with up to 5 active PLC transceivers, as shown in Table 5.1. Note D is the number active PLC users. Table 5.1: Latency: Scheduling-based solution. D (active users) T LAT,Scheduling (ms) It is possible that this PLC scheduling scheme may reduce PLC throughput if the transmission from a particular user is much shorter than a DSL super-frame. However, this loss in throughput will be more than made up by allowing PLC networks full use of the DSL operating spectrum. Furthermore, while VDSL2 has a super-frame duration of ms [2], it is expected that the super-frame duration for G.Fast will be less than 10 ms [8] Cancellation Algorithm The interference cancelling scheme is composed of two phases: training phase and cancellation phase. The objective of the training phase is to determine a ratio between the differential mode interference and the common mode interference. This ratio allows the FDIC to use the CM signal to estimate the DM interference component in order to subtract it off of the desired signal. As discussed in Section 5.2, the C-DSL and C-PLC are utilized to estimate a ratio between the DM and CM cross-coupling channels. The measurements in Chapter 3 show that the relationship between DM and CM cross-coupling channels is very frequency selective. However, since the FDICs processes each of the sub-channels 101

120 independently, this frequency selectivity does not affect the interference cancellation operation. To trigger the training process of the C-DSL, interference from the PLC network has to be present. Similarly, the training of C-PLC only occurs if and only if the DSL interference is significant. Various techniques could be used to trigger the training process. For instance, the interference could be identified by applying a threshold to the CM signal, which if surpassed the training of the FDICs begins. A similar concept is proposed in [24], where once the CM interference is detected, the FDIC is trained and utilized to remove the DM interference. In this scheme, however, since the PLC-DC is connected to both the DSL and PLC transceivers, it is responsible for triggering the training process. Note that C-PLC does not require re-training unless a change in the physical channel occurs, while C-DLC has to be re-trained each time a new PLC user occupies the channel. This is due to the fact that there is only one DSL transceiver and its effect on each of the PLC transceiver does not change unless the location of the PLC transceiver changes or a change in the DSL channel occurs. On the other hand, as shown in Chapter 3 the effect of the PLC transceivers vary significantly from one power outlet to the other [14, 13] C-DSL Training C-DSL estimates the ratio of the DM interference to the CM interference by dividing the DM signal by the CM signal and then averaging the result. In order for training to occur, it is necessary to subtract the desired signal component from the DM received signal. This is possible only when the desired signal is known. Thus, training of the C-DSL occurs only during the sync frame of each DSL super-frame when y can be correctly estimated and subtracted from the received DM signal r d,dsl. As the C-DSL trains, interference will be present in the DSL super-frame sync 102

121 symbols and the synchronization of the DSL super-frame will degrade. However, Section will show that the C-DSL trains extremely quickly, with acceptable performance achieved after only a handful of symbols. As a result, it is assumed the C-DSL converges quickly enough for the sync symbols to benefit from interference cancellation so that DSL modem synchronization will not be adversely affected. After removing the DSL desired signal component, the received DM signal r d,dsl defined (5.5a) is reduced to r d,dsl = v d,dsl + η d,dsl. (5.9) The C-DSL tap-coefficient for sub-channel i during the k th symbol index, C DSL,k (i, i), is calculated by C DSL,k (i, i) = r d,dsl(i) r c,dsl (i) = v d,dsl(i) + η d,dsl (i) v c,dsl (i) + η c,dsl (i) (5.10) Note that since the C-DSL tap-coefficient matrix is a square diagonal matrix, C DSL (i, j) = 0 when j i. To minimize the effect of the background noise on the estimate of the ratio of the DM interference to the CM interference, we average C DSL,k (i, i) over K symbols. This average is utilized in the cancellation phase of the interference cancelling scheme. Thus, for sub-channel i, the averaged C-DSL tap-coefficient C DSL,K (i, i), defined by (5.11), is the estimate of the ratio of the DM interference to the CM interference. C DSL,K (i, i) = 1 K K C DSL,k (i, i) (5.11) k=1 After training the C-DSL, an estimate of the DM interference is extracted from the CM signal through the C-DSL (i.e., v d,dsl = C DSL,K r c,dsl ). This estimate of the DM interference is subtracted from the corresponding DM signal r d,dsl (i) to extract the DSL signal y(i). Finally, the DSL signal is passed through a frequency domain equalizer (FEQ) to calculate the estimate of the transmitted DSL symbol 103

122 x(i). Note that the FEQ coefficients are the inverse of the the elements of the DSL channel frequency response matrix. The effectiveness of this cancellation scheme will be demonstrated in Section using measured PLC-to-DSL cross-coupling channels C-PLC Training In Chapter 4, it was shown that the DSL interference effect on the PLC network is not as substantial as the effect of the PLC interference on the performance of the DSL network; however, as was shown in Fig. 4.9, interference due to US DSL transmission causes up to 20% loss in the achieved PLC data rates. Thus, an adaptive frequency domain interference canceller to mitigate the effects of the DSL-to-PLC interference is required to mitigate the effect of the DSL interference on the PLC system. Each time a PLC transceiver joins the PLC network, all PLC transceivers are forced into listening mode to estimate the interference caused by the DSL transceiver. Note that it was found in [13] that the cross-coupling channel between the PLC and DSL systems are stationary; thus, only the PLC transceiver joining the PLC network requires training. However, the PLC transceivers that previously trained their cancellers, if needed, can retrain their respective cancellers during the listening mode. Note that, unlike C-DSL, there is no restriction on when the C-PLC should be trained. The ratio of the DM to the CM cross-coupling channels (i.e., H f DSL d,dtx PRx to H f DSL c,dtx PRx in Fig. 5.4) is estimated by each PLC transceiver. Since all PLC transceivers are listening to the PLC channel, r d,plc is reduced to v d,plc and η d,plc, while r d,plc is equal to v c,plc and η c,plc. Note that v d,plc and V c,plc are equal to H f DSL d,dtx PRx y and H f DSL c,dtx PRx y respectively, sampled at integer multiples of f DSL. The training of C-PLC is similar to the training of C-DSL, which has been discussed in Section In essence, to estimate the ratio of the DM to CM DSL- 104

123 to-plc cross-coupling channels for the i th sub-carrier, while in listening mode, each PLC transceiver divides r d,plc (i) by r c,plc (i) to obtain the tap-coefficient C PLC (i, i, p) during the p th symbol interval, as shown in (5.12a). Note that since the background noise has an effect on C PLC, the tap-coefficient for each sub-carrier is averaged over P consecutive symbols to minimize the effect of the background noise on the estimate C PLC (i, i, p), as shown in (5.12b). C PLC (i, i, p) = r d,plc(i) r c,plc (i) = v d,plc(i) + η d,plc (i) v c,plc (i) + η c,plc (i) C PLC,P (i, i) = 1 P C DSL (i, i, p) P p=1 (5.12a) (5.12b) Performance Evaluation In this section, we evaluate the efficacy of the interference cancelling scheme presented in Section in mitigating the effect of the DM PLC interference on the DSL signal. Recall, our proposed interference cancelling scheme estimates the DM interference from the CM signal. The DSL and both the DM and CM cross-coupling channel frequency response matrices are required to evaluate the performance of the proposed cancelling scheme. The DSL channel frequency response matrix H DSL is obtained from the standard two-port model defined in [29]. However, the measured DM and CM PLC-to-DSL cross-coupling channels (i.e., matrices H d,ptx DRx and H c,ptx DRx respectively), which were presented in Chapter 3, are utilized to determine the PLC interference. The maximum transmission power for DSL symbols and PLC symbols are ε D = 60 dbm Hz [2] and ε P = 50 dbm Hz [9] respectively. Note that we assume that the background noises on the DM and CM channels are AWGN and not correlated. The PSDs of η d,dsl and η c,dsl are N 0,d and N 0,c, respectively. In the literature, N 0,d is usually assumed to be -140 dbm. As shown in Chapter 3, the measured background noise on Hz 105

124 the DSL line in the common mode setup, N 0,c, has a PSD of approximately -120 dbm Hz. The interference cancelling scheme is evaluated in three steps. In Section , the mean square error (MSE) of the proposed scheme when the C-DSL utilized is determined and compared with the MSE of the proposed scheme when an optimum Wiener filter is utilized. After determining the MSE, the SINR after utilizing the interference cancelling scheme is used to calculate the achieved improvement in available bit rates, which is then compared with the bit rates in absence of the cancellation scheme in Section Finally, in Section 5.5, the performance of the scheduling-based interference mitigation solution is compared with the performance of the spectral management mitigation solutions proposed in [27] and [28] Mean Square Error Analysis When determining MSE for both the Desk Modem Scenario and the Entry Point Scenario, the measured DM and CM PLC-to-DSL cross-coupling channels are assumed to be the noise free actual channels. Using these channels, our proposed cancelling scheme estimates the DM interference from the CM interference in presence of simulated background noise, represented by η d,dsl and η c,dsl in Fig As discussed in Section 5.3.2, the proposed interference cancelling scheme utilizes the C-DSL to estimate the ratio of the DM interference to the CM interference. From (5.3), the DM PLC interference for the i th sub-channel is defined as v d,dsl (i) = H d,ptx DRx (i, i)q(i), (5.13) while the CM interference for the i th sub-channel is calculated by v c,dsl (i) = H c,ptx DRx (i, i)q(i). (5.14) The coefficient of the canceller C(i, i) (5.11), which is an estimate of the ratio of the DM interference to the CM interference, is multiplied by the corresponding CM signal 106

125 to yield an estimate of the DM PLC interference v d,dsl (i). Thus, the estimate of the DM PLC interference for the i th sub-channel is calculated by v d,dsl (i) = C(i, i) (v c,dsl (i) + η c,dsl (i)). (5.15) Note that the MSE has the same units as the square of the estimated quantity and is therefore expressed in dbm Hz. After training the canceller, the estimation error for sub-channel i, which is the difference between v d,dsl (i) and its estimate v d,dsl, is given by e(i) = v d,dsl (i) v d,dsl =v d,dsl (i) C(i, i) (v c,dsl (i) + η c,dsl (i)). (5.16) Thus, the MSE for sub-channel i, φ(i) = E[e(i)e (i)], is defined by (5.17). φ(i) =E[v d,dsl (i)u d,dsl(i)] E[u d,dsl(i)c(i, i) (v c,dsl (i) + η c,dsl (i))] E[v d,dsl (i)(c(i, i) (v c,dsl (i) + η c,dsl (i))) ] + E[C(i, i)c (i, i)v c,dsl (i)v c,dsl(i)] (5.17) + E[C(i, i)c (i, i)η c,dsl (i)η c,dsl(i)] It is assumed that all the signals are complex wide sense stationary with zero mean, and the PLC symbol transmit power is ε P. Thus, E[v d,dsl (i)u d,dsl (i)] is equal to E[qq ] H d,ptx DRx (i, i) 2, where E[qq ] = ε P. Similarly, E[v c,dsl (i)u c,dsl (i)] is equal to ε P H c,ptx DRx (i, i) 2. Furthermore, the term E[v d,dsl (i)u c,dsl (i)] is simplified to E[qq ]H d,ptx DRx (i, i)h c,ptx DRx (i, i), where as the term E[u d,dsl (i)v c,dsl(i)] is simplified to E[qq ]H d,ptx DRx (i, i)h c,ptx DRx(i, i), respectively. Finally, the term E[η c,dsl (i)η c,dsl (i)] is equal to the PSD of the CM background noise N 0,c. Thus, 107

126 (5.17) simplifies to φ(i) =ε P H d,ptx DRx (i, i) 2 ε P C(i, i)h c,ptx DRx (i, i)h d,ptx DRx(i, i) ε P C (i, i)h c,ptx DRx(i, i)h d,ptx DRx (i, i) + ε P C(i, i) 2 H c,ptx DRx (i, i) 2 + C(i, i) 2 N 0,c. (5.18) The canceller tap-coefficient C(i, i) affects the value of the MSE. Thus, the weights C DSL,K (i, i) calculated by (5.11) are substituted in (5.18) to calculate the MSE achieved by the C-DSL, which is denoted φ F,K (i). Similarly, to calculate the MSE achieved by an optimum Wiener filter φ W (i), the weights C W (i, i) are substituted in (5.18). Note that C W (i, i) (5.19) is determined from the Wiener-Hopf equation [68]. C W (i, i) = ε PH d,ptx DRx (i, i)hc,ptx DRx (i, i) (5.19) ε P H c,ptx DRx (i, i) 2 + N 0,c The average MSE achieved by the C-DSL, φ F,K, is calculated for various values of K and compared to the MSE achieved by an optimum Wiener filter φ W, to study the effect of background noise on the performance of the C-DSL. The average MSE is calculated by averaging the achieved MSE for each sub-channel according to φ = 1 M M i=1 φ(i). For both measurement scenarios, φ F,K is determined for K=1, 5, and 10 symbols. Figs. 5.5 and 5.6 show the average achieved MSE versus the relative distance between the DSL and the PLC modems for the Desk Modem Scenario and the Entry Point Scenario respectively. Note that it was established in Chapter 3 that relative distance between the DSL and the PLC modems does not have a significant impact on the cross-coupling channel gains. However, for ease of comparison and to maintain consistency, the euclidean distance is used as a reference point. Note that, as mention at the beginning of Section 5.3.3, the maximum transmission power for DSL symbols and PLC symbols are ε D = 60 dbm and ε Hz P = 50 dbm Hz respectively. Additionally, η d,dsl has a PSD of N 0,d =-140 dbm/hz, while η c,dsl has 108

127 a PSD of N 0,c =-120 dbm/hz[13] φ F,1 φ F,5 128 φ F,10 φ W M SE in db m Hz Eucledean Distance in ft (not to scale) Figure 5.5: Desk Modem Scenario: MSE of the proposed scheme versus the relative distance between the DSL and PLC modems. From Figs. 5.5 and 5.6, we can first observe that φ W is at the same level of N 0,d, which highlights the potential of the proposed interference cancelling scheme to eliminate the DM PLC interference and offer performance comparable to a noise only environment. In addition, the number of training symbols required for the C-DSL to achieve an MSE comparable to φ W is quite small. This is due to the fact that the CM signal provides a very low noise estimate of the EMI corrupting the DM signal. Finally, the achieved MSE for the Entry Point Scenario, shown in Fig. 5.6, is lower than the achieved MSE for the Desk Modem Scenario, shown in Fig This is expected since in the Entry Point Scenario the in-line LPF reduces the DM PLC interference levels while the CM PLC interference levels remains unaffected, as shown in Fig Improvement in Bit Rate The proposed interference cancelling scheme, presented in Section 5.3.2, estimates the DM interference signal and subtract this estimate from the DM signal. The power of 109

128 136 φ F,1 137 φ F,5 φ F, φ W M SE in db m Hz Eucledean Distance in ft (not to scale) Figure 5.6: Entry Point Scenario: MSE of the proposed scheme versus the relative distance between the DSL and PLC modems. the remaining interference is equal to the MSE of the C-DSL. In this section, we use the MSE of the C-DSL and the optimum Wiener filter, discussed in Section , to calculate the SINR when the proposed interference cancelling scheme is utilized. The SINR achieved after utilizing the C-DSL and the optimum Wiener filter is used to calculate the available bit rates. These bit rates are compared with the bit rates in absence of the proposed interference cancelling scheme. The SNR in presence of only background noise for sub-channel i, γ 0, is the ratio of the signal power to the background noise power (5.20). If a co-located PLC network exists, the SINR for sub-channel i, γ(i), is the ratio of the desired DSL signal power to the DM PLC interference power and the background noise (5.21). γ(i) = γ 0 (i) = ε D H DSL (i, i) 2 N 0,d (5.20) ε D H DSL (i, i) 2 N 0,d + ε P H d,ptx DRx (i, i) 2 (5.21) When the interference cancelling scheme is utilized, the DM PLC interference power is reduced to the MSE of the utilized canceller. Let the SINR when the C-DSL and Wiener filter are utilized be γ F,K and γ W respectively. Thus, for frequency bin 110

129 i, γ F,K (i) and γ W (i) are calculated by (5.22a) and (5.22b) respectively. γ F,K (i) = ε D H DSL (i, i) 2 N 0,d + φ F,K (i) γ W (i) = ε D H DSL (i, i) 2 N 0,d + φ W (i) (5.22a) (5.22b) Recall, bit loading, where the number of bits allocated to each sub-channel is dependent on the sub-channel s SINR, is used in DSL [22]. The number of bits b(i) that can be loaded on frequency bin i, in absence of any interference cancelling scheme is calculated by (5.23), where the SNR gap Γ for DSL is 9.45 db and the maximum number of bits that can be allocated to a DSL frequency bin b max is 15 bits [27]. ( ( b(i) = min b max, log γ(i) ) ) Γ To calculate the number of bits for frequency bin i when C DSL,K utilized, γ(i) in (5.23) is replaced by γ F,K (i) and γ W (i) respectively. (5.23) and C W are Finally, in presence of AWGN noise only, the number of bits for frequency bin i is determine by replacing γ(i) in (5.23) with γ 0. After determining the number of bits that can be loaded on each frequency bin, the total bit rate R is then calculated by (5.24), where, for VDSL2, the normalizing factor λ is 0.79 and the sub-carrier spacing f is 8.6 khz [27]. R = λ f i b(i) (5.24) To study the effectiveness of our proposed interference cancelling scheme, we calculate the total bit rate when the C-DSL and Wiener filter are utilized (i.e., R F,K and R W respectively) and compare the results with the total bit rates in presence of only AWGN R 0 and in absence of any interference cancellation scheme R W/O, for both the Desk Modem Scenario and the Entry Point Scenario. 111

130 DSL Bit Rate in Mb/s R F,1 R F,5 R F,10 R W R 0 R W/ O Euclidean Distance in ft (not to scale) Figure 5.7: Desk Modem Scenario: Achieved improvement in bit rates versus the Euclidean distance between the DSL and PLC modems. Figs. 5.7 and 5.8 show the total bit rate, achieved by C DSL,K and C W, in absence of any interference cancellation, and by complete removal of the PLC interference versus the relative distance between the DSL and the PLC modem, for the Desk Modem Scenario and the Entry Point Scenario respectively. It is evident that the C-DSL successfully reduces the DM PLC interference levels. As K increases the C-DSL approaches the optimum performance by a Wiener filter. Note that despite the optimality of the Wiener filter, it does not achieve the maximum bit rates of R 0 because the weights of the Wiener filter are dependent on the cross-correlation between the DM and the CM interference, as shown in (5.19). 5.4 Pre-Distortion-Based Interference Mitigation Solution Utilizing the scheduling-based interference mitigation solution, proposed in Section 5.3, restricts channel access to one PLC user per DSL super-frame. Restricting channel access to one PLC user per DSL super-frame has two drawbacks. First drawback is underutilization of the PLC channel, which affects the overall PLC network throughput. This occurs in the event that a PLC user that is granted access to the channel 112

131 DSL Bit Rate in Mb/s R F,1 R F,5 R F,10 R W R 0 R W/ O Euclidean Distance in ft (not to scale) Figure 5.8: Entry Point Scenario: Achieved improvement in bit rates versus the Euclidean distance between the DSL and PLC modems. does not have enough information to transmit for the entire duration of the DSL super-frame. The second drawback is the latency experience by the PLC users due to the scheduling restriction of one PLC user per DSL super frame. This latency affects the QoS of interactive multimedia applications. The pre-distortion-based interference mitigation solution, proposed in this section, is suitable for interactive multimedia applications and is not significantly affected by the number of active PLC users. Since the PLC-to-DSL cross-coupling channel is reciprocal, the PLC transceivers can estimate the cross-coupling channel using the DSL super-frame training symbols when directed by the PLC-DC. The PLC transceivers then pre-multiply their symbols by the inverse of the DM cross-coupling channel. The PLC transceivers transmit two types of symbols: training symbols and data symbols. Training symbols are known to all PLC transceivers and to the DSL transceiver. Additionally, the same sequence of training symbols are always transmitted during the training of the FDIC used by the DSL system, i.e., C-DSL. This means that all interfering signals, during the training of the FDIC, arrive at the DSL receiver at a 113

132 constant gain offset relative to the desired DSL signal, regardless of source or subcarrier. Note that the constant gain offset is removed from the desired DSL signal, during the training of C-DSL. This allows multiple PLC users to access the PLC channel during a single a DSL super-frame, which was not the case for the schedulingbased interference mitigation solution. Note that the proposed modification to the PLC frame structure complies with the requirement of G.hn [9]. Recall, the DM and CM cross-coupling channels between a PLC transmitter and the DSL receiver, sampled at integer multiples of f DSL, are denoted by the channel frequency response matrices H f DSL d,ptx DRx and H f DSL c,ptx DRx. Similarly, the DM crosscoupling channel between the PLC transmitter and the DSL receiver when sampled at integer multiples of f PLC is defined by the channel frequency response matrix H f PLC d,ptx DRx. Additionally, recall that the received DSL DM signal r d,dsl is the summation of the desired DSL signal y = H DSL x, the DM PLC interference v d,dsl = H f DSL d,ptx DRx q, and the DM additive white Gaussian noise η d,dsl. Note that H DSL is the direct DSL channel, while x and q are the transmitted DSL and PLC symbols, respectively. Similarly, the received DSL CM signal r c,dsl is the summation of only the CM PLC interference v c,dsl = H f DSL c,ptx DRx q and the CM additive white Gaussian noise η c,dsl. Finally, recall that the received DM PLC signal r d,plc is the summation of the desired PLC signal z = H PLC q, the DM DSL interference v d,plc = H f PLC d,dtx PRx y, and the DM additive white Gaussian noise η d,plc. Also, the received CM PLC signal r c,plc is the summation of only the CM DSL interference v c,plc = H f PLC c,dtx PRx y and the CM additive white Gaussian noise η c,plc. The rest of this section is organized as follows. The technique with which the PLC symbol is pre-distorted is presented in Section In Section 5.4.2, the predistortion cancellation algorithm and the training of C-DSL are discussed. Finally, 114

133 in Section 5.4.3, the performance of pre-distortion solution is analyzed and compared with the performance of the scheduling-based solution PLC Symbol Pre-distortion Before transmission, the generated PLC symbol u is pre-multiplied by the inverse of the DM PLC-to-DSL cross-coupling channel H f PLC d,ptx DRx and the normalization factor a to form the transmitted PLC symbol q. Thus, the transmitted PLC symbol q, which causes interference to the DSL system as shown in Fig. 5.4, is defined by q = au H f DSL d,ptx DRx. (5.25) Recall, the main obstacle to allow more than one PLC user to access the PLC channel during a single DSL super-frame is the outlet-to-outlet variations of the PLCto-DSL cross-coupling channels. This obstacle is overcome in two steps. The first step is adding a training prefix (TP) at the beginning of each PLC frame, during which a training sequence that is known to all PLC transceivers and the DSL transceiver is transmitted. Thus, each PLC frame is composed of known training symbols followed by data symbols. The second step is pre-multiplying the PLC symbol by the inverse of the DM cross-coupling channel before transmission. This means that during the PLC frame TP transmission, no matter which PLC user is transmitting, the effect of the PLC interference on the DSL receiver is at a constant level across all sub-carriers. This constant level can be estimated and removed from the DSL DM signal. Additionally, during the PLC frame TP transmission, the C-DSL is trained to estimate the DM to CM cross-coupling channel ratio of each individual PLC user. This ratio is required to mitigate the effects of the PLC interference during the transmission of the PLC data symbols. Further detail on the training of the C-DSL is presented in Section The normalization factor a, on the other hand, is utilized to confine the PLC 115

134 signal power to the values stated in [9]. In addition, the normalization factor controls the interference level of the PLC symbol on the DSL system, which prevents the PLC interference from masking the DSL desired signal. Note that the effective direct PLC channel has to be estimated by each PLC transceiver. Recall, each time a PLC transceiver joins the PLC network, all PLC transceivers are forced into listening mode to estimate the interference caused by the DSL transceiver. In the event that an C-PLC is utilized to mitigated the effect of the DSL-to-PLC interference, the C-PLC is trained as discussed in Section After which, the PLC transceiver estimates the effective PLC channel (i.e., H PLC /H f PLC d,ptx DRx ). In absence of the pre-multiplication imposed by the proposed pre-distortion-based interference mitigation solution, a PLC receiver would have to estimate the direct PLC channel H PLC. However, since in our proposed cancellation scheme the PLC transmitter pre-multiples the generated PLC symbols u by the inverse of the DM cross-coupling channel H f PLC d,ptx DRx before transmission, the PLC receiver has to estimate H PLC /H f PLC d,ptx DRx and not only H PLC. Direct PLC channel estimation is beyond the scope of this thesis; however, the standard channel estimation approach utilized by G.hn to estimate the direct PLC channel could be utilized to estimate the product of the pre-distortion and the actual PLC channel Cancellation Algorithm In this section, we introduce the pre-distortion based interference cancellation scheme, where the training of the canceller C-DSL is discussed in Recall, the block diagram on which this scheme is based was introduced in Section 5.2. The C-DSL has to be re-trained each time a new PLC user occupies the channel because of the outlet-to-outlet variation in the PLC-to-DSL cross-coupling channels. Note that the training of the C-DSL occurs during the transmission of the PLC 116

135 frame TP. Thus, impact of the PLC interference on the DSL DM signal during the TP duration is reduced to a constant interference level, which can be removed from the DSL signal because the training sequence is known to the DSL transceiver. Once the C-DSL is trained, it is utilized to mitigate the effect of the PLC interference during the data symbol transmission portion of the PLC frame. As will be discussed in detail in Section , the TP added to each PLC frame along with the pre-distortion of the PLC symbols before transmission allow multiple PLC users to access the PLC channel during a DSL super-frame. Thus, the latency introduced by the solution proposed in Section 5.3 is eliminated, which allows the PLC network to support interactive multimedia traffic, with very low latency tolerance. Additionally, by allowing more than one PLC user to access the channel during the DSL superframe, the unused time-slots issue is overcome. Note that during the PLC frame TP transmission, the C-DSL is trained with no significant impact on the DSL system, as will be shown in Section C-DSL Training A predefined training sequence is transmitted by the PLC transceiver during the training portion of the PLC frame. The same sequence is used by all PLC transceivers, and it is known to the DSL transceiver. The PLC-DC initiates the training of the C-DSL each time a new PLC user access the channel. Recall, once the C-DSL is trained, the tap-coefficient of C-DSL remains valid until a new PLC user access the PLC channel. Each PLC frame is preceded with a training period, where the TP is transmitted. During this training period, the transmitting PLC transceiver transmits q (5.25), where the generated PLC symbol u is a pre-defined training sequence. The DSL canceller C-DSL tap-coefficient matrix C DSL is an M M square diagonal matrix. During training, the received CM DSL signal r c,dsl is utilized to estimate the ratio of the DM to CM PLC-to-DSL cross-coupling channels. Recall, the transmitted 117

136 PLC symbol q is an N 1 column vector, which is converted to an analog signal via an A/D before transmission; the received interference signals v d,dsl (5.3a) and v c,dsl (5.3b), on the other hand, are sampled at integer multiples of f DSL (i.e., v d,dsl and v c,dsl are an M 1 column vector). During the training period, the CM PLC interference for the i th sub-carrier v c,dsl (i) is defined in (5.26), where q f DSL (i) is the transmitted PLC symbols sampled at the DSL sampling frequency. During the training period, u is the pre-defined training sequence, which is known to all PLC transceivers and to the DSL transceiver. v c,dsl (i) =H f DSL c,dtx PRx (i)q f DSL (i) ( =H f DSL c,dtx PRx (i) au(i) H f PLC d,ptx DRx (i) ) fdsl (5.26) To extract an estimate of the DM to the CM PLC cross-coupling channel ratio, the C-DSL is trained by dividing the known training sequence by the received CM signal. Thus, for the l th symbol interval, the C-DSL tap-coefficient for the i th sub-carrier C DSL (i, i, l) is calculated by (5.27), where v c,dsl (i) is defined in (5.26). Note that the PLC-DC is connected to the C-DSL to communicate the value of the normalizing factor a as shown in Fig C DSL (i, i, l) = au(i) r c,dsl (i) = au(i) v c,dsl (i) + η c,dsl (i) (5.27) To minimize the effect of the AWGN noise on the estimate of C DSL, the value of C DSL (i, i, l) is averaged over L successive training symbols. Thus, for the i th subcarrier, the averaged tap-coefficient C DSL,L (i, i), defined by (5.28), is the estimate of the the DM interference to the CM interference. C DSL,L (i, i) = 1 L C DSL (i, i, l) (5.28) L l=1 Note that while training C-DSL, the pre-defined training sequence is subtracted from r d,dsl. During the training of C-DSL, for the i th sub-carrier, the DM PLC inter- 118

137 ference v d,dsl (i) is approximately equal to au, as shown in (5.29). Thus, subtracting au(i) from r d,dsl (i) mitigates the PLC DM interference during the training phase of C-DSL, as will be confirmed by the analysis presented in Section v d,dsl (i) =H f DSL d,dtx PRx (i)q f DSL (i) H f DSL d,dtx PRx (i) au(i) (5.29) ( ) H f PLC d,ptx DRx (i) fdsl The reason than v d,dsl (i) is not exactly equal au(i) is the difference in sampling frequencies utilized by the DSL and PLC systems. Since each system samples the cross-coupling channel at different frequencies, the resulting samples are not equal ( ) (i.e.,h f DSL d,dtx PRx (i) is only equal H f PLC d,ptx DRx (i) fdsl when the samples are taken at the same frequency). However, as will be shown in Section 5.4.3, this difference does not impact the performance of the DSL system, either during the training phase or during the transmission phase of the C-DSL canceller Performance Analysis To evaluate the performance of the pre-distortion-based interference mitigation solution, the MSEs during the the training phase and transmission phases of C-DSL are determined. During the training phase of C-DSL, the known training sequence is subtracted from the received DSL signal. Thus, the MSE of the pre-distortion-based solution during the training phase of C-DSL does not depend on its tap-coefficients. However, during transmission phase, the tap-coefficients of C-DSL affects the MSE of the solution because the error during transmission phase is not simply the difference between the known training sequence and the interference due to the transmission of the TP. Rather, the error is the difference between the output of the C-DSL canceller and the DM PLC interference due to the transmission of the data symbols portion of the PLC frame, as shown in Fig

138 In this section, we evaluate the performance of C-DSL during both the training phase and transmission phase in Sections and respectively. To evaluate the performance of C-DSL the mean square error (MSE) is analyzed for both training and transmission phase. After which, the available bit rates based on the SINR per sub-carrier is calculated before and after utilizing the proposed canceller for both the training and transmission phase. It was shown in Fig. 5.2, the PLC interference s couples on each of the DSL twisted-pair differently, according to the imbalance factor of each of the twistedpairs. Let us assume s = a.u, where a is the normalization factor, and u is the generated PLC symbol. For the twisted-pair imbalance factors k and j, the DM to a.u(k j) CM interference ratio is. Thus, variations in a will appear in both the 0.5 a.u(k + j) DM and the CM PLC interference signals, and since C-DSL estimates a ratio, these variations should cancel out. Therefore, if needed to maintain the performance of the PLC system, during the transmission phase each PLC transceiver can choose a normalization factor that maintains the power spectral density of the received PLC signal. However, during the training phase, all PLC transceiver have to utilize the same known value for the normalization factor a. By ensuring that, no matter which PLC user is active, the PLC interference level at the DSL receiver is constant during the training phase of C-DSL, the PLC interference can be easily removed while training C-DSL. The DSL direct channel frequency response matrix, along with the DM PLC-to- DSL cross-coupling channel response matrices sampled at both f DSL and f PLC, and the CM PLC-to-DSL cross-coupling channel response matrices sampled at f DSL are required in the evaluation process. Recall, the DSL direct channel is obtained from the standard two port model discussed in [29]; while, the cross-coupling channels are obtained from the measurements presented in Chapter 3. Note that we utilize the 120

139 measurements for the Entry Point Scenario in the following analysis. Recall, DSL and PLC utilize frequency separations of f DSL = 8.6 khz and f PLC =24.4 khz respectively. However, for simplicity, we assume that f PLC is three times f DSL (i.e., f DSL = 8.6 khz and f PLC = 25.8 khz). Thus, the DSL and PLC system utilize M and N orthogonal sub-carriers respectively, where N = M/3. The maximum transmission power for DSL symbols and PLC symbols are ε D = 60 dbm Hz [2] and ε P = a dbm Hz respectively. Note that we assume that both η d,dsl and η d,plc have a PSD of N 0,d =-140 dbm/hz, while η c,dsl and η c,plc have a PSD of N 0,c =-120 dbm/hz [13]. Based on each sub-carrier s SINR, a number of bits are allocated to each subcarrier [22]. The number of bits b(i) that can be transmitted over the i th sub-carrier is calculated by (5.23) Training Phase During the training phase, the known training sequence au is subtracted from the received DSL signal r d,dsl = v d,dsl + η d,dsl. Therefore, the error for the i th subcarrier, e Tr (i) = v d,dsl (i) au(i). Note that if DSL and PLC system utilize the same frequency separation with aligned sub-carrier locations, v d,dsl (i) will be equal to au(i). However, since the PLC-to-DSL cross-coupling channel is sampled at different frequencies by the PLC and DSL systems, v d,dsl (i) au(i), which results in errors. Thus, the MSE during the training phase of the C-DSL for sub-carrier i, φ Tr (i) = E[e Tr (i)e Tr (i)], is defined by (5.30). φ Tr (i) = v d,dsl (i) 2 + a 2 u 2 (i) au(i)(v d,dsl (i) + v d,dsl(i)) (5.30) The SNR in presence of only background noise is the maximum bound achieved by complete removal of the interference. Thus, for the i th sub-carrier, the SNR bound 121

140 γ Bound (i) is the ratio of the signal power to the background noise power (5.20). If a colocated PLC network exists, the SINR for the i th sub-carrier, γ W/O (i), is the ratio of the desired DSL signal power to the DM PLC interference power and the background noise (5.21). Utilizing the MSE calculated by (5.30), we calculate the SINR after interference mitigation during the training phase. This is performed by substituting the interference power with the calculated MSE. Thus, for the i th sub-carrier, the SINR after interference cancellation is calculated by (5.22). To calculate the available bit rates for the DSL system during the training phase of the C-DSL, we calculate the number of bits that can be loaded to each of the DSL sub-carriers, based on each sub-carrier s SINR, from (5.23). The number of bits for the SNR bound, SINR without interference cancellation, and the SINR during the training phase are calculated from the SNR (or SINR) for each case in (5.23). After determining the number of bits that can be loaded on each sub-carrier, for each case, the total bit rate R is then calculated by (5.24). where, for VDSL2, the factor λ is 0.79 and the sub-carrier spacing f DSL is 8.6 khz [27]. Table 5.2 shows the available bit rates during the training phase of C-DSL. Four cases are considered. The first and the second are the total bit rates in presence of only background noise R Bound,Tr and without interference cancellation R W/O,Tr. The other two cases are when the interference cancellation scheme is utilized; however, we study the total bit rates if both DSL and PLC utilize the same frequency separation R M=N,Tr (i.e., f PLC = f DSL ) and the total bit rates when the frequency separation between the PLC sub-carriers is 3 times the frequency separation utilized by the DSL system R M=N/3,Tr (i.e., f PLC = 3 f DSL ). From Table 5.2, it is evident that during the training phase, subtracting au from v d,dsl achieves optimal bit rates if both PLC and DSL utilized the same frequency separation, and near-optimal bit rates when the frequency separation utilized by PLC is 3 times the frequency separation utilized by 122

141 Table 5.2: Training phase available bit rates. Case Bits in Mb/S R Bound,Tr R W/O,Tr R M=N,Tr R M=N/3,Tr DSL Transmission Phase During transmission phase, the canceller tap-coefficient C DSL (i, i) affects the value of the MSE. Thus, the number of symbols utilized in training C-DSL will affect the available bit rates. The error during transmission phase is not simply the difference between the known training sequence and the DM PLC interference estimate v d,dsl, but rather the error is the difference between the output of the C-DSL canceller v d,dsl and v d,dsl, as shown in Fig The transmission phase error for the i th sub-carrier, e Tx (i) = v d,dsl (i) v d,dsl (i). Thus, The MSE during the transmission phase for sub-carrier i, φ Tx (i) = E[e Tx (i)e Tx (i)], is defined by (5.31). φ Tx (i) = v d,dsl (i) 2 + v d,dsl (i) 2 v d,dsl(i)v d,dsl (i) v d,dsl (i)v d,dsl(i) (5.31) Similar to the approach used to evaluate the C-DSL performance during training, we utilize the MSE to calculate the SINR. The SINR is then utilized to calculate the available bit rates for the various cases presented in Table 5.2. However, since the the number of symbols used in training affects the MSE, and consequently the achieved SINR and available bit rates, we calculate the available bit rates for various numbers of training symbols. Recall, C DSL (i, i, l) is averaged over l symbols to mitigate the 123

142 Table 5.3: Transmission phase available bit rates. l (symbols) R M=N,Tx (Mb/s) R M=N/3,Tx (Mb/s) effect of the AWGN noise on the estimate, as shown by (5.28). Table 5.3 shows the available bit rates for two cases, where both the DSL and PLC systems utilize the same frequency separation and when the frequency separation between the PLC sub-carriers is 3 times the frequency separation utilized by the DSL system during the transmission phase of C-DSL (i.e., R M=N,Tx and R M=N/3,Tx respectively), for DSL cable run length of 2000 ft and for various values of l. The total available bit rates for the bound case R Bound,Tx and in absence of interference cancellation R W/O,Tx are and 60.3 M b/s respectively. From Table 5.3, it is clear that the canceller C-DSL successfully mitigates the effect of the PLC interference on the DSL system, which is reflected in the available bit rates achieved by the canceller. Fig. 5.9 shows the total bit rates achieved by the C-DSL, for N=M and N=M/3, and by the same C-DSL if it was trained according to the scheme discussed in Section 5.3 versus the total bit rates in presence of only background noise (denoted by Bound) and without interference cancellation (indicated by W/O), for various DSL cable run lengths. Note that the results in Fig. 5.9 are the achieved DSL DSL bit rates for band plan 998E30 [2], after training each of the C-DSLs for 5 symbols. It is evident that C-DSL achieve slightly better performance that the canceller presented in Section 5.3, both for N=M and N=M/3. It was shown in Section that the scheduling based interference mitigation solution is suitable for multimedia applications that require the latency to be in the 124

143 Bound C DSL in Section 5.3 W/O C DSL N=M C DSL N=M/3 DSL Bit Rate in Mb/s DSL loop length in ft Figure 5.9: Achieved total DSL bit rates for DS transmission, for various DSL cable run length. range of 100 to 300 ms, such as IPTV and high definition video streaming. However, interactive multimedia applications, such as VoIP and Internet video conferencing, have very low latency tolerance that lies is the range of 10 to 30 ms [67]. The purpose of the pre-distortion-based interference mitigation solution is to mitigate the PLCto-DSL interference without introducing significant latency to the PLC system. For a saturated PLC system, i.e., all users have packets to transmit, the latency is calculated by T LAT = (n 1)τ, (5.32) where n is the number of active PLC users and τ is the duration the PLC channel is restricted to a single user. In Section 5.3.2, once a PLC user takes over the channel, other PLC users cannot utilize it for the duration of the DSL super-frame T SF. Thus, for the scheduling-based solution, and in presence of a VDSL2 system, T τ = T SF = ms is substituted in (5.32) to calculate the latency of the scheduling-based algorithm T LAT,Scheduling. On the other hand, if C-DSL is trained according to the algorithm discussed in Section 5.4.2, which does not impose any additional restriction 125

144 on the PLC channel access, T τ = T PLC =100 µs, where T PLC is the duration of the PLC time-slot duration. Thus, to calculate the latency due to the pre-distortion algorithm T LAT,Pre distortion, T τ = 100 ms is substituted in (5.32). Table 5.4: Latency: Scheduling-based versus pre-distortion-based solutions. D (active users) T LAT,Scheduling (ms) T LAT,Pre distortion (ms) Table 5.4 shows the latency experience by the PLC network due to both the scheduling-based and the pre-distortion-based interference mitigation solutions. For each algorithm, the latency is calculated by (5.32) based on the number of active PLC users. It is evident that the latency introduced by the pre-distortion based solution is significantly lower than the latency caused by the scheduling based solution. Additionally, only the pre-distortion based solution is suitable for interactive multimedia applications with very low latency tolerance. 5.5 Comparison with Spectral Management Solutions In this section, we compare the achieved bit rates of the scheduling-based interference mitigation solution presented in Section 5.3 with the achieved bit rates of the spectral management (SM) solutions proposed in [27] and [28], over the VDSL2 spectrum. Note that both scheduling-based and pre-distortion-based solutions achieve similar bit rates; however, in the pre-distortion-based solution, the simplicity of the scheme is sacrificed to reduce the latency experienced by the PLC users. In [27], the PLC transmit power for all sub-carriers within the VDSL2 spectrum is reduced; however, only the transmit power for the PLC sub-carriers that overlap with 126

145 the DSL downstream (DS) frequencies is reduced in [28]. Henceforth, the solutions proposed in [27] and [28], will be referred to as SMFR (SM flat reduction) and SMSR (SM selective reduction) respectively. The factor used to compare the performance of the three PLC-to-DSL interference mitigation solutions is the available bit rates. We utilize band plan 998E30 [2] and study the bit rates achieved by each solution for three cases, namely: DSL downstream (DSL-DS), DSL upstream (DSL-US), and PLC. In each case, we calculate the available bit rate based on the SINR as discussed in Section In Figs to 5.15, the available bit rate in presence of only AWGN noise is indicated by Bound, while the bit rate without any interference cancellation is denoted by W/O. The achieved bit rates by the C-DSL when trained for 5 symbols C DSL,5 are indicated by F,5. If the performance of both SMFR and SMSR is identical (i.e., in the case of DSL-DS), SM is used to indicated the achieved bit rates by both SMFR and SMSR. Note that the dbm/hz, -75 dbm/hz is utilized to donate that the achieved bit rates when the PLC transmit power ε P is set to -50, -60, and -70 dbm/hz, respectively. DSL Bit Rate in Mb/s Bound 50 dbm/hz 75 dbm/hz 50 dbm/hz 60 dbm/hz 75 dbm/hz DSL loop length in ft Figure 5.10: Desk Modem Scenario: Achieved bit rates for DSL-DS. 127

146 Fig shows the DSL-DS bit rates achieved by the PLC-to-DSL interference mitigation solutions versus the DSL loop length for the Desk Modem Scenario. The percentage improvement of a bit rate R relative to no interference cancellation is calculated according to ( ) R R W/O /RW/O. For a short DSL loop length (i.e., DSL loop length less than 500 ft in Fig. 5.10), the percentage of improvement achieved by the proposed scheduling-based interference mitigation solution ranges from 50% to 80%, while for long DSL loops (i.e., DSL loop length greater than 500 ft in Fig. 5.10), the percentage of improvement achieved by our proposed scheme ranges from 100% to 300%. The variation in the improvement percentage is because for short DSL loops the limiting factor for the number of bits per frequency bin is b max, while for longer loops, the PLC-to-DSL interference plays a role in reducing the SINR, which consequently affects the total bit rates. On the other hand, the improvement achieved by both SMFR and SMSR by reducing ε P to -60 dbm/hz is significantly lower than the improvement achieved by our scheme. While both SMFR and SMSR achieve comparable results to our scheme when ε P to -75 dbm/hz, this reduction in PLC transmit power significantly hinders the achievable PLC bit rates, as will be shown in Figs and Similarly, for the Entry Point Scenario, the scheduling-based interference mitigation solution achieves higher SINR than both SMFR and SMSR, which is reflected in the available bit rates shown in Fig. 5.11, especially for ε P less than -75 dbm/hz. However, for ε P equal to -75 dbm/hz, both SMFR and SMSR achieve bit rate improvement that is identical to the scheduling-based interference mitigation solution. Note that the percentage of bit rate improvement in the Entry Point Scenario is lower than that of the Desk Modem Scenario, because the PLC-to-DSL interference levels are lower due to the in-line LPF. To study the effect of reducing ε P on the PLC bit rates, we simulated a PLC 128

147 DSL Bit Rate in Mb/s Bound 50 dbm/hz 75 dbm/hz 50 dbm/hz 60 dbm/hz 75 dbm/hz DSL loop length in ft Figure 5.11: Entry Point Scenario: Achieved bit rates for DSL-DS. channel based on the model presented in [66]. A medium PLC channel [69] was generated and the simulation included both AWGN on the PLC channel at N o = -140 dbm/hz and the interference from the DSL signal that couples across to the PLC receiver through H D,d channel shown in Fig For the PLC simulation, we assumed Γ = 9.45 db, λ = 0.75, f = 24.4 khz, and b max = 12 bits [27] PLC Bit Rate in Mb/s dbm/hz 60 dbm/hz 60 dbm/hz 75 dbm/hz 75 dbm/hz DSL loop length in ft Figure 5.12: Desk Modem Scenario: Achieved bit rates for PLC. In Figs and 5.13, FDIC represents the PLC bit rates available when the 129

148 PLC Bit Rate in Mb/s dbm/hz dbm/hz 60 dbm/hz dbm/hz dbm/hz DSL loop length in ft Figure 5.13: Entry Point Scenario: Achieved bit rates for PLC. scheduling-based interference mitigation solution is utilized, which are the available PLC bit rates without reducing the PLC transmit power. The achieved DSL-DS bit rates by both SMFR and SMSR was comparable to the DSL-DS bit rates achieved by the scheduling-based interference mitigation solution when ε P is set to -75 dbm/hz. However, as shown in Fig. 5.12, reducing ε P to -75 dbm/hz results in a 65% to 85% reduction in the PLC bit rates for SMFR and a 45% to 55% reduction in the PLC bit rates for SMSR, for the Desk Modem Scenario. Our proposed solution, on the other hand, does not hinder the PLC bit rates because our scheme relies on estimating the DM PLC interference and subtracting it from the DSL signal rather than reducing the PLC transmit power. Similarly, for the Entry Point Scenario, shown in Fig. 5.13, reducing ε P to -75 dbm/hz results in a reduction in the PLC bit rates for both SMFR and SMSR. However, the reduction is not as large as for the Desk Modem Scenario, because the interference from the DSL signal via the H D,d channel is reduced by the in-line LPF. Figs and 5.15 show the DSL-US bit rates achieved by the PLC-to-DSL interference mitigation solutions versus the DSL loop length for both the Desk Modem 130

149 Bound 50 dbm/hz 50 dbm/hz DSL Bit Rate in Mb/s DSL loop length in ft Figure 5.14: Desk Modem Scenario: Achieved bit rates for DSL-US. Scenario and The Entry Point Scenario respectively. SMSR achieves higher PLC bit rates than SMFR, because SMSR only reduces the PLC transmit power for subcarriers that overlap with the DS frequencies of the VDSL2 spectrum. However, not reducing the transmit power of sub-carriers that overlap with the US frequencies of the VDSL2 spectrum negatively affects the DSL-US bit rates Bound 50 dbm/hz 50 dbm/hz DSL Bit Rate in Mb/s DSL loop length in ft Figure 5.15: Entry Point Scenario: Achieved bit rates for DSL-US. For the Desk Modem Scenario, shown in Fig. 5.14, the DSL-US bit rates when 131

150 SMSR is utilized are significantly lower than DSL-US bit rates achieved by our proposed solution, especially for short DSL loops. For short DSL loops, b max is usually the limiting factor, if the PLC-to-DSL interference is removed. However, in presence of PLC interference, which is the case when SMSR is utilized, the numbers of bits loaded to each sub-carrier is dependent on the SINR. For the Entry point Scenario, shown in Fig. 5.15, where the PLC interference is reduced by the in-line LPF, the DSL-US bit rates achieved by SMSR is lower than the rates achieved by our proposed solution. However, the gap between the achieved bit rates between our solution and SMSR is not as pronounced as it was for the Desk Modem Scenario. 132

151 Chapter 6 CONCLUSION Recent advances in PLC have made it popular for in-home networking. This makes PLC an increasingly relevant source of interference for DSL networks within the home environment. Current solutions proved to be favouring the performance of the DSL network over the PLC achievable bit rates. In fact, with the increase of the usable DSL bandwidth, these current solutions will render the PLC network inoperable. The objective of this thesis is to provide solutions that mitigate the interference between co-located DSL and in-home PLC networks, without degrading the performance of the PLC networks. The main hypothesis of this is that the common mode signal, which can be determined at the receiver and is usually ignored, contains information about the external EMI. Via an adaptive frequency domain canceller, the common signal can be utilized to estimate differential mode interference EMI. To the best of my knowledge, neither field measurements of PLC-to-DSL crosscoupling channels have been performed nor a model that characterizes the interference environment between DSL and in-home PLC networks exists. Thus, field measurements that characterize the interference environment in a residential setting are required. As discussed in Chapter 3, a measurement campaign that has been designed to characterize the PLC-to-DSL cross-coupling channels was conducted in two residential test-sites. The findings of the measurement campaign are summarized in Section 6.1. Both the DM and CM PLC-to-DSL cross-coupling channels are studied within two residential houses as test sites. For each of the two test sites, the DM and CM PLC-to-DSL cross-coupling channels are measured for various rooms. Mitigating the PLC-to-DSL interference via utilizing adaptive filters have never 133

152 been performed in the literature. Only time domain adaptive cancellers have been proposed to mitigate the near end crosstalk in ADSL networks. However, in spite of the relative narrow-bandwidth utilized by the ADSL technology, these time domain filters required a long training time. In addition, due to the frequency selective nature of the PLC interference, these time domain adaptive filters cannot mitigate the DM PLC interference. Two PLC-to-DSL interference mitigation solutions were presented in Chapter 5. Both solutions utilize frequency domain adaptive cancellers to combat the frequency selectivity of the PLC-to-DSL cross-coupling channel. Via these cancellers, the proposed solutions estimate the differential mode interference from the usually ignored common mode signal, on a tone by tone basis. The functionality of the proposed solutions is summarized in Section 6.2. Finally, in Section 6.3, recommendations for future research is proposed. 6.1 Measurement Campaign Findings The main findings of the measurement campaign are: The interference from an in-home PLC network to a co-located DSL modem is significant. Effect of the PLC interference on the DSL bit rates varies according to the distance between the DSL modem and the CO. For short DSL cable run length, interference from DSL negatively impacts the performance of the PLC network. Utilizing an LPF to prevent the travel of the DSL signal over the house internal telephone wires reduces but does not eliminate the degradation of the DSL system performance due to the PLC interference. 134

153 Spatial separation between the DSL modem and the PLC modems has no significant impact on the interference. The in-home cross-coupling channels, both DM and CM, are frequency selective. The ratio of the DM to CM PLC-to-DSL cross-coupling channels is not a smooth function of frequency. Both cross-coupling channels significantly vary from one power outlet to the other. In a non-electrically active test site, the cross-coupling channels are stationary. 6.2 Interference Mitigation Solutions Both interference solutions utilize FDICs. The FDICs are composed of parallel singletap filters, where the number of taps equals the number of sub-channels. Each tap of the FDIC is an estimate of the ratio of the DM to the CM interference for a given subchannel. Since each sub-channel is processed independently, the frequency selectivity of the cross-coupling channels is overcome. Additionally, since the FDIC estimates a ratio, variations in the PLC channel due to its non-stationarity does not affect the performance of the FDIC. The first interference mitigation solution is a scheduling-based solution, where access to the PLC channel is restricted to one PLC user per DSL super-frame. This restriction overcomes the variation in cross-coupling channels due to changes in the location of the PLC transceiver. For less than 6 active users in the PLC network, the added delay due to this restriction does not deteriorate the QoS for non-interactive multimedia applications. However, as number of active users in the PLC network increases, this restriction severely degrades the QoS for the multimedia applications. 135

154 The second interference mitigation solution is a pre-distortion based solution, where the PLC symbols are multiplied by the inverse of the DM PLC-to-DSL crosscoupling channel and a training period is added to the beginning of each PLC frame. This allows the FDIC to train during the PLC frame training period instead of restricting channel access to one PLC user per DSL super-frame. The pre-distortion based solution does not add any latency to PLC frame transmissions, and it meets the necessary QoS requirements for interactive multimedia applications. However, it is more complex to implement than the scheduling-based solution because of the required signal processing. 6.3 Recommendation for Future Research Future work motivated by the findings of this thesis is discussed in this section. For consistency, the future work is categorized according to the two focus points of the thesis. First, field measurements for cross-coupling channels over the G.fast spectrum are discussed in Section The proposed approach to solve the interference source in presence of interference from multiple sources and developing a replacement system for the current DSL transmission scheme is presented in Section Interference Channel Characterization Two measurement campaigns are required to characterize the DM and CM PLCto-DSL cross-coupling channels and the DSL DM and CM direct channels over the G.fast spectrum are discussed in Sections and , respectively. Note that G.fast has a usable bandwidth that goes up to 212 MHz PLC-to-DSL Cross-Coupling Channels A measurement campaign that studies the PLC-to-DSL coupling environment over the VDSL2 spectrum has been performed in [13, 14]. While this study provides insight 136

155 into the relation between the CM interference and the DM interference, results from only two test-sites is not sufficient to extract a model for this relation. In addition, only interference over the VDSL2 spectrum, i.e., up to 30 MHz, was considered in the measurement campaign. The goal of this future campaign is to define a model for the relationship between the CM and DM PLC-to-DSL coupling channels that considers frequencies up to 212 MHz. To achieve this goal, more measurements in residential houses are need. Note that these measurements will be performed using the setup discussed in Chapter 3; however, modification to the balun and the PLC coupler is necessary to accommodate the wider spectrum of G.fast. Various factors will affect the number of houses in which the measurements will be conducted. First is the variance of the measurements. If it is determined that the C2DTF does not significantly vary from house to house, fewer houses will be utilized to extract a model for the C2DTF of the PLC-to-DSL cross-coupling channels. Another factor that will influence the number of houses used in this measurement campaign is the availability of the houses. Collaboration with a Canadian telecommunications company is underway to determine the possibility of locating various houses for measurements. Once these houses are identified, measurements to study the PLC-to-DSL coupling channels will be performed DM and CM DSL Direct Channels To characterize the interference in DSL networks over the 212 MHz spectrum, the DSL direct channels are needed. The DM DSL channels over the G.fast spectrum do not conform to the standard two-port model [8]. Despite various research efforts and proposed models, such as the model proposed in [70], there is much to learn about the behaviour of the DSL channel in the 30 to 212 MHz frequencies [8]. In addition, the current DSL channel models do not model the CM signal. This measurement campaign will be performed in two phases. The first phase will 137

156 constitute measurements in the lab environment, since these high frequencies are only suitable for short distances. The setup discussed in Chapter 3 is required to perform these measurements with the exception of utilizing a balun with a 212 MHz range. Once lab measurements have been conducted, field measurements will be performed to substantiate the lab measurements. It is suggested that these filed measurements should be performed in collaboration with a local exchange carrier because access to both ends of the twisted-pairs is required. Note that since utilizing a VNA to perform these field measurements is impractical, software radios will be utilized Interference Cancellation Currently, for VDSL2 and similar DSL technologies, NEXT is eliminated via FDD, while FEXT is mitigated via vectoring. The FDIC has the potential to eliminate intrinsic interference, without the complexity of vectoring and the restriction of FDD. In addition, given that TDD is proposed as the duplexing scheme, for G.fast DSL technology, NEXT will return as the dominant crosstalk. However, the performance of the FDIC in mitigating multiple sources of interference has to be studied in further detail. Preliminary results indicate that the FDIC has the ability to reduce interference from multiple sources; however, the achieved improvement in SINR in presence of multiple sources is not as high as the achieved improvement in SINR in presences of a single source of interference. This is because the FDIC estimates the equivalent interference levels based on the average (or maximum) transmission power. Thus, once actual transmission occurs in presence of multiple sources, the ratio of the summed differential mode interference to the summed common mode interference does not remain constant, while this ratio remains constant for a single interference source. A potential solution to this problem is to utilize FDICs at the DSLAM (an FDIC for each of the twisted-pair). At the DSLAM, all transceivers are co-located, and thus, 138

157 information about the DSL signals on each of the twisted-pairs can be shared among the transceivers. This information can be utilized by the FDIC in the calculation of the C2DTF. For instance, the actual transmitted symbol for each of the disturbers, rather than the average (or maximum) transmission power, can be utilized by the FDIC to calculate the C2DTF. This solution has to be examined in further details to ensure its functionality and feasibility. In addition, the effect of uncontrolled lines on this solution has to be studied as well. 139

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