Low Distortion, Wide Bandwidth Voltage Feedback Clamp Amps AD8036/AD8037

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1 a FEATURES Superb Clamping Characteristics 3 mv Clamp Error 1.5 ns Overdrive Recovery Minimized Nonlinear Clamping Region 24 MHz Clamp Input Bandwidth 3.9 V Clamp Input Range Wide Bandwidth AD836 AD837 Small Signal 24 MHz 27 MHz Large Signal (4 V p-p) 195 MHz 19 MHz Good DC Characteristics 2 mv Offset 1 V/ C Drift Ultralow Distortion, Low Noise 72 dbc 2 MHz 4.5 nv/ Hz Input Voltage Noise High Speed Slew Rate 15 V/ s Settling 1 ns to.1%, 16 ns to.1% 3 V to 5 V Supply Operation APPLICATIONS ADC Buffer IF/RF Signal Processing High Quality Imaging Broadcast Video Systems Video Amplifier Full Wave Rectifier PRODUCT DESCRIPTION The AD836 and AD837 are wide bandwidth, low distortion clamping amplifiers. The AD836 is unity gain stable. The AD837 is stable at a gain of two or greater. These devices allow the designer to specify a high (V CH ) and low (V CL ) output clamp voltage. The output signal will clamp at these specified levels. Utilizing a unique patent pending CLAMPIN input clamp architecture, the AD836 and AD837 offer a 1 improvement in clamp performance compared to traditional output clamping devices. In particular, clamp error is typically 3 mv or less and distortion in the clamp region is minimized. This product can be used as a classical op amp or a clamp amplifier where a high and low output voltage are specified. The AD836 and AD837, which utilize a voltage feedback architecture, meet the requirements of many applications which previously depended on current feedback amplifiers. The AD836 and AD837 exhibit an exceptionally fast and accurate pulse response (16 ns to.1%), extremely wide small-signal and CLAMPIN is a trademark of Analog Devices, Inc. Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Low Distortion, Wide Bandwidth Voltage Feedback Clamp Amps AD836/AD837 large-signal bandwidths and ultralow distortion. The AD836 achieves 66 dbc at 2 MHz, and 24 MHz small-signal and 195 MHz large-signal bandwidths. The AD836 and AD837 s recover from 2 clamp overdrive within 1.5 ns. These characteristics position the AD836/AD837 ideally for driving as well as buffering flash and high resolution ADCs. In addition to traditional output clamp amplifier applications, the input clamp architecture supports the clamp levels as additional inputs to the amplifier. As such, in addition to static dc clamp levels, signals with speeds up to 24 MHz can be applied to the clamp pins. The clamp values can also be set to any value within the output voltage range provided that is greater that. Due to these clamp characteristics, the AD836 and AD837 can be used in nontraditional applications such as a full-wave rectifier, a pulse generator, or an amplitude modulator. These novel applications are only examples of some of the diverse applications which can be designed with input clamps. The AD836 is offered in chips, industrial ( 4 C to +85 C) and military ( 55 C to +125 C) package temperature ranges and the AD837 in industrial. Industrial versions are available in plastic DIP and SOIC; MIL versions are packaged in cerdip. OUTPUT VOLTAGE Volts FUNCTIONAL BLOCK DIAGRAM 8-Lead Plastic DIP (N), Cerdip (Q), and SO Packages AD836 = 1V = 2V = 3V NC INPUT +INPUT AD836/ AD837 (Top View) NC = NO CONNECT = 3V = 2V = 1V INPUT VOLTAGE Volts Figure 1. Clamp DC Accuracy vs. Input Voltage OUTPUT One Technology Way, P.O. Box 916, Norwood, MA , U.S.A. Tel: 781/ World Wide Web Site: Fax: 781/ Analog Devices, Inc., 2

2 AD836/AD837 SPECIFICATIONS ( V S = 5 V; R LOAD = 1 ; A V = +1 (AD836); A V = +2 (AD837),, open, unless ELECTRICAL CHARACTERISTICS otherwise noted) AD836A AD837A Parameter Conditions Min Typ Max Min Typ Max Unit DYNAMIC PERFORMANCE Bandwidth ( 3 db) Small Signal V OUT.4 V p-p MHz Large Signal 1 836, V OUT = 2.5 V p-p; 837, V OUT = 3.5 V p-p MHz Bandwidth for.1 db Flatness V OUT.4 V p-p 836, = 14 Ω; 837, = 274 Ω MHz Slew Rate, Average +/ V OUT = 4 V Step, 1 9% V/µs Rise/Fall Time V OUT =.5 V Step, 1 9% ns V OUT = 4 V Step, 1 9% ns Settling Time To.1% V OUT = 2 V Step 1 1 ns To.1% V OUT = 2 V Step ns HARMONIC/NOISE PERFORMANCE 2nd Harmonic Distortion 2 V p-p; 2 MHz, R L = 1 Ω dbc R L = 5 Ω dbc 3rd Harmonic Distortion 2 V p-p; 2 MHz, R L = 1 Ω dbc R L = 5 Ω dbc 3rd Order Intercept 25 MHz dbm Noise Figure R S = 5 Ω db Input Voltage Noise 1 MHz to 2 MHz nv Hz Input Current Noise 1 MHz to 2 MHz pa Hz Average Equivalent Integrated Input Noise Voltage.1 MHz to 2 MHz 95 6 µv rms Differential Gain Error (3.58 MHz) R L = 15 Ω % Differential Phase Error (3.58 MHz) R L = 15 Ω Degree Phase Nonlinearity DC to 1 MHz Degree CLAMP PERFORMANCE Clamp Voltage Range 2 V CH or V CL ± 3.3 ± 3.9 ± 3.3 ± 3.9 V Clamp Accuracy 2 Overdrive, V CH = +2 V, V CL = 2 V ± 3 ± 1 ± 3 ± 1 mv T MIN T MAX ± 2 ± 2 mv Clamp Nonlinearity Range mv Clamp Input Bias Current ( or ) 836,, L = ± 1 V; 837,, L = ±.5 V ± 4 ± 6 ± 5 ± 7 µa T MIN T MAX ± 8 ± 9 µa Clamp Input Bandwidth ( 3 db) V CH or V CL = 2 V p-p MHz Clamp Overshoot 2 Overdrive, V CH or V CL = 2 V p-p % Overdrive Recovery 2 Overdrive ns DC PERFORMANCE 4, R L = 15 Ω Input Offset Voltage mv T MIN T MAX 11 1 mv Offset Voltage Drift ± 1 ± 1 µv/ C Input Bias Current µa T MIN T MAX µa Input Offset Current µa T MIN T MAX 5 5 µa Common-Mode Rejection Ratio V CM = ±2 V db Open-Loop Gain V OUT = ± 2.5 V db T MIN T MAX 4 46 db INPUT CHARACTERISTICS Input Resistance 5 5 kω Input Capacitance pf Input Common-Mode Voltage Range ± 2.5 ± 2.5 V OUTPUT CHARACTERISTICS Output Voltage Range, R L = 15 Ω ±3.2 ± 3.9 ± 3.2 ± 3.9 V Output Current 7 7 ma Output Resistance.3.3 Ω Short Circuit Current ma POWER SUPPLY Operating Range ± 3. ± 5. ± 6. ± 3. ± 5. ± 6. V Quiescent Current ma T MIN T MAX ma Power Supply Rejection Ratio T MIN T MAX db NOTES 1 See Max Ratings and Theory of Operation sections of data sheet. 2 See Max Ratings. 3 Nonlinearity is defined as the voltage delta between the set input clamp voltage ( or ) and the voltage at which V OUT starts deviating from V IN (see Figure 73). 4 Measured at A V = 5. 5 Measured with respect to the inverting input. Specifications subject to change without notice. 2

3 AD836/AD837 ABSOLUTE MAXIMUM RATINGS 1 Supply Voltage V Voltage Swing Bandwidth Product V-MHz V IN V V IN V Internal Power Dissipation 2 Plastic DIP Package (N) Watts Small Outline Package (SO) Watts Input Voltage (Common Mode) ± V S Differential Input Voltage ± 1.2 V Output Short Circuit Duration Observe Power Derating Curves Storage Temperature Range N, R C to +125 C Operating Temperature Range (A Grade)... 4 C to +85 C Lead Temperature Range (Soldering 1 sec) C NOTES 1 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Specification is for device in free air: 8-Lead Plastic DIP: θ JA = 9 C/W 8-Lead SOIC: θ JA = 155 C/W 8-Lead Cerdip: θ JA = 11 C/W. METALIZATION PHOTO Dimensions shown in inches and (mm). Connect Substrate to. IN MAXIMUM POWER DISSIPATION The maximum power that can be safely dissipated by these devices is limited by the associated rise in junction temperature. The maximum safe junction temperature for plastic encapsulated devices is determined by the glass transition temperature of the plastic, approximately 15 C. Exceeding this limit temporarily may cause a shift in parametric performance due to a change in the stresses exerted on the die by the package. Exceeding a junction temperature of 175 C for an extended period can result in device failure. While the AD836 and AD837 are internally short circuit protected, this may not be sufficient to guarantee that the maximum junction temperature (15 C) is not exceeded under all conditions. To ensure proper operation, it is necessary to observe the maximum power derating curves. MAXIMUM POWER DISSIPATION Watts LEAD PLASTIC DIP PACKAGE 8-LEAD SOIC PACKAGE T J = +15 C AMBIENT TEMPERATURE C.46 (1.17) 6 OUT Figure 2. Plot of Maximum Power Dissipation vs. Temperature ORDERING GUIDE.46 (1.17) 3 +IN IN +IN (1.27) 836 AD AD (1.27) CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD836/AD837 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. OUT Temperature Package Package Model Range Description Option AD836AN 4 C to +85 C Plastic DIP N-8 AD836AR 4 C to +85 C SOIC SO-8 AD836AR-REEL 4 C to +85 C 13" Tape and Reel SO-8 AD836AR-REEL7 4 C to +85 C 7" Tape and Reel SO-8 AD836ACHIPS 4 C to +85 C Die AD836-EB Evaluation Board MPA 55 C to +125 C Cerdip Q-8 AD837AN 4 C to +85 C Plastic DIP N-8 AD837AR 4 C to +85 C SOIC SO-8 AD837AR-REEL 4 C to +85 C 13" Tape and Reel SO-8 AD837AR-REEL7 4 C to +85 C 7" Tape and Reel SO-8 AD837ACHIPS 4 C to +85 C Die AD837-EB Evaluation Board WARNING! ESD SENSITIVE DEVICE

4 AD836/AD837 AD836 Typical Characteristics PULSE GENERATOR 1 F PULSE GENERATOR + 1 F T R /T F = 35ps T R /T F = 35ps V IN AD836 1 F V OUT R L = 1 V IN AD836 1 F V OUT R L = 1 TPC 1. Noninverting Configuration, G = +1 TPC 4. Noninverting Clamp Configuration, G = +1 TPC 2. Large Signal Transient Response; V O = 4 V p-p, G = +1, = 14 Ω TPC 5. Clamped Large Signal Transient Response (2 Overdrive); V O = 2 V p-p, G = +1, = 14 Ω, = +1 V, = 1 V TPC 3. Small Signal Transient Response; V O = 4 mv p-p, G = +1, = 14 Ω TPC 6. Clamped Small Signal Transient Response (2 Overdrive); V O = 4 mv p-p, G = +1, = 14 Ω, = +.2 V, =.2 V 4

5 AD836/AD837 AD837 Typical Characteristics PULSE GENERATOR T R /T F = 35ps 1 F PULSE GENERATOR T R /T F = 35ps + 1 F R IN R IN V IN AD837 1 F V OUT R L = 1 V IN AD837 1 F V OUT R L = 1 TPC 7. Noninverting Configuration, G = +2 TPC 1. Noninverting Clamp Configuration, G = +2 TPC 8. Large Signal Transient Response; V O = 4 V p-p, G = +2, = R IN = 274 Ω TPC 11. Clamped Large Signal Transient Response (2 Overdrive); V O = 2 V p-p, G = +2, = R IN = 274 Ω, = +.5 V, =.5 V TPC 9. Small Signal Transient Response; V O = 4 mv p-p, G = +2, = R IN = 274 Ω TPC 12. Clamped Small Signal Transient Response (2 Overdrive); V O = 4 mv p-p, G = +2, = R IN = 274 Ω, = +.1 V, =.1 V 5

6 AD836/AD837 AD836 Typical Characteristics GAIN db V O = 3mV p-p R L = dB BANDWIDTH MHz R PACKAGE R L = 1 GAIN = +1 N PACKAGE AD836 R L M 1M 1M 1G VALUE OF FEEDBACK RESISTOR ( ) TPC 13. AD836 Small Signal Frequency Response, G = +1 TPC 16. AD836 Small Signal 3 db Bandwidth vs GAIN db V O = 3mV p-p R L = OUTPUT db V O = 2.5V p-p R L = 1 5 = 5 TO 25 BY M 1M 1M 1G 8 1M 1M 1M 1G TPC 14. AD836.1 db Flatness, N Package (for R Package Add 2 Ω to ) TPC 17. AD836 Large Signal Frequency Response, G = OPEN -LOOP GAIN db GAIN k 1k 1M 1M PHASE 1M G PHASE MARGIN Degrees GAIN db k V O = 3mV p-p R L = 1 1V 14 AD836 (V IN ) 1 (V O ) 1M 1M 1M 1G TPC 15. AD836 Open-Loop Gain and Phase Margin vs. Frequency, R L = 1 Ω TPC 18. AD836 Clamp Input Bandwidth,, 6

7 AD836/AD837 HARMONIC DISTORTION dbc k V O = 2V p-p R L = 5 G = +1 1k 2ND HARMONIC 3RD HARMONIC 1M 1M 1M DIFF PHASE Degrees DIFF GAIN % st 2nd 3rd 4th 5th 6th 7th 8th 9th 1th 11th 1st 2nd 3rd 4th 5th 6th 7th 8th 9th 1th 11th TPC 19. AD836 Harmonic Distortion vs. Frequency, R L = 5 Ω TPC 22. AD836 Differential Gain and Phase Error, G = +1, R L = 15 Ω, F = 3.58 MHz HARMONIC DISTORTION dbc V O = 2V p-p R L = 1 G = +1 2ND HARMONIC 3RD HARMONIC ERROR % k 1k 1M 1M 1M SETTLING TIME ns TPC 2. AD836 Harmonic Distortion vs. Frequency, R L = 1 Ω TPC 23. AD836 Short-Term Settling Time to.1%, 2 V Step, G = +1, R L = 1 Ω INTERCEPT +dbm ERROR % FREQUENCY MHz SETTLING TIME - s TPC 21. AD836 Third Order Intercept vs. Frequency TPC 24. AD836 Long-Term Settling Time, 2 V Step, G = +1, R L = 1 Ω 7

8 AD836/AD837 AD837 Typical Characteristics GAIN db V O = 3mV p-p R L = dB BANDWIDTH MHz R L = 1 GAIN = +2 R PACKAGE 49.9 N PACKAGE R IN AD837 1 R L M 1M 1M 1G VALUE OF,R IN TPC 25. AD837 Small Signal Frequency Response, G = +2 TPC 28. AD837 Small Signal 3 db Bandwidth vs., R IN GAIN db M V O = 3.mV p-p R L = M 1M 1G TPC 26. AD837.1 db Flatness, N Package (for R Package Add 2 Ω to ) GAIN db M V O = 3.5 V p-p R L = 1 = 75 RF = 75 TO 475 BY 1 = 475 1M 1M 1G TPC 29. AD837 Large Signal Frequency Response, G = +2 OPEN -LOOP GAIN db GAIN PHASE k 1k 1M 1M 1M 1G TPC 27. AD837 Open-Loop Gain and Phase Margin vs. Frequency, R L = 1 Ω PHASE MARGIN Degrees GAIN db V O = 3mV p-p R L = V 274 AD837 1 (V IN ) (V O ) 2 1k 1M 1M 1M 1G TPC 3. AD837 Clamp Input Bandwidth,, 8

9 AD836/AD837 HARMONIC DISTORTION dbc k V O = 2V p-p R L = 5 G = +2 1k 2ND HARMONIC 3RD HARMONIC 1M 1M 1M DIFF PHASE Degrees DIFF GAIN % st 2nd 3rd 4th 5th 6th 7th 8th 9th 1th 11th 1st 2nd 3rd 4th 5th 6th 7th 8th 9th 1th 11th TPC 31. AD837 Harmonic Distortion vs. Frequency, R L = 5 Ω TPC 34. AD837 Differential Gain and Phase Error G = +2, R L = 15 Ω, F = 3.58 MHz 3.5 HARMONIC DISTORTION dbc V O = 2V p-p R L = 1 G = +2 2ND HARMONIC 3RD HARMONIC ERROR % k 1k 1M 1M 1M SETTLING TIME ns TPC 32. AD837 Harmonic Distortion vs. Frequency, R L = 1 Ω TPC 35. AD837 Short-Term Settling Time to.1%, 2 V Step, G = +2, R L = 1 Ω INTERCEPT +dbm ERROR % FREQUENCY MHz SETTLING TIME s TPC 33. AD837 Third Order Intercept vs. Frequency TPC 36. AD837 Long-Term Settling Time 2 V Step, R L = 1 Ω 9

10 AD836/AD837 Typical Characteristics INPUT NOISE VOLTAGE nv/ Hz INPUT NOISE VOLTAGE nv/ Hz k 1k 1k k 1k 1k TPC 37. AD836 Noise vs. Frequency TPC 4. AD837 Noise vs. Frequency PSRR db PSRR 65 +PSRR k 1k 1M 1M 1M 1G TPC 38. AD836 PSRR vs. Frequency PSRR db PSRR PSRR k 1k 1M 1M 1M 1G TPC 41. AD837 PSRR vs. Frequency V CM = 1V R L = V CM = 1V R L = 1 CMRR db 7 6 CMRR db k 1M 1M 1M 1G 2 1k 1M 1M 1M 1G TPC 39. AD836 CMRR vs. Frequency TPC 42. AD837 CMRR vs. Frequency 1

11 AD836/AD837 1k R OUT G = +1 OPEN -LOOP GAIN V/ V AD837 +A OL A OL AD836 +A OL.1.1M 1.M 1M 1M 3M A OL JUNCTION TEMPERATURE C TPC 43. AD836 Output Resistance vs. Frequency TPC 46. Open-Loop Gain vs. Temperature 1k 74 1 G = PSRR +PSRR AD837 R OUT 1 1 PSRR db PSRR AD837 AD PSRR.1 62 AD M 1.M 1M 1M 3M JUNCTION TEMPERATURE C TPC 44. AD837 Output Resistance vs. Frequency TPC 47. PSRR vs. Temperature V OUT R L =15 95 V CM = 2V OUTPUT SWING Volts V OUT +V OUT V OUT R L = 5 CMRR db JUNCTION TEMPERATURE C TPC 45. AD836/AD837 Output Swing vs. Temperature JUNCTION TEMPERATURE C TPC 48. AD836/AD837 CMRR vs. Temperature 11

12 AD836/AD837 Typical Characteristics SUPPLY CURRENT ma AD836, V S = 6V AD837, V S = 6V AD836, AD837, SHORT CIRCUIT CURRENT ma AD837 AD836 AD837 SINK AD836 SOURCE JUNCTION TEMPERATURE C TPC 49. Supply Current vs. Temperature JUNCTION TEMPERATURE C TPC 52. Short Circuit Current vs. Temperature INPUT OFFSET VOLTAGE mv AD837 AD836 V S = 6V V S = 6V INPUT BIAS CURRENT A AD836 AD837 IB +IB IB +IB JUNCTION TEMPERATURE C TPC 5. Input Offset Voltage vs. Temperature JUNCTION TEMPERATURE C TPC 53. Input Bias Current vs. Temperature COUNT WAFER LOTS COUNT = 632 FREQ. DIST INPUT OFFSET VOLTAGE mv TPC 51. AD836 Input Offset Voltage Distribution COUNT WAFER LOTS COUNT = 853 FREQ. DIST INPUT OFFSET VOLTAGE mv TPC 54. AD837 Input Offset Voltage Distribution 12

13 Clamp Characteristics AD836/AD837 INPUT ERROR VOLTAGE mv V CL = 3V V CL = 2V V CL = 1V V CH = +1V AD836, A CL = +1 AD837, A CL = +2 AD836 AD837 V CH = +2V V CH = +3V OUTPUT VOLTAGE Volts TPC 55. Input Error Voltage vs. Clamped Output Voltage HARMONIC DISTORTION dbc AD837 2ND HARMONIC AD836 2ND HARMONIC AD836 3RD HARMONIC AD837 3RD HARMONIC AD836 AD837 +1V +.5V 1V.5V G +1V +2V ABSOLUTE VALUE OF OUTPUT VOLTAGE Volts TPC 58. Harmonic Distortion as Output Approaches Clamp Voltage; V O = 2 V p-p, R L = 1, f = 2 MHz 2 8 NONLINEARITY mv = + 1V = 1V CLAMP INPUT BIAS CURRENT A I BL POSITIVE I BH, I BL DENOTES CURRENT FLOW INTO CLAMP INPUTS, I BH INPUT VOLTAGE A V Volts INPUT CLAMP VOLTAGE (, ) Volts TPC 56. AD836/AD837 Nonlinearity Near Clamp Voltage TPC 59. AD836/AD837 Clamp Input Bias Current vs. Input Clamp Voltage +2V +2V +1V +1V V V REF TPC 57. AD836 Clamp Overdrive (2 ) Recovery REF TPC 6. AD837 Clamp Overdrive (2 ) Recovery 13

14 AD836/AD837 Clamp Characteristics ERROR %.1.1 ERROR % SETTLING TIME ns SETTLING TIME ns TPC 61. AD836 Clamp Settling (.1%), = +1 V, = 1 V, 2 Overdrive TPC 64. AD837 Clamp Settling (.1%), = +.5 V, =.5 V, 2 Overdrive ERROR %.1.1 ERROR % SETTLING TIME ns SETTLING TIME ns TPC 62. AD836 Clamp Recovery Settling Time (High), from +2 Overdrive to V TPC 65. AD837 Clamp Recovery Settling Time (High), from +2 Overdrive to V ERROR %.1.1 ERROR % SETTLING TIME ns SETTLING TIME ns TPC 63. AD836 Clamp Recovery Settling Time (Low), from 2 Overdrive to V TPC 66. AD837 Clamp Recovery Settling Time (Low), from 2 Overdrive to V 14

15 AD836/AD837 THEORY OF OPERATION General The AD836 and AD837 are wide bandwidth, voltage feedback clamp amplifiers. Since their open-loop frequency response follows the conventional 6 db/octave roll-off, their gain bandwidth product is basically constant. Increasing their closed-loop gain results in a corresponding decrease in small signal bandwidth. This can be observed by noting the bandwidth specification, between the AD836 (gain of 1) and AD837 (gain of 2). The AD836/ AD837 typically maintain 65 degrees of phase margin. This high margin minimizes the effects of signal and noise peaking. While the AD836 and AD837 can be used in either an inverting or noninverting configuration, the clamp function will only work in the noninverting mode. As such, this section shows connections only in the noninverting configuration. Applications that require an inverting configuration will be discussed in the Applications section. In applications that do not require clamping, Pins 5 and 8 (respectively and ) may be left floating. See Input Clamp Amp Operation and Applications sections otherwise. Feedback Resistor Choice The value of the feedback resistor is critical for optimum performance on the AD836 (gain +1) and less critical as the gain increases. Therefore, this section is specifically targeted at the AD836. At minimum stable gain (+1), the AD836 provides optimum dynamic performance with = 14 Ω. This resistor acts only as a parasitic suppressor against damped RF oscillations that can occur due to lead (input, feedback) inductance and parasitic capacitance. This value of provides the best combination of wide bandwidth, low parasitic peaking, and fast settling time. In fact, for the same reasons, a 1 13 Ω resistor should be placed in series with the positive input for other AD836 noninverting configurations. The correct connection is shown in Figure 3. V IN R G = 1+ F R G 1-13 R TERM AD836/ AD837 1 F 1 F V OUT This estimation loses accuracy for gains of +2/ 1 or lower due to the amplifier s damping factor. For these low gain cases, the bandwidth will actually extend beyond the calculated value (see Closed-Loop BW plots, TPCs 13 and 25). Pulse Response Unlike a traditional voltage feedback amplifier, where the slew speed is dictated by its front end dc quiescent current and gain bandwidth product, the AD836 and AD837 provide on demand current that increases proportionally to the input step signal amplitude. This results in slew rates (12 V/µs) comparable to wideband current feedback designs. This, combined with relatively low input noise current (2.1 pa/ Hz), gives the AD836 and AD837 the best attributes of both voltage and current feedback amplifiers. Large Signal Performance The outstanding large signal operation of the AD836 and AD837 is due to a unique, proprietary design architecture. In order to maintain this level of performance, the maximum 35 V-MHz product must be observed, 1 MHz, V O 3.5 V p-p). Power Supply and Input Clamp Bypassing Adequate power supply bypassing can be critical when optimizing the performance of a high frequency circuit. Inductance in the power supply leads can form resonant circuits that produce peaking in the amplifier s response. In addition, if large current transients must be delivered to the load, then bypass capacitors (typically greater than 1 µf) will be required to provide the best settling time and lowest distortion. A parallel combination of at least 4.7 µf, and between.1 µf and.1 µf, is recommended. Some brands of electrolytic capacitors will require a small series damping resistor 4.7 Ω for optimum results. When the AD836 and AD837 are used in clamping mode, and a dc voltage is connected to clamp inputs and, a.1 µf bypassing capacitor is required between each input pin and ground in order to maintain stability. Driving Capacitive Loads The AD836 and AD837 were designed primarily to drive nonreactive loads. If driving loads with a capacitive component is desired, the best frequency response is obtained by the addition of a small series resistance as shown in Figure 4. The accompanying graph shows the optimum value for R SERIES vs. capacitive load. It is worth noting that the frequency response of the circuit when driving large capacitive loads will be dominated by the passive roll-off of R SERIES and C L. For capacitive loads of 6 pf or less, no R SERIES is necessary. R G R IN Figure 3. Noninverting Operation For general voltage gain applications, the amplifier bandwidth can be closely estimated as: R IN AD836/ AD837 R SERIES R L 1k C L f 3 db 2π 1+ ω O R G Figure 4. Driving Capacitive Loads 15

16 AD836/AD837 R SERIES C L pf Figure 5. Recommended R SERIES vs. Capacitive Load INPUT CLAMPING AMPLIFIER OPERATION The key to the AD836 and AD837 s fast, accurate clamp and amplifier performance is their unique patent pending CLAMPIN input clamp architecture. This new design reduces clamp errors by more than 1 over previous output clamp based circuits, as well as substantially increasing the bandwidth, precision and versatility of the clamp inputs. Figure 6 is an idealized block diagram of the AD836 connected as a unity gain voltage follower. The primary signal path comprises A1 (a 12 V/µs, 24 MHz high voltage gain, differential to single-ended amplifier) and A2 (a G = +1 high current gain output buffer). The AD837 differs from the AD836 only in that A1 is optimized for closed-loop gains of two or greater. The CLAMPIN section is comprised of comparators C H and C L, which drive switch S1 through a decoder. The unity-gain buffers in series with +V IN,, and inputs isolate the input pins from the comparators and S1 without reducing bandwidth or precision. The two comparators have about the same bandwidth as A1 (24 MHz), so they can keep up with signals within the useful bandwidth of the AD836. To illustrate the operation of the CLAMPIN circuit, consider the case where is referenced to 1 V, is open, and the AD836 is set for a gain of +1, by connecting its output back to its inverting input through the recommended 14 Ω feedback resistor. Note that the main signal path always operates closed loop, since the CLAMPIN circuit only affects A1 s noninverting input. If a V to 2 V voltage ramp is applied to the AD836 s +V IN for the connection just described, V OUT should track +V IN perfectly up to 1 V, then should limit at exactly 1 V as +V IN continues to 2 V. In practice, the AD836 comes close to this ideal behavior. As the +V IN input voltage ramps from zero to 1 V, the output of the high limit comparator C H starts in the off state, as does the output of C L. When +V IN just exceeds V IN (ideally, by say 1 µv, practically by about 18 mv), C H changes state, switching S1 from A to B reference level. Since the + input of A1 is now connected to, further increases in +V IN have no effect on the AD836 s output voltage. In short, the AD836 is now operating as a unity-gain buffer for the input, as any variation in, for > 1 V, will be faithfully reproduced at V OUT. Operation of the AD836 for negative input voltages and negative clamp levels on is similar, with comparator C L controlling S1. Since the comparators see the voltage on the +V IN pin as their common reference level, then the voltage and are defined as High or Low with respect to +V IN. For example, if V IN is set to zero volts, is open, and is +1 V, comparator C L will switch S1 to C, so the AD836 will buffer the voltage on and ignore +V IN. The performance of the AD836 and AD837 closely matches the ideal just described. The comparator s threshold extends from 6 mv inside the clamp window defined by the voltages on and to 6 mv beyond the window s edge. Switch S1 is implemented with current steering, so that A1 s +input makes a continuous transition from say, V IN to as the input voltage traverses the comparator s input threshold from.9 V to 1. V for = 1. V. The practical effect of these nonidealities is to soften the transition from amplification to clamping modes, without compromising the absolute clamp limit set by the CLAMPIN circuit. Figure 7 is a graph of V OUT vs. V IN for the AD836 and a typical output clamp amplifier. Both amplifiers are set for G = +1 and = 1 V. The worst case error between V OUT (ideally clamped) and V OUT (actual) is typically 18 mv times the amplifier closed-loop gain. This occurs when V IN equals (or ). As V IN goes above and/or below this limit, V OUT will settle to within 5 mv of the ideal value. In contrast, the output clamp amplifier s transfer curve typically will show some compression starting at an input of.8 V, and can have an output voltage as far as 2 mv over the clamp limit. In addition, since the output clamp in effect causes the amplifier to operate open loop in clamp mode, the amplifier s output impedance will increase, potentially causing additional errors. The AD836 s and AD837 s CLAMPIN input clamp architecture works only for noninverting or follower applications and, since it operates on the input, the clamp voltage levels and, and input error limits will be multiplied by the amplifier s V IN +V IN C H C L B A C 14 S1 A1 A2 +1 S1 A B C V IN > 1 V IN 1 V IN < 1 Figure 6. AD836/AD837 Clamp Amp System V OUT 16

17 AD836/AD837 closed-loop gain at the output. For instance, to set an output limit of ± 1 V for an AD837 operating at a gain of 3., and would need to be set to V and.333 V, respectively. The only restriction on using the AD836 s and AD837 s +V IN,, pins as inputs is that the maximum voltage difference between +V IN and or should not exceed 6.3 V, and all three voltages be within the supply voltage range. For example, if is set at 3 V, then V IN should not exceed +3.3 V. OUTPUT VOLTAGE V OUT CLAMP ERROR 25mV AD836 AD836 OUTPUT CLAMP AMP CLAMP ERROR >2mV OUTPUT CLAMP INPUT VOLTAGE +V IN Figure 7. Output Clamp Error vs. Input Clamp Error AD836/AD837 APPLICATIONS The AD836 and AD837 use a unique input clamping circuit to perform the clamping function. As a result, they provide the clamping function better than traditional output clamping devices and provide additional flexibility to perform other unique applications. There are, however, some restrictions on circuit configurations; and some calculations need to be performed in order to figure the clamping level, as a result of clamping being performed at the input stage. The major restriction on the clamping feature of the AD836/ AD837 is that clamping occurs only when using the amplifiers in the noninverting mode. To clamp in an inverting circuit, an additional inverting gain stage is required. Another restriction is that be greater than, and that each be within the output voltage range of the amplifier (±3.9 V). can go below ground and can go above ground as long as is kept higher than. Unity Gain Clamping The simplest circuit for calculating the clamp levels is a unity gain follower as shown in Figure 8. In this case, the AD836 should be used since it is compensated for noninverting unity gain. This circuit will clamp at an upper voltage set by (the voltage applied to Pin 8) and a lower voltage set by (the voltage applied to Pin 5). Clamping with Gain Figure 9 shows an AD837 configured for a noninverting gain of two. The AD837 is used in this circuit since it is compensated for gains of two or greater and provides greater bandwidth. In this case, the high clamping level at the output will occur at V IN 13 AD836 +5V 5V 14 1 F 1 F V OUT Figure 8. Unity Gain Noninverting Clamp 2 and the low clamping level at the output will be 2. The equations governing the output clamp levels in circuits configured for noninverting gain are: V CH = G V CL = G where: V CH is the high output clamping level V CL is the low output clamping level G is the gain of the amplifier configuration is the high input clamping level (Pin 8) is the low input clamping level (Pin 5) *Amplifier offset is assumed to be zero. V IN 49.9 R G AD837 +5V 5V F 1 F V OUT Figure 9. Gain of Two Noninverting Clamp 17

18 AD836/AD837 +5V 86 1µF +5V AD78 2.5V V IN.5V to +.5V R AD V 2V to V 1 F 1 F 1N CLAMPING RANGE 2.1V to +.1V AD92 V IN = 2V TO V SUBSTRATE DIODE 5V 86 R V R V Figure 1. Gain of Two, Noninverting with Offset AD837 Driving an AD92 8-Bit, 125 MSPS A/D Converter Clamping with an Offset Some op amp circuits are required to operate with an offset voltage. These are generally configured in the inverting mode where the offset voltage can be summed in as one of the inputs. Since AD836/AD837 clamping does not function in the inverting mode, it is not possible to clamp with this configuration. Figure 1 shows a noninverting configuration of an AD837 that provides clamping and also has an offset. The circuit shows the AD837 as a driver for an AD92, an 8-bit, 125 MSPS A/D converter and illustrates some of the considerations for using an AD837 with offset and clamping. The analog input range of the AD92 is from ground to 2 V. The input should not go more than.5 V outside this range in order to prevent disruptions to the internal workings of the A/D and to avoid drawing excess current. These requirements make the AD837 a prime candidate for signal conditioning. When an offset is added to a noninverting op amp circuit, it is fed in through a resistor to the inverting input. The result is that the op amp must now operate at a closed-loop gain greater than unity. For this circuit a gain of two was chosen which allows the use of the AD837. The feedback resistor, R2, is set at 31 Ω for optimum performance of the AD837 at a gain of two. There is an interaction between the offset and the gain, so some calculations must be performed to arrive at the proper values for R1 and R3. For a gain of two the parallel combination of resistors R1 and R3 must be equal to the feedback resistor R2. Thus R1 R3/R1 + R3 = R2 = 31 Ω The reference used to provide the offset is the AD78 whose output is 2.5 V. This must be divided down to provide the 1 V offset desired. Thus 2.5 V R1/(R1 + R3) = 1 V When the two equations are solved simultaneously we get R1 = 499 Ω and R3 = 75 Ω (using closest 1% resistor values in all cases). This positive 1 V offset at the input translates to a 1 V offset at the output. The usable input signal swing of the AD92 is 2 V p-p. This is centered about the 1 V offset making the usable signal range from V to 2 V. It is desirable to clamp the input signal so that it goes no more than 1 mv outside of this range in either direction. Thus, the high clamping level should be set at +.1 V and the low clamping level should be set at 2.1 V as seen at the input of the AD92 (output of AD837). Because the clamping is done at the input stage of the AD837, the clamping level as seen at the output is affected by not only the gain of the circuit as previously described, but also by the offset. Thus, in order to obtain the desired clamp levels, must be biased at +.55 V while must be biased at.55 V. The clamping levels as seen at the output can be calculated by the following: V CH = V OFF + G V CL = V OFF + G Where V OFF is the offset voltage that appears at the output. The resistors used to generate the voltages for and should be kept to a minimum in order to reduce errors due to clamp bias current. This current is dependent on and (see TPC 59) and will create a voltage drop across whatever resistance is in series with each clamp input. This extra error voltage is multiplied by the closed-loop gain of the amplifier and can be substantial, especially in high closed-loop gain configurations. A.1 µf bypass capacitor should be placed between input clamp pins and and ground to ensure stable operation. The 1N5712 Schottky diode is used for protection from forward biasing the substrate diode in the AD92 during power-up transients. Programmable Pulse Generator The AD836/AD837 s clamp output can be set accurately and has a well controlled flat level. This along with wide bandwidth and high slew rate make them very well suited for programmable level pulse generators. Figure 11 is a schematic for a pulse generator that can directly accept TTL generated timing signals for its input and generate pulses at the output up to 24 V p-p with 25 V/µs slew rate. The output levels can be programmed to anywhere in the range 12 V to +12 V. 18

19 AD836/AD837 +5V TTL IN k 15V AD837 1 F 1 AD V 1 F PULSE OUT 1 1 5V 1 F 1 F V Figure 11. Programmable Pulse Generator The circuit uses an AD837 operating at a gain of two with an AD811 to boost the output to the ± 12 V range. The AD811 was chosen for its ability to operate with ± 15 V supplies and its high slew rate. R1 and R2 act as a level shifter to make the TTL signal levels be approximately symmetrical above and below ground. This ensures that both the high and low logic levels will be clamped by the AD837. For well controlled signal levels in the output pulse, the high and low output levels should result from the clamping action of the AD837 and not be controlled by either the high or low logic levels passing through a linear amplifier. For good rise and fall times at the output pulse, a logic family with high speed edges should be used. The high logic levels are clamped at two times the voltage at, while the low logic levels are clamped at two times the voltage at. The output of the AD837 is amplified by the AD811 operating at a gain of 5. The overall gain of 1 will cause the high output level to be 1 times the voltage at, and the low output level to be 1 times the voltage at. High Speed, Full-Wave Rectifier The clamping inputs are additional inputs to the input stage of the op amp. As such they have an input bandwidth comparable to the amplifier inputs and lend themselves to some unique functions when they are driven dynamically. Figure 12 is a schematic for a full-wave rectifier, sometimes called an absolute value generator. It works well up to 2 MHz and can operate at significantly higher frequencies with some degradation in performance. The distortion performance is significantly better than diode based full-wave rectifiers, especially at high frequencies. 1 AD837 +5V 1 F V OUT = V IN The circuit is configured as an inverting amplifier with a gain of one. The input drives the inverting amplifier and also directly drives, the lower level clamping input. The high level clamping input,, is left floating and plays no role in this circuit. When the input is negative, the amplifier acts as a regular unitygain inverting amplifier and outputs a positive signal at the same amplitude as the input with opposite polarity. is driven negative by the input, so it performs no clamping action, because the positive output signal is always higher than the negative level driving. When the input is positive, the output result is the sum of two separate effects. First, the inverting amplifier multiplies the input by 1 because of its unity-gain inverting configuration. This effectively produces an offset as explained above, but with a dynamic level that is equal to 1 times the input. Second, although the positive input is grounded (through 1 Ω), the output is clamped at two times the voltage applied to (a positive, dynamic voltage in this case). The factor of two is because the noise gain of the amplifier is two. The sum of these two actions results in an output that is equal to unity times the input signal for positive input signals, see Figure 13. For a input/output scope photo with an input signal of 2 MHz and amplitude ±1 V, see Figure 14. LOWER CLAMPING LEVEL WITH NO NEG INPUT INPUT FULL WAVE RECTIFIED OUTPUT V IN R G V 1 F 1 INPUT OUTPUT Figure 13. LOWER CLAMPING LEVEL Figure 12. Full-Wave Rectifier 19

20 AD836/AD837 Figure 14. Full-Wave Rectifier Scope Thus for either positive or negative input signals, the output is unity times the absolute value of the input signal. The circuit can be easily configured to produce the negative absolute value of the input by applying the input to instead of. The circuit can get to within about 4 mv of ground during the time when the input crosses zero. This voltage is fixed over a wide frequency range and is a result of the switching between the conventional op amp input and the clamp input. But because there are no diodes to rapidly switch from forward to reverse bias, the performance far exceeds that of diode based full wave rectifiers. The 4 mv offset mentioned can be removed by adding an offset to the circuit. A 27.4 kω input resistor to the inverting input will have a gain of.1, while changing the gain of the circuit by only 1%. A plus or minus 4 V dc level (depending on the polarity of the rectifier) into this resistor will compensate for the offset. Full wave rectifiers are useful in many applications including AM signal detection, high frequency ac voltmeters and various arithmetic operations. Amplitude Modulator In addition to being able to be configured as an amplitude demodulator (AM detector), the AD837 can also be configured as an amplitude modulator as shown in Figure 15. The modulation signal is applied to both the input of a unity gain inverting amplifier and to, the lower clamping input. is biased at.5 V dc. To understand the circuit operation, it is helpful to first consider a simpler circuit. If both and were dc biased at.5 V and the carrier and modulation inputs driven as above, the output would be a 2 V p-p square wave at the carrier frequency riding on a waveform at the modulating frequency. The inverting input (modulation signal) is creating a varying offset to the 2 V p-p square wave at the output. Both the high and low levels clamp at twice the input levels on the clamps because the noise gain of the circuit is two. When is driven by the modulation signal instead of being held at a dc level, a more complicated situation results. The resulting waveform is composed of an upper envelope and a lower envelope with the carrier square wave in between. The upper and lower envelope waveforms are 18 out of phase as in a typical AM waveform. The upper envelope is produced by the upper clamp level being offset by the waveform applied to the inverting input. This offset is the opposite polarity of the input waveform because of the inverting configuration. The lower envelope is produced by the sum of two effects. First, it is offset by the waveform applied to the inverting input as in the case of the simplified circuit above. The polarity of this offset is in the same direction as the upper envelope. Second, the output is driven in the opposite direction of the offset at twice the offset voltage by the modulation signal being applied to. This results from the noise gain being equal to two, and since there is no inversion in this connection, it is opposite polarity from the offset. The result at the output for the lower envelope is the sum of these two effects, which produces the lower envelope of an amplitude modulated waveform. See Figure V CARRIER IN 1 AD837 1 F AM OUT 1 F MODULATION IN R G V Figure 15. Amplitude Modulator The positive input of the AD837 is driven with a square wave of sufficient amplitude to produce clamping action at both the high and low levels. This is the higher frequency carrier signal. Figure 16. AM Waveform The depth of modulation can be modified in this circuit by changing the amplitude of the modulation signal. This changes the amplitude of the upper and lower envelope waveforms. The modulation depth can also be changed by changing the dc bias applied to. In this case the amplitudes of the upper and lower envelope waveforms stay constant, but the spacing between them changes. This alters the ratio of the envelope amplitude to the amplitude of the overall waveform. 2

21 AD836/AD837 Layout Considerations The specified high speed performance of the AD836 and AD837 requires careful attention to board layout and component selection. Proper RF design techniques and low pass parasitic component selection are mandatory. The PCB should have a ground plane covering all unused portions of the component side of the board to provide a low impedance path. The ground plane should be removed from the area near the input pins to reduce stray capacitance. Chip capacitors should be used for supply and input clamp bypassing (see Figure 17). One end should be connected to the ground plane and the other within 1/8 inch of each power and clamp pin. An additional large (.47 µf 1 µf) tantalum electrolytic capacitor should be connected in parallel, though not necessarily so close, to supply current for fast, large signal changes at the output. The feedback resistor should be located close to the inverting input pin in order to keep the stray capacitance at this node to a minimum. Capacitance variations of less than 1 pf at the inverting input will significantly affect high speed performance. Stripline design techniques should be used for long signal traces (greater than about 1 inch). These should be designed with a characteristic impedance of 5 Ω or 75 Ω and be properly terminated at each end. Evaluation Board An evaluation board for both the AD836 and AD837 is available that has been carefully laid out and tested to demonstrate that the specified high speed performance of the device can be realized. For ordering information, please refer to the Ordering Guide. The layout of the evaluation board can be used as shown or serve as a guide for a board layout. IN R G R T R S OPTIONAL AD836/ AD837 1k 1k R O NONINVERTING CONFIGURATION C1.1 F C2.1 F C3 C4 SUPPLY BYPASSING C5 1 F C6 1 F V OUT Figure 17. Noninverting Configurations for Evaluation Boards Table I. AD836A AD837A Gain Gain Component Ω 274 Ω 2 kω 2 kω 274 Ω 2 kω 2 kω R G 274 Ω 221 Ω 2.5 Ω 274 Ω 221 Ω 2.5 Ω R O (Nominal) 49.9 Ω 49.9 Ω 49.9 Ω 49.9 Ω 49.9 Ω 49.9 Ω 49.9 Ω R S 13 Ω 1 Ω 1 Ω 1 Ω 1 Ω 1 Ω 1 Ω R T (Nominal) 49.9 Ω 49.9 Ω 49.9 Ω 49.9 Ω 49.9 Ω 49.9 Ω 49.9 Ω Small Signal BW (MHz)

22 AD836/AD837 Figure 18. Evaluation Board Silkscreen (Top) Figure 2. Board Layout (Solder Side) Figure 19. Evaluation Board Silkscreen (Bottom) Figure 21. Board Layout (Component Side) 22

23 AD836/AD837 OUTLINE DIMENSIONS Dimensions shown in inches and (mm)..43 (1.92).348 (8.84) 8-Lead Plastic DIP (N Package) PIN 1.21 (5.33) MAX.16 (4.6).115 (2.93).1574 (4.).1497 (3.8) 8.22 (.558).14 (.356) PIN 1.98 (.25).4 (.1) SEATING PLANE PIN 1.2.(5.8) MAX.2 (5.8).125 (3.18) (2.54) BSC (.13) MIN.1968 (5.).189 (4.8).7 (1.77).45 (1.15).28 (7.11).24 (6.1).6 (1.52).15 (.38).13 (3.3) MIN SEATING PLANE 8-Lead Plastic SOIC (SO Package) 4.5 (1.27) BSC.192 (.49).138 (.35) (6.2).2284 (5.8).688 (1.75).532 (1.35) 8-Lead Cerdip (Q Package).55 (1.4) MAX.1 (2.54) BSC.45 (1.29) MAX.31 (7.87).22 (5.59).6 (1.52).15 (.38).15 (3.81) MIN SEATING.23 (.58).7 (1.78) PLANE.14 (.36).3 (.76).325 (8.25).3 (7.62).98 (.25).75 (.19) (4.95).115 (2.93).15 (.381).8 (.24) 8.32 (8.13).29 (7.37).196 (.5) (.25).5 (1.27).16 (.41).15 (.38).8 (.2) PRINTED IN U.S.A. C157 12/ (rev. B) 23