# Intended audience This document is intended for design engineers who want to design CCM PFC boost converter.

 To view this video please enable JavaScript, and consider upgrading to a web browser that supports HTML5 video
Save this PDF as:

Size: px
Start display at page:

Download "Intended audience This document is intended for design engineers who want to design CCM PFC boost converter."

## Transcription

1 200 W Design Example Sam Abdel-Rahman Franz Stückler Ken Siu Application Note About this document Scope and purpose This document introduces a design methodology for a Power Factor Correction (PFC) Continuous Conduction Mode (CCM) Boost Converter, including: Equations for design and power losses Selection guide of semiconductor devices and passive components Charts for CoolMOS optimum R DS(ON) selection 200 W design example with calculated and experimental results. Schematics, layout and bill of material Intended audience This document is intended for design engineers who want to design CCM PFC boost converter. Table of Contents Introduction... 2 Power Stage Design... 4 ICE3PCS0G PFC Boost Controller... 7 Efficiency and Power Losses Modeling... 9 Experimental Results Board Design References Symbols used in formulas Application Note Revision.,

2 Design Note DN Introduction Power Factor Correction (PFC) shapes the input current of the power supply to be in synchronization with the mains voltage, in order to maximize the real power drawn from the mains. In a perfect PFC circuit, the input current follows the input voltage as a pure resistor, without any input current harmonics. This document is to introduce a design methodology for the CCM PFC Boost converter, including equations for power losses estimation, selection guide of semiconductor devices and passive components, and a design example with experimental results.. Boost topology Although active PFC can be achieved by several topologies, the boost converter (Figure ) is the most popular topology used in PFC applications, for the following reasons: The line voltage varies from zero to some peak value typically 375 V; hence a step up converter is needed to output a DC bus voltage of 380 V or more. For that reason the buck converter is eliminated, and the buck-boost converter has high switch voltage stress (V in+v o), therefore it is also not the popular one. The boost converter has the filter inductor on the input side, which provides a smooth continuous input current waveform as opposed to the discontinuous input current of the buck or buck-boost topology. The continuous input current is much easier to filter, which is a major advantage of this design because any additional filtering needed on the converter input will increase the cost and reduces the power factor due to capacitive loading of the line. Boost Key Waveforms Vac AC PFC Converter DC Bus DC/DC Converter Load S V_L DT Vin T=/f AC Vac L S D Co DC Bus Ro + Vo - I_L I_S I_D Vin-Vo I_Lmax I_Lmax I_Lmax I_Lmin I_Lmin Figure Structure and key waveforms of a boost converter V o = V in D (CCM operation).2 PFC Modes of Operation The boost converter can operate in three modes: continuous conduction mode (CCM), discontinuous conduction mode (DCM), and critical conduction mode (CrCM). Figure 2 shows modeled waveforms to illustrate the inductor and input currents in the three operating modes, for the same exact voltage and power conditions. By comparing DCM among the others, DCM operation seems simpler than CrCM, since it may operate in constant frequency operation; however DCM has the disadvantage that it has the highest peak current compared to CrCM and also to CCM, without any performance advantage compared to CrCM. For that reason, CrCM is a more common practice design than DCM, therefore, this document will exclude the DCM design. Application Note 2 Revision.,

3 Design Note DN CrCM may be considered a special case of CCM, where the operation is controlled to stay at the boundary between CCM and DCM. CrCM usually uses constant on-time control; the line voltage is changing across the 60 Hz line cycle, the reset time for the boost inductor is varying, and the operating frequency will change as well in order to maintain the boundary mode operation. CrCM dictates the controller to sense the inductor current zero crossing in order to trigger the start of the next switching cycle. The inductor current ripple (or the peak current) in CrCM is twice of the average value, which greatly increases the MOSFET RMS currents and turn-off current. But since every switching cycle starts at zero current, and usually with ZVS operation, turn-on loss of MOSFET is usually eliminated. Also, since the boost rectifier diode turns off at zero current as well, reverse recovery losses and noise in the boost diode are eliminated too, another major advantage of CrCM mode. Still, on the balance, the high input ripple current and its impact on the input EMI filter tends to eliminate CrCM mode for high power designs unless interleaved stages are used to reduce the input HF current ripple. A high efficiency design can be realized that way, but at substantially higher cost. That discussion is beyond the scope of this application note. The power stage equations and transfer functions for CrCM are the same as CCM. The main differences relate to the current ripple profile and switching frequency, which affects RMS current and switching power losses and filter design. CCM operation requires a larger filter inductor compared to CrCM. While the main design concerns for a CrCM inductor are low HF core loss, low HF winding loss, and the stable value over the operating range (the inductor is essentially part of the timing circuit), the CCM mode inductor takes a different approach. For the CCM PFC, the full load inductor current ripple is typically designed to be 20-40% of the average input current. This has several advantages: Peak current is lower, and the RMS current factor with a trapezoidal waveform is reduced compared to a triangular waveform, reducing device conduction losses. Turn-off losses are lower due to switch off at much lower maximum current. The HF ripple current to be smoothed by the EMI filter is much lower in amplitude. On the other side, CCM encounters the turn-on losses in the MOSFET, which can be exacerbated by the boost rectifier reverse recovery loss due to reverse recovery charge, Q rr. For this reason, ultra-fast recovery diodes or silicon carbide Schottky diodes with extreme low Q rr are needed for CCM mode. In conclusion, we can say that for low power applications, the CrCM boost has the advantages in power saving and improving power density. This advantage may extend to medium power ranges, however at some medium power level the low filtering ability and the high peak current starts to become severe disadvantages. At this point the CCM boost starts being a better choice for high power applications. Since this document is intended to support high power PFC applications, therefore a CCM PFC boost converter has been chosen in the application note with detailed design discussions and design examples for demonstration I in ( t) I L ( t) 5 0 I in ( t) I L ( t) 5 0 I in ( t) I L ( t) Continuous Conduction Mode (CCM) Critical Conduction - 4 Discontinuous Conduction Mode - (DCM) 4 t Mode (CrCM) t 0 0 t Figure 2 PFC Inductor and input line current waveforms in the three different operating modes Application Note 3 Revision.,

4 Design Note DN Power Stage Design The following are the converter design and power losses equations for the CCM operated boost. The design example specifications listed in Table will be used for all of the equations calculations. Also the boost converter encounters the maximum current stress and power losses at the minimum line voltage condition (V ac.min ); hence, all design equations and power losses will be calculated using the low-line voltage condition as an extreme case. Table Input voltage Specifications of the power stage Vac 60 Hz Output voltage Maximum power steady state Switching frequency Inductor current ripple Output voltage 20Hz ripple Hold-up time 400 V 200 W 00 khz low line/full load 0 V p-p 6.6 V O,min=340 V Figure 3 Block Schematic for boost power stage with input rectifier 2. Main PFC Inductor Off-the-shelf inductors are available and usable for a first pass design, typically with single layer windings and a permeability drop of 30% or less. In some circumstances it may be desirable to further optimize the inductor configuration in order to meet the requirements for high power factor over a wide input line current range. Many popular PFC controllers use single cycle current loop control, which can provide good performance provided that the inductor remains in CCM operation. At low-line this is no problem, but for the operation in the high-line band (76 Vac to 265 Vac), the operating current will be much lower. If an inductor is used with a nominal stable value of inductance, it works well at low-line but results in DCM mode operation for a significant part of the load range at high-line, with poorer power factor, THDi and higher EMI. A swinging choke (also called powder core), such as Arnold/Micrometals Sendust or Magnetics Inc Kool Mu, can address this if designed with the Application Note 4 Revision.,

5 L (Henry) Design Note DN right energy capability and with full load permeability drop by 75-80%, so that at lighter load the inductance swings up. The filter inductor value and its maximum current are determined based on the specified maximum inductor current ripple as shown below: L = %Ripple V ac.min P o 2 ( 2 V ac.min ) T = V o 0.25 (85V)2 2 85V ( 200W 400V ) Hz = 68.5 μh Eq. I L.max = 2 P o ( + %Ripple ) = V ac.min W 85V ( ) = 22.5 A Eq. 2 Note: Inductor saturation current must be rated at > 22.5A. In this evaluation board design, a 60 μ permeability Kool Mu core from Magnetics Inc. is used. It consists of two stacked of Kool Mμ 77083A7 toroids cores from Magnetics Inc., with 64 turns of.5 mm copper wire, the DC resistance is about 70 mω, and the inductance ranges from 680 µh at no load dropping to about 65 µh at low-line full load, which is very close to the desired value calculated above. Figure 4 Main PFC inductor Since the inductance value is varying across the line and load range, and also across the line cycle, it would be more accurate to model the inductance value as a function of V in, P o and t. in order to obtain better estimation of the switching currents and losses. This specific inductor value was modeled as shown in Figure 5. P o (W) Figure 5 Swinging Inductance accros load/line range Application Note 5 Revision.,

6 Design Note DN Inductor copper loss: The inductor RMS current and the corresponding copper loss are: I L.rms I in.rms = P o 200 W = V ac.min 85 V = 4.2 A Eq. 3 P L.cond = I L.rms 2 DCR = (4.2 A) Ω = 3.95 W Eq. 4 Inductor core losses: At low-line, core loss across the line cycle is found to be close to a sinusoidal shape (Figure 6, left). Therefore a simple and accurate enough method to estimate the average core loss is to calculate the peak core loss at the peak of the line cycle point, then multiply by 2/π. However, at high line, core losses are far from the sinusoidal shape (Figure 6, right), so the aforementioned method is not valid anymore, so it is necessary to model the core loss across the line cycle as a function of time, and then integrate it to obtain the average loss. Figure 6 Inductor core loss across the line cycle: low-line (left), high-line (right) Since in this document we are calculating losses at the minimum line voltage, as the worst case scenario, thus we will use the first method discussed above to obtain the average core loss in the low-line, as detailed below: In order to calculate the core loss, we must calculate the minimum and maximum inductor current and the associated minimum and maximum magnetic force (H), then we can use the fitted equation of that magnetic material to calculate the minimum and maximum magnetic flux (B). Then the AC flux swing can be used to calculate the core loss by using another fitted equation. For 2 stacked Kool Mμ 77083A7 toroids, we get: Path length l e = 98.4 mm Cross section area A e = 2 07 mm 2 Volume V e = mm 3 Using the maximum inductor current calculated in Eq. 2, the magnetic force at the peak of the line cycle can be found as: H max = 0.4 π N I L.max l e (in cm) = 0.4 π 64 turns 22.5A 98.4mm/0 = 84 Oersteds Eq. 5 The minimum inductor current and magnetic force at the peak of the line cycle are: I L.min = P o 2 ( %Ripple ) = V ac.min 2 200W 2 ( 25% 85V 2 ) = 7.42 A Eq. 6 H min = 0.4 π N I L.min l e (in cm) = 0.4 π 64turns 7.42A 98.4mm/0 = Oersteds Eq. 7 Application Note 6 Revision.,

7 Design Note DN Flux density for 60 Koolu material is: x a + b H + c H2 B = ( a + d H + e H 2) Eq. 8 where a =.658e -2 b =.83e -3 c = 4.62e -3 d = 4.7e -3 e = 3.833e -5 x = 0.5 The minimum and maximum flux densities at the peak of the line cycle are: B max = ( a + b H 2 x max + c H max 2 a + d H max + e H ) = kgauss max B min = ( a + b H 2 x min + c H min 2 a + d H min + e H ) = 8.04 kgauss min The AC flux swing at the peak of the line cycle is: B = B max B min 2 = kgaus Eq. 9 Eq. 0 Eq. Peak core loss at the peak of the line cycle is: P core.pk = B 2 ( f ) V e 0 6 = ( 0 3 ) = 0.97 W Eq. 2 Average core loss across the line cycle is: P core.av = P core.pk 2 π = W 2 π = 0.62 W Eq Rectifier Bridge Using a higher rated current bridge can reduce the forward voltage drop V f, which reduces the total power dissipation at a small incremental cost. Also using two parallel bridges is another approach to distribute the thermal dissipation. This is often a sound strategy, as with modern components, the bridge rectifier usually has the highest semiconductor loss for the PFC stage. In this evaluation board design, 2 parallel GSIB2580 are used. The bridge total power loss is calculated using the average input current flowing through two of the bridge rectifying diodes and is shown as: I average = 2 π 2 P o = W = 2.7 A V ac.min π 85 V Eq. 4 P bridge = 2 I average V f.bridge = A V = 25.4 W Eq. 5 The selection on the heat sink is based on the process discussed in chapter 2.6 Heat sink design. 2.3 MOSFET In order to select the optimum MOSFET, one must understand the MOSFET requirements in a CCM boost converter. High voltage MOSFETS have several families based on different technologies. For a boost converter, the following are some major MOSFET selection considerations for high efficiency application design: Low Figure-of Merits - R DS(ON)*Q g and R DS(ON)*E oss Application Note 7 Revision.,

8 Design Note DN Fast turn-on/off switching to reduce the device switching losses Gate plateau near middle of gate drive range to balance turn-on/off losses Low output capacitance Coss for low switching energy and to increase light load efficiency Drain-source breakdown voltage V BR(DSS) to handle spikes/overshoots Low thermal resistance R thjc. Package selection must consider the resulting total thermal resistance from junction to ambient, and the worst case surge dissipation, typically under low-line cycle skipping and recovery into highline while ramping the bulk voltage back up. The body diode commutation speed and reverse recovery charge are not important, since body diode never conducts in the CCM boost converter. Several CoolMOS series can be used for boost applications. C7 followed by CP provides the fastest switching (Figure 7) and best performance, but require careful design in terms of gate driving circuit and PCB layout. The P6/C6/E6 series provides a cost advantage, with easier design. The P6 series approaches CP performance closely at a better price point, and is recommended for new designs that are cost sensitive. In this Evaluation board C7 is used to reach the best efficiency. Figure 7 shows that CoolMOS C7 total gate charge Q g is less than one third of the C6 charge, and less than two thirds of the CP charge. Moreover, it shows gate-drain charge Q gd reduction in the much shorter length of the Miller Plateau compared to previous generations. These improvements in the gate charge profile are an indication of the improvement of the gate driving related losses as well as of the MOSFET switching times and losses. Figure 7 Gate charge and E oss comparision for 4-45 mω CoolMOS C6, CP, C CoolMOS Optimum RDS(ON) Selection Charts Selection of the optimum on-state resistance of a specific CoolMOS series is based on the balancing between switching losses and conduction losses of the device at a targeted load point. This can be done by modeling all losses in software tool such as Mathcad by evaluating different technologies and different values of the on-state resistance. This requires few iterations and entry of several parameters from the datasheet of each part. An alternative way is using the CoolMOS selection charts shown in Figure 9 to Figure 2, specific charts exist for each CoolMOS family, optimized at half load or full load. The following is a guide on how to use these CoolMOS R DS(ON) selection charts. Let s use the specifications in Table as our design example: P o= 200 W, f=00 khz; full load efficiency is critical; CoolMOS C7 is preferred for best performance. Application Note 8 Revision.,

9 Optimum FET On-Resistance (ohm) Max Power Loss (W) for Optimum 5 Vac Design Note DN Step : Find the correct chart, Figure 8 shows the chart for CoolMOS C7 optimized at full load. Step 2: Find the 00 khz curves (blue curves in this example) Step3: Mark the 200 W on the x-axis Step 3: Find the 200 W intersection with the solid blue line for the 00 khz, read the left side y-axis, we find that Ohm is the optimum for CoolMOS C7, then we may choose 045C7. Step 4: Find the 200 W intersection with the dashed blue line for the 00 khz, read the right side y-axis, we find that the Ohm will result in 3.9 W power loss at full load and low line 5 Vac. Ron optimized at full load Best thermal performance 50kHz 00kHz C7 90C7 25C7 095C W 0 065C7 045C7 09C ohm 50kHz 00kHz W 0,000 Output Power (W) Figure 8 Example on how to use the CoolMOS Optimum R DS(on) selection charts. Application Note 9 Revision.,

10 Optimum FET On-Resistance (ohm) Max Power Loss (W) for Optimum 5 Vac Optimum FET On-Resistance (ohm) Max Power Loss (W) for Optimum 5 Vac Optimum FET On-Resistance (ohm) Max Power Loss (W) for Optimum 5 Vac Optimum FET On-Resistance (ohm) Max Power Loss (W) for Optimum 5 Vac Design Note DN Ron optimized at full load Best thermal performance 50kHz 00kHz 00 Ron optimized at 50% load Balanced light/full load efficiency 50kHz 00kHz C7 90C7 225C7 90C7 25C7 095C C7 095C C7 045C7 09C7 50kHz 00kHz 065C7 045C7 09C7 50kHz 00kHz ,000 Output Power (W) Figure 9 CoolMOS C7 Optimum R DS(on) selection charts ,000 Output Power (W) 600CP 520CP 385CP 299CP 250CP 99CP 65CP 25CP 099CP 075CP 045CP 0. Ron optimized at full load Best thermal performance 50kHz 00kHz 50kHz 00kHz CP 520CP 385CP 299CP 250CP 99CP 65CP 25CP 099CP 075CP 045CP 0. Ron optimized at 50% load Balanced light/full load efficiency 50kHz 00kHz 50kHz 00kHz 00 0 Figure , ,000 Output Power (W) Output Power (W) CoolMOS CP Optimum R DS(on) selection charts Application Note 0 Revision.,

11 Optimum FET On-Resistance (ohm) Max Power Loss (W) for Optimum 5 Vac Optimum FET On-Resistance (ohm) Max Power Loss (W) for Optimum 5 Vac Optimum FET On-Resistance (ohm) Max Power Loss (W) for Optimum 5 Vac Optimum FET On-Resistance (ohm) Max Power Loss (W) for Optimum 5 Vac Design Note DN P6 380P6 330P6 280P6 230P6 90P6 60P6 25P6 099P6 070P6 04P6 0. Ron optimized at full load Best thermal performance 50kHz 00kHz 50kHz 00kHz P6 380P6 330P6 280P6 230P6 90P6 60P6 25P6 099P6 070P6 04P6 0. Ron optimized at 50% load Balanced light/full load efficiency 50kHz 00kHz 50kHz 00kHz 00 0 Figure , ,000 Output Power (W) Output Power (W) CoolMOS P6 Optimum R DS(on) selection charts 950C6 600C6 520C6 Ron optimized at full load Best thermal performance C6 600C6 520C6 Ron optimized at 50% load Balanced light/full load efficiency C6 380C6 280C6 280C6 90C6 60C6 25C6 099C6 070C6 04C kHz 50kHz 50kHz 00kHz 0 90C6 60C6 25C6 099C6 070C6 04C kHz 00kHz 0 Figure , ,000 Output Power (W) Output Power (W) CoolMOS C6 Optimum R DS(on) selection charts Since we found that 45 mω CoolMOS C7 IPW65R045C7 is the optimum device for our design, we can base on it to calculate different power losses at the worst case of 85 Vac full load condition, as follows: The MOSFET RMS current across the 60 Hz line cycle can be calculated by the following equation, and consequently the MOSFET conduction loss can be obtained as: Application Note Revision.,

12 Design Note DN I S.rms = P o 8 2 V ac.min 200 W = V ac.min 3 π V o 85 V V = 2.2 A 3 π 400V P S.cond = I S.rms 2 R on(00 ) = ( ) = 2 W Eq. 6 Eq. 7 (Assuming R on(00 ) =.8 R on(25 ) ) For switching losses calculation, the average input current is used to estimate the losses over the line cycle. The calculation is based on the switching time consideration, where the triangular area between current and voltage changing references to the switching losses. Figure 3 Simplified turn-on and turn-off waveforms The average input current is given as: I L.avg = P o 2 2 V ac.min π = = 200 W 85 V 2 2 π = 2.7 A Eq. 8 Turn-on time and loss are: t on = C iss R g ln ( V g V th V g V pl ) + C rss R g ( V ds V pl V g V pl ) = V 3.5 V F.8Ω ln ( 2V 5.4 V ) V 5.4 V 0 2 F.8Ω ( 2V 5.4 V ) = s Eq. 9 (V ds = V o = 400 V, C rss = Q gd V ds = nc 400V = F) P S.on = 0.5 I L.avg V o t on f = A 400V s Hz = 2.5 W Eq. 20 Turn-off time and loss are: t off = C rss R g ( V ds V pl V pl ) + C iss R g ln ( V pl V th ) = V 5.4V C.8Ω ( ) C.8Ω ln ( 5.4V 5.4V 3.5V ) = s Eq. 2 P S.off = 0.5 I L.avg V o t off f = A 400V s Hz = 3.4 W Eq. 22 The above is the classic format for calculating turn-off time and loss; due to the high Q oss of Super Junction MOSFETs, the C oss acts like a nonlinear capacitive snubber, and actual turn-off losses with fast switching can be up to 50% lower than calculated. The current flow through the drain during turn-off under these Application Note 2 Revision.,

13 Design Note DN conditions is non-dissipative capacitive current, and with fast drive, the channel may be completely turned off by the onset of drain voltage rise. Output capacitance C oss switching loss are: P S.oss = E oss f = Joule Hz =.7 W Eq. 23 Gate drive loss is defined as: P S.gate = V g Q g f = 2 V nc Hz = 0. W Eq. 24 Total MOSFET loss is defined as: P S.total = P S.cond + P S.on + P S.off + P S.oss = 9.2 W Eq TO pin package with Kelvin source connection In common gate drive arrangements, the fast current transient causes a voltage drop V LS across the parasitic inductance of the source of the MOSFET that can counteracts the driving voltage. The induced source voltage, V LS = L*di/dt, can reduce the gate current (Figure 4), therefore lead to slowing down the switching transient and increasing the associated energy loss. On the other hand, the kelvin-source package concept is to exclude the package source inductance and layout inductance from the driving loop, so that the L*di/dt induced voltage is outside the gate drive loop and not affecting the gate current and switching losses. The 4pin MOSFET recommended for this design would be IPZ65R045C7. Figure 4 a) Conventional package b) TO-247 4pin package 2.4 Boost Diode Selection of the boost diode is a major design decision in CCM boost converter. Since the diode is hard commutated at a high current, and the reverse recovery can cause significant power loss, noise and current spikes. Reverse recovery can be a bottle neck for high switching frequency and high power density power supplies. Additionally, at low line, the available diode conduction duty cycle is quite low, and the forward current quite high in proportion to the average current. For that reason, the first criteria for selecting a diode in CCM boost are fast recovery with low reverse recovery charge, followed by V f operating at high forward current. Since Silicon Carbide (SiC) Schottky diodes have capacitive charge,qc, rather than reverse recovery charge, Q rr. Their switching loss and recovery time are much lower compared to Silicon Ultrafast diode, and will show an enhanced performance. Moreover, SiC diodes allow higher switching frequency designs, hence, higher power density converters is achieved. Application Note 3 Revision.,

14 Design Note DN The capacitive charge for SiC diodes are not only low, but also independent on di/dt, current level, and temperature; which is different from Si diodes that have strong dependency on these conditions, as shown in Figure 5. Figure 5 Capacitive charge as a function of di/dt for Si pin double diode and SiC diode The newer generations of SiC diodes are not just Schottky devices, but are merged structure diodes known as MPS diodes - Merged PN/Schottky (Figure 6). They combine the relatively low V f and capacitive charge characteristics of Schottky diodes with the high peak current capability of PN diodes, while avoiding the high junction voltage penalty (typically V at room temperature) of a pure PN wide bandgap diode. Figure 6 Schottky and Merged PN/Schottky compared The recommended diode for CCM boost applications is the 650V thinq! SiC Schottky Diode Generation 5, which include Infineon s leading edge technologies, such as diffusion soldering process and wafer thinning technology. The result is a new family of products showing improved efficiency over all load conditions, coming from both the improved thermal characteristics and an improved Figure-of-Merit (Q c x V f). With the high surge current capability of the MPS diode, there is some latitude for the selection of the boost diode. A simple rule of thumb that works well for a wide input range PFC is A diode rating for each 50 W of output power for good cost/performance tradeoffs, or A diode rating for each 75 W for a premium performance. For example, a 600 W application will only need a 4 A rated diode, but an 8 A diode would perform better at full load. Especially at low-line operation, where the input current is quite high with a Application Note 4 Revision.,

15 Design Note DN short duty cycle, the higher rated diode will have a much lower V f at the actual operating current, reducing the conduction losses. Note that even when using the MPS type SiC diode, it is still preferred to use a bulk pre-charge diode as shown earlier in Figure 3. This is a low frequency standard diode with high I2t rating to support pre-charging the bulk capacitor to the peak of the AC line voltage; this is a high initial surge current stress (which should be limited by a series NTC) that is best avoided for the HF boost rectifier diode. In this design example, we are using the A/75 W rule, so for a 200 W we require a 6 A diode, therefore SiC diode IDH6G65C5 is selected. The boost diode average current and conduction loss are: I D.avg = P o 200 W = V o 400 V = 3 A Eq. 26 P D.cond = I D.avg V f.diode = 3 A.5 V = 4.5 W Eq. 27 Diode switching loss, which is carried by the boost MOSFET is: P D.swit = 0.5 V o Q C f = V nc Hz = 0.46 W Eq. 28 Diode total loss is: P D.total = P D.cond + P D.swit = 4.5 W W = 4.96 W Eq Output Capacitor The output capacitor is sized to meet both of the hold-up time (6.6ms) and the low frequency voltage ripple (0 V) requirements. The capacitor value is selected to have the larger value among the two equations in below: C o 2 P o t hold V 2 2 o V = 2 200W sec o.min (400V) 2 (340V) 2 = μf Eq. 30 C o P o 200W = 2 π f line V o V o 2 π 60Hz 0V 400V = μf Eq. 3 C o = max (900.9 μf, μf) = μf In this design we use two parallel 560 μf, 450 V, with dissipation factor DF=0.2, consequently the capacitor ESR loss is obtained as below: ESR = DF 0.2 = 2 π f C o 2 π 20Hz (2 560 μf) = Ω Eq. 32 The capacitor RMS current across the 60Hz line cycle can be calculated by the following equation. I Co.rms = 8 2 P o 2 P 2 o 3 π V ac.min V 2 o V = 8 2 (200 W)2 o 3 π 85V 400 V (200 W)2 = 6.47 A (400 V) 2 Eq. 33 P Co = I Co.rms 2 ESR = (6.47A) Ω = 9.9 W Eq. 34 Application Note 5 Revision.,

16 Design Note DN Heat sink design The MOSFET and boost diode share the same heat sink, thermal resistors are modeled as in Figure 7. In this evaluation board the maximum heat sink temperature T S is regulated to 60 C, so we can calculate the average junction temperature for the MOSFET and diode as follows: T J.diode = T S + P diode (R thcs.diode + R thjc.diode ) Eq. 35 T J.FET = T S + P FET (R thcs.fet + R thjc.fet ) Eq. 36 The T S can be regulated by choosing a heatsink that with certain airflow can reach the thermal resistance (R thsa ) calculated below: R thsa = Where: T S T A P FET + P diode Eq. 37 R thjc : Thermal resistance from junction to case, this is specified in the MOSFET and Diode datasheets. R thcs : Thermal resistace from case to heatsink, typically low compared to the overall thermal resistance, its value depends on the the interface material, for example, thermal grease and thermal pad. R thsa : Thermal resistance from heatsink to ambient, this is specified in the heatsink datasheets, it depends on the heatsink size and design, and is a function of the surroundings, for example, a heat sink could have difference values for R thsa for different airflow conditions. T S : Heatsink temperature. T C : Case temperature. T A : Ambient temperature. P FET : FET total power loss. P diode : Diode total power loss. P FET R thjc.fet R thcs.fet T J.FET T C.FET T S R thsa T A P diode T J.diode R thjc.diode T C.diode R thcs.diode P FET +P diode Figure 7 Schematic of thermal network Application Note 6 Revision.,

17 Frequency/kHz Design Note DN ICE3PCS0G PFC Boost Controller The ICE3PCS0G is a 4pins controller IC for power factor correction converters. It is suitable for wide range line input applications from 85 to 265 Vac and with overall efficiency above 97%. The IC supports the converters in boost topology and operates in continuous conduction mode (CCM) with average current control. The IC operates with a cascaded control; the inner current loop and the outer voltage loop. The inner current loop of the IC controls the sinusoidal profile for the average input current. It uses the dependency of the PWM duty cycle on the line input voltage to determine the corresponding input current. This means the average input current follows the input voltage as long as the device operates in CCM. Under light load condition, depending on the choke inductance, the system may enter into discontinuous conduction mode (DCM) resulting in a higher harmonics but still meeting the Class D requirement of IEC The outer voltage loop of the IC controls the output bulk voltage and is integrated digitally within the IC. Depending on the load condition, internal PI compensation output is converted to an appropriate DC voltage which controls the amplitude of the average input current. The IC is equipped with various protection features to ensure safe operating for the system and the device. 3. Soft Startup During power up when the V OUT is less than 96% of the rated level, internal voltage loop output increases from initial voltage under the soft-start control. This results in a controlled linear increase of the input current from 0 A as can be seen in Figure 7. This helps to reduce the current stress in power components. Once V OUT has reached 96% of the rated level, the soft-start control is released to achieve good regulation and dynamic response and VB_OK pin outputs 5V indicating PFC output voltage in normal range. 3.2 Gate Switching Frequency The switching frequency of the PFC converter can be set with an external resistor R FREQ at pin FREQ with reference to pin SGND. The voltage at pin FREQ is typical V. The corresponding capacitor for the oscillator is integrated in the device and the R FREQ/frequency is given in Figure 8. The recommended operating frequency range is from 2 khz to 250 khz. As an example, a R FREQ of 43 kω at pin FREQ will set a switching frequency f SW of 00 khz typically. Frequency vs Resistance Resistance /kohm Frequency /khz Resistance /kohm Frequency /khz Resistance/kohm Figure 8 Frequency setting Application Note 7 Revision.,

18 Design Note DN Protection Features 3.3. Open loop protection (OLP) The open loop protection is available for this IC to safe-guard the output. Whenever voltage at pin VSENSE falls below 0.5 V, or equivalently V OUT falls below 20% of its rated value, it indicates an open loop condition (i.e. VSENSE pin not connected). In this case, most of the blocks within the IC will be shutdown. It is implemented using a comparator with a threshold of 0.5 V First over-voltage protection (OVP) Whenever V OUT exceeds the rated value by 8%, the first over-voltage protection OVP is active. This is implemented by sensing the voltage at pin VSENSE with respect to a reference voltage of 2.7 V. A VSENSE voltage higher than 2.7 V will immediately block the gate signal. After bulk voltage falls below the rated value, gate drive resumes switching again Peak current limit The IC provides a cycle by cycle peak current limitation (PCL). It is active when the voltage at pin ISENSE reaches -0.2 V. This voltage is amplified by a factor of -5 and connected to comparator with a reference voltage of.0 V. A deglitcher with 200 ns after the comparator improves noise immunity to the activation of this protection. In other words, the current sense resistor should be designed lower than -0.2 V PCL for normal operation IC supply under voltage lockout When V CC voltage is below the under voltage lockout threshold V CC,UVLO, typical V, IC is off and the gate drive is internally pull low to maintain the off state. The current consumption is down to.4 ma only Bulk Voltage Monitor and Enable Function (VBTHL_EN) The IC monitors the bulk voltage status through VSENSE pin and output a TTL signal to enable PWM IC or control inrush relay. During soft-start once the bulk voltage is higher than 95% rated value, pin VB_OK outputs a high level. The threshold to trigger the low level is decided by the pin VBTHL voltage adjustable externally. When pin VBTHL is pulled down externally lower than 0.5V most function blocks are turned off and the IC enters into standby mode for low power consumption. When the disable signal is released the IC recovers by soft-start. Application Note 8 Revision.,

19 Design Note DN Efficiency and Power Losses Modeling The design example discussed in this document was modeled in Mathcad. All of the power losses equations were written as a function of the output power in order to be able to plot an estimated efficiency curve across the output power range as shown in Figure 9. Figure 9 Calculated 85Vac vs Experimental 90Vac Figure 20 shows a breakdown of main power losses at the full load of both low and high line conditions. Output capacitor loss 3% 85Vac 00% load Total power loss = 74.W MOSFET losses 26% Output capacitor loss 9% 230Vac 00% load Total power loss = 25.3W MOSFET losses 6% Rectifier bridge loss 34% Inductor losses 20% Boost diode losses 7% Rectifier bridge loss 37% Inductor losses 8% Boost diode losses 20% Figure 20 Breakdown of main power losses Figure 2 shows the simulated and experimental inductor current at the low line full load condition. Figure 2 Simulated and experimental Inductor current waveforms at low-line full load condition Application Note 9 Revision.,

20 Design Note DN Experimental Results Figure 22 Evaluation board All test conditions are based on 60 C heat sink temperature. 5. Load and Line measurement data The test data of the evaluation board under different test conditions are listed in Table 2. The corresponding efficiency waveforms are shown in Figure 23 and Figure 24. Table 2 Input Efficiency test data with IPZ65R045C7 & 00kHz, Rg=.8ohms VIN [V] IIN [A] PIN [W] UOUT [V] IOUT [A] POUT [W] Efficiency [%] Power Factor 90Vac Vac Application Note 20 Revision.,

21 Design Note DN Figure 23 High-line efficiency with IPZ65R045C7 & 00kHz, R g=.8 Ω Figure 24 Low-line efficiency with IPZ65R045C7 & 00kHz, R g=.8 Ω 5.2 Conductive EMI Test Compliance with EN55022 standard is a very important quality factor for a power supply. The EMI has to consider the whole SMPS and is split into radiated and conductive EMI consideration. For the described evaluation PFC board it is most important to investigate on the conducted EMI-behavior since it is the input stage of any SMPS below a certain power range, as shown in Figure 25. Application Note 2 Revision.,

22 Design Note DN Figure 25 Conductive EMI measurement of the board with resistive load 5.3 Startup behavior During power up when the VOUT is less than 96% of the rated level, internal voltage loop output increases from initial voltage under the soft-start control. This results in a controlled linear increase of the input current from 0A thus reducing the current stress in the power components as can be seen on the yellow waveform in Figure 26. Figure 26 Waveform capture during low-line startup Application Note 22 Revision.,

23 Design Note DN Board Design 6. Schematics Figure 27 Evaluation board schematic Application Note 23 Revision.,

24 Design Note DN PCB Layout Figure 28 PCB top layer view Figure 29 PCB bottom layer view Application Note 24 Revision.,

Infineon Component Librar y Installation Guide Keysight ADS Installation Guide About this document Infineon Technologies AG provides its component libraries as Design Kits for the ADS simulation environment.

### About this document. 32-bit Microcontroller Series for Industrial Applications AP32301. Application Note

XMC4000 32-bit Microcontroller Series for Industrial Applications D igital to Analog Converter (DAC) About this document Scope and purpose This document describes the features of the DAC peripheral and

650V Rapid Diode for Industrial Applications Application Note About this document Scope and purpose This document introduces the Rapid Diodes, high voltage hyperfast silicon diodes from Infineon. Based

### Scope and purpose This document provides hints for using the RGB LED Lighting Shield to drive and control LED strip lights.

XMC1000 32-bit Microcontroller Series for Industrial Applications D riving LED Strips with the RGB LED Lighting Shield Application Note About this document Scope and purpose This document provides hints

### TLE7368 Pre-regulator Filters Dimensioning

Multi Voltage Power Supply System Application Note Rev. 1.01, 2015-09-22 Automotive Power Table of Contents Table of Contents................................................................. 2 1 Abstract.........................................................................

### Power Management & Multimarket

TVS Diodes Transient Voltage Suppressor Diodes TVS3V3L4U Low Capacitance ESD / Transient / Surge Protection Array TVS3V3L4U Data Sheet Revision 2.4, 213-2-6 Final Power Management & Multimarket Edition

### SLE 66R01L Intelligent 512 bit EEPROM with Contactless Interface compliant to ISO/IEC 14443 Type A and support of NFC Forum Type 2 Tag Operation

Intelligent 512 bit EEPROM with Contactless Interface compliant to ISO/IEC 14443 Type A and support of NFC Forum Type 2 Tag Operation Short Product Information 2010-05-10 Chip Card & Security Edition 2010-05-10

### Angle Sensors. Magnetic Design Tool. Application Note. Sense & Control. xmr-based Angular Sensors TLE5009(D) TLE5011 TLE5012B(D) TLE5309D TLE5014

Angle Sensors xmr-based Angular Sensors Magnetic Design Tool TLE5009(D) TLE5011 TLE5012B(D) TLE5309D TLE5014 Application Note Ver. 1.0, 2014-12-03 Sense & Control Edition 2014-12-03 Published by Infineon

### Anti-Tampering Solution for E-Meter Application

TLV493D Application Note Revision 1.0 2015-07-24 Sense & Control Table of Contents 1 Introduction..................................................................... 3 2.................................................................

### Power Management and Multimarket

LED Driver ICs for High Power LEDs ILD65 Data Sheet Revision 3.2, 24-7-9 Power Management and Multimarket Edition 24-7-9 Published by Infineon Technologies AG 8726 Munich, Germany 24 Infineon Technologies

### Power Management & Multimarket

Protection Devices TVS (Transient Voltage Suppressor Diodes) ESD18-B1-CSP21 Bi-directional, 5.5 V,.28 pf, 21, RoHS and Halogen Free compliant ESD18-B1-CSP21 Data Sheet Revision 1.4, 216-4-21 Final Power

### LED Drivers for High Power LEDs

LED Drivers for High Power LEDs ILD435 Data Sheet Revision 2., 211-8-17 Industrial and Multimarket Edition 211-8-17 Published by Infineon Technologies AG 81726 Munich, Germany 211 Infineon Technologies

### Power Management & Multimarket

TVS Diodes Transient Voltage Suppressor Diodes ESD112-B1-02 Series Bi-directional Ultra-low Capacitance ESD / Transient Protection Diode ESD112-B1-02ELS ESD112-B1-02EL Data Sheet Rev. 1.3, 2013-11-27 Final

### Power Management & Multimarket

Protection Device TVS (Transient Voltage Suppressor) ESD231-B1-W0201 Bi-directional, 5.5 V, 3.5 pf, 0201, RoHS and Halogen Free compliant ESD231-B1-W0201 Data Sheet Revision 1.1, 2015-10-02 Final Power

### Microcontroller Series for Industrial Applications

XMC4000 Microcontroller Series for Industrial Applications Power Management Bus ( PMBus) Host with XMC4000 Application Guide V1.1 2014-03 Microcontrollers Edition 2014-03 Published by Infineon Technologies

### LED Driver for High Power LEDs ILD4001. Data Sheet. Industrial and Multimarket. Step down LED Controller for high power LEDs. Revision 2.

LED Driver for High Power LEDs ILD4001 Data Sheet Revision 2.0, 2011-06-09 Industrial and Multimarket Edition 2011-06-09 Published by Infineon Technologies AG 81726 Munich, Germany 2011 Infineon Technologies

### AURIX, TriCore, XC2000, XE166, XC800 Families DAP Connector

AURIX, TriCore, XC2000, XE166, XC800 Families DAP Connector Application Note V1.4, 2014-05 Microcontrollers Edition 2014-05 Published by Infineon Technologies AG 81726 Munich, Germany 2014 Infineon Technologies

### White LED Card. inlight_whiteled_v4. Board User's Manual. Microcontroller. For XMC1000 Family. White LED Card. Revision 1.

For XMC1000 Family inlight_whiteled_v4 Board User's Manual Revision 1.0, 2013-03-08 Microcontroller Edition 2013-03-08 Published by Infineon Technologies AG 81726 Munich, Germany 2013 Infineon Technologies

### OptiMOS 3 Power-Transistor

Type IPD36N4L G OptiMOS 3 Power-Transistor Features Fast switching MOSFET for SMPS Optimized technology for DC/DC converters Qualified according to JEDEC ) for target applications Product Summary V DS

### Magnetic Circuit Design for Correct Direction Detection of TLE4966V

Magnetic Circuit Design for Correct Direction Detection of TLE4966V Application Note Rev. 1.0, 2013-10-28 Sense & Control Edition 2013-10-28 Published by Infineon Technologies AG 81726 Munich, Germany

### High Precision Hall Effect Latch for Consumer Applications

High Precision Hall Effect Latch for Consumer Applications Hall Effect Latch TLV4961-1T TLV4961-1TA TLV4961-1TB TLV4961-1T Data Sheet Revision 1.0, 2015-05-19 Sense & Control Table of Contents 1 Product

### BGM1032N7. Data Sheet. RF & Protection Devices. GPS and GLONASS Front-End Module. Revision 3.2, 2011-07-18

Data Sheet Revision 3.2, 2011-07-18 RF & Protection Devices Edition 2011-07-18 Published by Infineon Technologies AG 81726 Munich, Germany 2011 Infineon Technologies AG All Rights Reserved. Legal Disclaimer

### 7-41 POWER FACTOR CORRECTION

POWER FTOR CORRECTION INTRODUCTION Modern electronic equipment can create noise that will cause problems with other equipment on the same supply system. To reduce system disturbances it is therefore essential

### For XMC1000 Family CPU-13A-V1. XMC1300 CPU Card. Board User's Manual. Revision 2.0, 2013-12-18. Microcontroller

For XMC1000 Family CPU-13A-V1 Board User's Manual Revision 2.0, 2013-12-18 Microcontroller Edition 2013-12-18 Published by Infineon Technologies AG 81726 Munich, Germany 2013 Infineon Technologies AG All

### Evaluation Kit. 3D Magnetic Sensor 2 Go Kit. User s Manual. for 3D Magnetic Sensor TLV493D-A1B6. Rev. 1.1 2016-01. Sense & Control

for 3D Magnetic Sensor TLV493D-A1B6 3D Magnetic Sensor 2 Go Kit User s Manual Rev. 1.1 2016-01 Sense & Control Table of Contents 1 Introduction...................................................................

### Colour LED Card. inlight_rgb_v3. Board User's Manual. Microcontroller. For XMC1000 Family. Colour LED Card. Revision 1.

For XMC1000 Family inlight_rgb_v3 Board User's Manual Revision 1.0, 2013-03-08 Microcontroller Edition 2013-03-08 Published by Infineon Technologies AG 81726 Munich, Germany 2013 Infineon Technologies

### OptiMOS TM 3 Power-Transistor

BSC7N1NS3 G OptiMOS TM 3 Power-Transistor Features Very low gate charge for high frequency applications Optimized for dc-dc conversion N-channel, normal level Excellent gate charge x R DS(on) product (FOM)

### Power Management & Multimarket

TVS Diodes Transient Voltage Suppressor Diodes ESD15-B1-2 Series Low Capacitance & Low Clamping Bi-directional ESD / Transient Protection Diodes ESD15-B1-2ELS ESD15-B1-2EL Data Sheet Revision 1., 213-12-12

### OptiMOS TM 3 Power-Transistor

BSC47N8NS3 G OptiMOS TM 3 Power-Transistor Features Ideal for high frequency switching and sync. rec. Optimized technology for DC/DC converters Excellent gate charge x R DS(on) product (FOM) Product Summary

### OptiMOS 3 Power-Transistor

Type IPD6N3L G OptiMOS 3 Power-Transistor Features Fast switching MOSFET for SMPS Optimized technology for DC/DC converters Qualified according to JEDEC 1) for target applications Product Summary V DS

### TriCore AURIX Family. PCB design Guidelines. Application Note. 32-bit (TC23x, TC22x) AP32261 V1.2 2015-10

TriCore AURIX Family 32-bit (TC23x, TC22x) PCB design Guidelines Application Note V1.2 2015-10 Microcontrollers Edition 2015-10 Published by Infineon Technologies AG, 81726 Munich, Germany. 2015 Infineon

### BGA725L6. Data Sheet. RF & Protection Devices

Silicon Germanium Low Noise Amplifier for Global Navigation Satellite Systems (GNSS) in ultra small package with 0.77mm² footprint Data Sheet Revision 2.0, 2012-03-09 Preliminary RF & Protection Devices

### OptiMOS Power-Transistor Product Summary

OptiMOS Power-Transistor Product Summary V DS 55 V R DS(on),max 4) 35 mω Features Dual N-channel Logic Level - Enhancement mode AEC Q11 qualified I D 2 A PG-TDSON-8-4 MSL1 up to 26 C peak reflow 175 C

### Power Management & Multimarket

TVS Diodes Transient Voltage Suppressor Diodes ESD0P2RF Series Bi-directional Ultra-low Capacitance ESD / Transient Protection Diode ESD0P2RF-02LS ESD0P2RF-02LRH Data Sheet Rev. 1.2, 2012-10-01 Final Power

### Application Note, Rev.1.0, September 2008 TLE8366. Application Information. Automotive Power

Application Note, Rev.1.0, September 2008 TLE8366 Automotive Power Table of Contents 1 Abstract...3 2 Introduction...3 3 Dimensioning the Output and Input Filter...4 3.1 Theory...4 3.2 Output Filter Capacitor(s)

### OptiMOS TM Power-Transistor

Type BSC28N6NS OptiMOS TM Power-Transistor Features Optimized for high performance SMPS, e.g. sync. rec. % avalanche tested Superior thermal resistance N-channel Qualified according to JEDEC ) for target

### BFR843EL3. Data Sheet. RF & Protection Devices. Robust Low Noise Broadband Pre-Matched Bipolar RF Transistor. Revision 1.

Robust Low Noise Broadband Pre-Matched Bipolar RF Transistor Data Sheet Revision 1., 214-8-5 RF & Protection Devices Edition 214-8-5 Published by Infineon Technologies AG 81726 Munich, Germany 214 Infineon

### Power Management & Multimarket

TVS Diodes Transient Voltage Suppressor Diodes ESD3V3U4ULC Ultra-low Capacitance ESD / Transient Protection Array ESD3V3U4ULC Data Sheet Rev. 1.6, 2013-02-20 Final Power Management & Multimarket Edition

### 1ED Compact A new high performance, cost efficient, high voltage gate driver IC family

1ED Compact A new high performance, cost efficient, high voltage gate driver IC family Heiko Rettinger, Infineon Technologies AG, Am Campeon 1-12, 85579 Neubiberg, Germany, heiko.rettinger@infineon.com

### BFP740ESD. Data Sheet. RF & Protection Devices. Robust Low Noise Silicon Germanium Bipolar RF Transistor. Revision 1.1, 2012-10-08

Robust Low Noise Silicon Germanium Bipolar RF Transistor Data Sheet Revision 1.1, 2012-10-08 RF & Protection Devices Edition 2012-10-08 Published by Infineon Technologies AG 81726 Munich, Germany 2013

### O p t i m u m M O S F E T S e l e c t i o n f o r S y n c h r o n o u s R e c t i f i c a t i o n

V2.4. May 2012 O p t i m u m M O S F E T S e l e c t i o n f o r S y n c h r o n o u s R e c t i f i c a t i o n IFAT PMM APS SE DS Mößlacher Christian Guillemant Olivier Edition 2011-02-02 Published by

### AC/DC Power Supply Reference Design. Advanced SMPS Applications using the dspic DSC SMPS Family

AC/DC Power Supply Reference Design Advanced SMPS Applications using the dspic DSC SMPS Family dspic30f SMPS Family Excellent for Digital Power Conversion Internal hi-res PWM Internal high speed ADC Internal

### EiceDRIVER. High voltage gate driver IC. Application Note. Revision 1.0, 2013-07-26

EiceDRIVER High voltage gate driver IC Evaluation Board Application Note EVAL_1ED020I12-B2 Application Note Revision 1.0, 2013-07-26 Infineon Technologies AG Edition 2013-07-26 Published by Infineon Technologies

### N-Channel 20-V (D-S) 175 C MOSFET

N-Channel -V (D-S) 75 C MOSFET SUD7N-4P PRODUCT SUMMARY V DS (V) r DS(on) ( ) (A) a.37 @ V GS = V 37.6 @ V GS = 4.5 V 9 TO-5 D FEATURES TrenchFET Power MOSFET 75 C Junction Temperature PWM Optimized for

### SLE 66R01P SLE 66R01PN SLE 66R16P SLE 66R32P SLE 66R01L

How to operate my-d devices in NFC Forum Type 2 Tag infrastructures SLE 66R01P SLE 66R01PN SLE 66R16P SLE 66R32P SLE 66R01L Application Note 2011-11-14 Chip Card & Security Edition 2011-11-14 Published

### TLE4961-4M. Data Sheet. Sense & Control. High Precision Automotive Hall Effect Latch. Revision 1.0,

High Precision Automotive Hall Effect Latch Data Sheet Revision 1.0, 2013-04-07 Sense & Control Edition 2013-04-07 Published by Infineon Technologies AG 81726 Munich, Germany 2013 Infineon Technologies

### For XMC4000 Family. Standard Human Machine Interface Card. Board User s Manual. Revision 1.0, 2012-02-28

Hexagon Application Kit For XMC4000 Family HMI_OLED-V1 Board User s Manual Revision 1.0, 2012-02-28 Microcontroller Edition 2012-02-28 Published by Infineon Technologies AG 81726 Munich, Germany 2012 Infineon

### CoolMOS TM Power Transistor

CoolMOS TM Power Transistor Features New revolutionary high voltage technology Intrinsic fast-recovery body diode Extremely low reverse recovery charge Ultra low gate charge Extreme dv /dt rated Product

### OptiMOS -3 Small-Signal-Transistor

BSS66N OptiMOS -3 Small-Signal-Transistor Features N-channel Enhancement mode Logic level (.5V rated) Avalanche rated Product Summary V DS 6 V R DS(on),max V GS = V 6 mw V GS =.5 V 9 I D 3. A Qualified

### Test Methods for DC/DC Power Modules

Test Methods for DC/DC Power Modules Design Note 027 Ericsson Power Modules Precautions Abstract A user may have the desire to verify or characterize the performance of DC/DC power modules outside the

### Revision: Rev. 1.0 2012-06-08

High-Gain Low Noise Amplifier for Global Navigation Satellite Systems (GNSS) from 1550 MHz to 1615 MHz Applications using 0201 components Application Note AN280 Revision: Rev. 1.0 RF and Protection Devices

### Performance Evaluation of Cree SiC Schottky Diode in a Non-isolated LED Light Bulb Application

Performance Evaluation of Cree SiC Schottky Diode in a Non-isolated LED Light Bulb Application Jimmy Liu, Kin Lap Wong Cree Inc Abstract: As the demand for low-cost energy efficient LED lighting grows,

### SIPMOS Small-Signal-Transistor

SIPMOS Small-Signal-Transistor Features N-channel Depletion mode dv /dt rated Product Summary V DS V R DS(on),max 3.5 Ω I DSS,min.4 A Available with V GS(th) indicator on reel Pb-free lead plating; RoHS

### ULRASONIC GENERATOR POWER CIRCUITRY. Will it fit on PC board

ULRASONIC GENERATOR POWER CIRCUITRY Will it fit on PC board MAJOR COMPONENTS HIGH POWER FACTOR RECTIFIER RECTIFIES POWER LINE RAIL SUPPLY SETS VOLTAGE AMPLITUDE INVERTER INVERTS RAIL VOLTAGE FILTER FILTERS

### RGB Wall Washer Using ILD4035

ILD4035 Application Note AN216 Revision: 1.0 Date: RF and Protection Devices Edition Published by Infineon Technologies AG 81726 Munich, Germany 2010 Infineon Technologies AG All Rights Reserved. Legal

### Power MOSFET FEATURES. IRF520PbF SiHF520-E3 IRF520 SiHF520. PARAMETER SYMBOL LIMIT UNIT Drain-Source Voltage V DS 100 V Gate-Source Voltage V GS ± 20

Power MOSFET PRODUCT SUMMARY (V) 100 R DS(on) ( ) = 0.7 Q g (Max.) (nc) 16 Q gs (nc) 4.4 Q gd (nc) 7.7 Configuration Single TO0AB G DS ORDERING INFORMATION Package Lead (Pb)free SnPb G D S NChannel MOSFET

### AEROSEMI MT3608 GENERAL DESCRIPTION FEATURES APPLICATIONS TYPICAL APPLICATION. High Efficiency 1.2MHz 2A Step Up Converter.

FEATURES AEROSEMI Integrated 80mΩ Power MOSFET 2V to 24V Input Voltage 1.2MHz Fixed Switching Frequency Internal 4A Switch Current Limit Adjustable Output Voltage Internal Compensation Up to 28V Output

### Design of an Auxiliary Power Distribution Network for an Electric Vehicle

Design of an Auxiliary Power Distribution Network for an Electric Vehicle William Chen, Simon Round and Richard Duke Department of Electrical & Computer Engineering University of Canterbury, Christchurch,

### UNISONIC TECHNOLOGIES CO., LTD

UPS61 UNISONIC TECHNOLOGIES CO., LTD HIGH PERFORMANCE CURRENT MODE POWER SWITCH DESCRIPTION The UTC UPS61 is designed to provide several special enhancements to satisfy the needs, for example, Power-Saving

### SEMICONDUCTOR FC Pin Switch-Mode LED Lamp Driver IC FC9921 TECHNICAL DATA

SEMICONDUCTOR TECHNICAL DATA FC99 3-Pin Switch-Mode LED Lamp Driver IC Features Constant output current: 0mA Universal 85-65VAC operation Fixed off-time buck converter Internal 475V power MOSFET Applications

### Features. Symbol JEDEC TO-220AB

Data Sheet June 1999 File Number 2253.2 3A, 5V,.4 Ohm, N-Channel Power MOSFET This is an N-Channel enhancement mode silicon gate power field effect transistor designed for applications such as switching

### Power MOSFET. IRF9520PbF SiHF9520-E3 IRF9520 SiHF9520. PARAMETER SYMBOL LIMIT UNIT Drain-Source Voltage V DS - 100 V Gate-Source Voltage V GS ± 20

Power MOSFET PRODUCT SUMMARY (V) 100 R DS(on) ( ) = 10 V 0.60 Q g (Max.) (nc) 18 Q gs (nc) 3.0 Q gd (nc) 9.0 Configuration Single TO220AB G DS ORDERING INFORMATION Package Lead (Pb)free SnPb G S D PChannel

### Power MOSFET. IRF510PbF SiHF510-E3 IRF510 SiHF510. PARAMETER SYMBOL LIMIT UNIT Drain-Source Voltage V DS 100 V Gate-Source Voltage V GS ± 20

Power MOSFET PRODUCT SUMMARY (V) 100 R DS(on) () = 0.54 Q g max. (nc) 8.3 Q gs (nc) 2.3 Q gd (nc) 3.8 Configuration Single D TO220AB G FEATURES Dynamic dv/dt rating Available Repetitive avalanche rated

### Preliminary Datasheet

Features Macroblock Preliminary Datasheet 1.2A Constant Output Current 93% Efficiency @ input voltage 13V, 350mA, 9~36V Input Voltage Range Hysteretic PFM Improves Efficiency at Light Loads Settable Output

### Power MOSFET FEATURES. IRF610PbF SiHF610-E3 IRF610 SiHF610. PARAMETER SYMBOL LIMIT UNIT Drain-Source Voltage V DS 200 V Gate-Source Voltage V GS ± 20

Power MOSFET PRODUCT SUMMARY (V) 00 R DS(on) ( ) = 1.5 Q g (Max.) (nc) 8. Q gs (nc) 1.8 Q gd (nc) 4.5 Configuration Single FEATURES Dynamic dv/dt Rating Repetitive Avalanche Rated Fast Switching Ease of

### DISCONTINUED. Product Marking Reel size (inches) Tape width (mm) Quantity per reel DMN4009LK3-13 N4009L ,500

40V N-CHANNEL ENHANCEMENT MODE MOSFET Product Summary V (BR)DSS 40V R DS(on) T A = 25 C 8.5mΩ @ = V 27.6A 4mΩ @ = 4.5V 2.5A Description and Applications This new generation MOSFET has been designed to

### Application Note AN-1070

Application Note AN-1070 Class D Audio Amplifier Performance Relationship to MOSFET Parameters By Jorge Cerezo, International Rectifier Table of Contents Page Abstract... 2 Introduction... 2 Key MOSFET

### Final data. Maximum Ratings Parameter Symbol Value Unit

SPPN8C3 SPN8C3 Cool MOS Power Transistor V DS 8 V Feature R DS(on).45 Ω New revolutionary high voltage technology Ultra low gate charge I D Periodic avalanche rated Extreme dv/dt rated Ultra low effective

### Revision: Rev

Driver Amplifier for LTE Band-3 (1805-1880 MHz) Applications Application Note AN390 Revision: Rev. 1.0 RF and Protection Devices Application Note AN390 Revision History: Previous Revision: Page Subjects

### BGS12AL7-6. Data Sheet. RF & Protection Devices. SPDT RF Switch. Revision 2.1,

Data Sheet Revision 2.1, 2009-12-09 RF & Protection Devices Edition 2009-12-09 Published by Infineon Technologies AG 81726 Munich, Germany 2009 Infineon Technologies AG All Rights Reserved. Legal Disclaimer

### 32-bit Microcontroller Series for Industrial Applications AP32294. Application Note

XMC1302 32-bit Microcontroller Series for Industrial Applications Server Fan Control Reference Design Application Note About this document This document is designed for the low voltage server fan motor

### N-Channel 60-V (D-S), 175 C MOSFET

N-Channel 6-V (D-S), 75 C MOSFET SUP/SUB7N6-4 V (BR)DSS (V) r DS(on) ( ) (A) 6.4 7 a TO-22AB D TO-263 DRAIN connected to TAB G G D S Top View SUP7N6-4 G D S Top View SUB7N6-4 S N-Channel MOSFET Parameter

### Motor Control Shield with BTN8982TA for Arduino

Motor Control Shield for Arduino Motor Control Shield with BTN8982TA for Arduino About this document Scope and purpose This document describes how to use the Motor Control Shield with BTN8982TA for Arduino.

### AN1489 Application note

Application note VIPower: non isolated power supply using VIPer20 with secondary regulation Introduction Output voltage regulation with adjustable feedback compensation loop is very simple when a VIPer

### Power MOSFET FEATURES. IRF740PbF SiHF740-E3 IRF740 SiHF740. PARAMETER SYMBOL LIMIT UNIT Drain-Source Voltage V DS 400 V Gate-Source Voltage V GS ± 20

Power MOSFET PRODUCT SUMMARY (V) 400 R DS(on) (Ω) = 0.55 Q g (Max.) (nc) 63 Q gs (nc) 9.0 Q gd (nc) 3 Configuration Single FEATURES Dynamic dv/dt Rating Repetitive Avalanche Rated Fast Switching Ease of

### Power MOSFET FEATURES. IRF740APbF SiHF740A-E3 IRF740A SiHF740A

Power MOSFET PRODUCT SUMMARY V DS (V) 400 R DS(on) ( ) = 0 V 0.55 Q g (Max.) (nc) 36 Q gs (nc) 9.9 Q gd (nc) 6 Configuration Single D TO220AB G FEATURES Low Gate Charge Q g Results in Simple Drive Requirement

### PMPB15XN. 1. Product profile. 20 V, single N-channel Trench MOSFET 13 September 2012 Product data sheet. 1.1 General description

3 September 22 Product data sheet. Product profile. General description N-channel enhancement mode Field-Effect Transistor (FET) in a leadless medium power DFN22MD-6 (SOT22) Surface-Mounted Device (SMD)

### UNISONIC TECHNOLOGIES CO., LTD 50N06 Power MOSFET

UNISONIC TECHNOLOGIES CO., LTD 50N06 50 Amps, 60 Volts N-CHANNEL POWER MOSFET DESCRIPTION TO-263 TO-25 The UTC 50N06 is three-terminal silicon device with current conduction capability of about 50A, fast

### Design Considerations for an LLC Resonant Converter

Design Considerations for an LLC Resonant Converter Hangseok Choi Power Conversion Team www.fairchildsemi.com 1. Introduction Growing demand for higher power density and low profile in power converter

### LDS8720. 184 WLED Matrix Driver with Boost Converter FEATURES APPLICATION DESCRIPTION TYPICAL APPLICATION CIRCUIT

184 WLED Matrix Driver with Boost Converter FEATURES High efficiency boost converter with the input voltage range from 2.7 to 5.5 V No external Schottky Required (Internal synchronous rectifier) 250 mv

### Features G D. TO-220 FQP Series

FQP33N10 FQP33N10 100V N-Channel MOSFET April 2000 QFET TM General Description These N-Channel enhancement mode power field effect transistors are produced using Fairchild s proprietary, planar stripe,

### BGB707L7ESD. Data Sheet. RF & Protection Devices. Wideband MMIC LNA with Integrated ESD Protection. Revision 3.3, 2012-11-09

Wideband MMIC LNA with Integrated ESD Protection Data Sheet Revision 3.3, 2012-11-09 RF & Protection Devices Edition 2012-11-09 Published by Infineon Technologies AG 81726 Munich, Germany 2013 Infineon

### DISCONTINUED. Product Marking Reel size (inches) Tape width (mm) Quantity per reel DMN4015LK3-13 N4015L ,500

40V N-CHANNEL ENHANCEMENT MODE MOSFET Product Summary V (BR)DSS 40V T A = 25 C 5mΩ @ = V 20.8A 20mΩ @ = 4.5V 8.0A Description and Applications This new generation MOSFET has been designed to minimize the

### Power MOSFET FEATURES. IRFZ44PbF SiHFZ44-E3 IRFZ44 SiHFZ44 T C = 25 C

Power MOSFET PRODUCT SUMMARY (V) 60 R DS(on) (Ω) V GS = 10 V 0.028 Q g (Max.) (nc) 67 Q gs (nc) 18 Q gd (nc) 25 Configuration Single FEATURES Dynamic dv/dt Rating 175 C Operating Temperature Fast Switching

### 6.6 mm x 4.5 mm x 0.6 mm

High-Performance DrBLADE 6.6 mm x 4.5 mm x 0.6 mm TDA21320 Data Sheet Revision 2.4, 2015-07-16 Power Management and Multi Market Edition 2015-07-1626 Published by Infineon Technologies AG 81726 Munich,

### Chapter 4. LLC Resonant Converter

Chapter 4 LLC Resonant Converter 4.1 Introduction In previous chapters, the trends and technical challenges for front end DC/DC converter were discussed. High power density, high efficiency and high power

### BFP840FESD for GHz WLAN

High Gain Low Noise Amplifier using BFP840FESD for 2.4-2.5 GHz WLAN Application Application Note AN320 Revision: Rev. 1.0 RF and Protection Devices Edition Published by Infineon Technologies AG 81726 Munich,

### BUZ11. 30A, 50V, 0.040 Ohm, N-Channel Power MOSFET. Features. [ /Title (BUZ1 1) /Subject. (30A, 50V, 0.040 Ohm, N- Channel. Ordering Information

Data Sheet June 1999 File Number 2253.2 [ /Title (BUZ1 1) /Subject (3A, 5V,.4 Ohm, N- Channel Power MOS- FET) /Autho r () /Keywords (Intersil Corporation, N- Channel Power MOS- FET, TO- 22AB ) /Creator

### AP4511GM Pb Free Plating Product Advanced Power N AND P-CHANNEL ENHANCEMENT Electronics Corp.

AP5GM Pb Free Plating Product Advanced Power N AND P-CHANNEL ENHANCEMENT Electronics Corp. MODE POWER MOSFET Simple Drive Requirement N-CH BV DSS 35V Low On-resistance D D D R DS(ON) 5mΩ D D D D D Fast

### PAM2841. Description. Pin Assignments. Features. Applications. Typical Applications Circuit. A Product Line of. Diodes Incorporated

1.5A SW CURRENT, 40V PRECISION WLED DRIVER Description Pin Assignments The is a white LED driver, capable of driving 10 or more WLEDs in series (depending on forward voltage of the LEDs) with a range of

### Power MOSFET FEATURES. IRF9640PbF SiHF9640-E3 IRF9640 SiHF9640

Power MOSFET PRODUCT SUMMARY V DS (V) 200 R DS(on) (Ω) = 10 V 0.50 Q g (Max.) (nc) 44 Q gs (nc) 7.1 Q gd (nc) 27 Configuration Single TO220AB G DS ORDERING INFORMATION Package Lead (Pb)free SnPb G S D

### Power MOSFET FEATURES. IRF740PbF SiHF740-E3 IRF740 SiHF740. PARAMETER SYMBOL LIMIT UNIT Drain-Source Voltage V DS 400 V Gate-Source Voltage V GS ± 20

Power MOSFET PRODUCT SUMMARY (V) 400 R DS(on) (Ω) = 0.55 Q g (Max.) (nc) 63 Q gs (nc) 9.0 Q gd (nc) 3 Configuration Single FEATURES Dynamic dv/dt Rating Repetitive Avalanche Rated Fast Switching Ease of

### Exercise 6-1. The Power MOSFET EXERCISE OBJECTIVES

Exercise 6-1 The Power MOSFET EXERCISE OBJECTIVES At the completion of this exercise, you will know the behaviour of the power MOSFET during switching operation. You will be able to explain how MOSFET

### Power MOSFET FEATURES. IRL540PbF SiHL540-E3 IRL540 SiHL540

Power MOSFET PRODUCT SUMMARY (V) 100 R DS(on) (Ω) = 5.0 V 0.077 Q g (Max.) (nc) 64 Q gs (nc) 9.4 Q gd (nc) 27 Configuration Single TO220AB G DS ORDERING INFORMATION Package Lead (Pb)free SnPb G D S NChannel

### (Single E n d C ap T8) C onverter with I C L8201

A N-REF-ICL8201_Single E nd C a p T8 18W 270mA Single Stage Floating B u ck L E D (Single E n d C ap T8) C onverter with I C L8201 & IPS65R1K5CE About this document Scope and purpose This document is a

### AND8147/D. An Innovative Approach to Achieving Single Stage PFC and Step-Down Conversion for Distributive Systems APPLICATION NOTE

An Innovative Approach to Achieving Single Stage PFC and Step-Down Conversion for Distributive Systems APPLICATION NOTE INTRODUCTION In most modern PFC circuits, to lower the input current harmonics and

### Power MOSFET FEATURES. IRF540PbF SiHF540-E3 IRF540 SiHF540. PARAMETER SYMBOL LIMIT UNIT Drain-Source Voltage V DS 100 V Gate-Source Voltage V GS ± 20

Power MOSFET PRODUCT SUMMARY (V) 100 R DS(on) ( ) = 0.077 Q g (Max.) (nc) 72 Q gs (nc) 11 Q gd (nc) 32 Configuration Single TO220AB G DS ORDERING INFORMATION Package Lead (Pb)free SnPb G D S NChannel MOSFET

### SPW32N50C3. Cool MOS Power Transistor V DS @ T jmax 560 V

SPW3N5C3 Cool MOS Power Transistor V DS @ T jmax 56 V Feature New revolutionary high voltage technology Ultra low gate charge Periodic avalanche rated Extreme dv/dt rated Ultra low effective capacitances