1KW CLASS-E 13.56MHz SINGLE DEVICE RF GENERATOR for INDUSTRIAL APPLICATIONS

Transcription

1 DIRECTED ENERGY, INC. APPLICATION NOTE KW CLASS-E 3.56MHz SINGLE DEVICE RF GENERATOR for INDUSTRIAL APPLICATIONS George J. Krausse, Directed Energy, Inc. Abstract There are a large variety of industrial processes that require reliable, low cost, regulated RF power. RF generators are well suited to these applications because of their high efficiency, high reliability and low system cost. The DE-SERIES POWER MOSFETs are ideally suited to RF generator applications. Fast switching speed combined with low parasitic capacitances, low on resistance and low thermal resistance allows them to dramatically outperform all competing MOSFET devices. This application note discusses the theory of operation and circuit design of the Class-E generator using DEI DE-SERIES MOSFETS.. An Directed Energy, Inc. IXYS Company 40 Research Blvd. Suite 08 Fort Collins, CO USA 8056 TEL (970) FAX (970) deiinfo@directedenergy.com Directed Energy, Inc. All Rights Reserved

2 TABLE OF CONTENTS INTRODUCTION... THE CLASS E RF GENERATOR... 3 Theory of Operation... 3 Calculating Class-E Element Values... 4 SPICE MODEL... 5 PROTOTYPE CIRCUIT... 6 FPS-4N Gate Drive... 7 PROTOTYPE CIRCUIT PERFORMANCE AND SPICE... 9 POWER MANAGEMENT... Power Measurement... Control... Protection... CONCLUSION... 3 REFERENCES... 4 APPENDIX... 5 Spice Model...5 RF Data...6 Page

3 INTRODUCTION There are a large variety of industrial processes that require reliable, low cost, regulated RF power. Applications include RF plasma processing of silicon and gallium arsenide wafers, induction heating, glass and lens coating, plastic forming and industrial laser power supplies, to name a few. In the past this high power RF was provided by RF amplifiers. More recently the RF generator has displaced the RF amplifier in this field. The RF generator is a non-linear system in which the RF power output is produced at very high efficiency. The efficiency in a typical kw RF generator can reach 90%. This is accomplished by operating the power device as a saturated switch. This is considerably different from the linear power amplifier, where the power device is operated in the linear mode, and the efficiency can reach 65%. These two parameters, mode of operation and system efficiency, are the key distinctions between RF power amplifiers and RF power generators. The RF generator is well suited to industrial applications because of its high efficiency. This high efficiency brings with it the added benefit of fewer components, therefore potentially higher reliability and lowered system cost (,, 3). The DE-SERIES POWER MOSFETs are ideally suited to RF generator applications. Fast switching speed combined with low parasitic capacitances, low on resistance and low thermal resistance allows them to dramatically outperform all competing MOSFET devices. (4) In this application note we will discuss the theory of operation and circuit design of the Class-E generator using DE-SERIES MOSFETs. A SPICE circuit model and device model will be developed to aid in the design phase, and the results will be bench tested and those results will be compared to the SPICE model output. The SPICE model will then be adjusted to reflect the performance of the prototype circuit. We will also explore circuit protection as well as power measurement and control. Page

4 THE CLASS E RF GENERATOR Theory of Operation A typical Class E power amplifier is a single-ended switch-mode topology, one active device and an output series resonant, tuned load principals network as shown in Figure. +Vin C.5UF 500V Q DE-375 0N0A FPS-4N Gate Drive 3 L uh L 00nH C4 90pF KV C6 77pF 3.5KV RLOAD.5 Figure Class E circuit The active device should have high speed turn on and off characteristics, low on resistance and low C OSS and C RSS, such that it can be an effective low loss switch in its saturated mode of operation. The resonant load network is designed so that its transient response reduces the power dissipation in the active device during the switching intervals. T0 T T T3 Vds Ids Vgs Figure Class E Waveforms Referring to Figure, in an ideal Class E circuit, during the off state of the active device, the current drain remains at zero while the voltage across the device, Vds, increases to a maximum of 3.5 (Duty Cycle=50%) times Vcc (T 0 -T ). At the end of the off cycle (T ), the voltage across the active device has decreased to zero. At (T ) Vgs is applied and the current through the active device increases toward a maximum of approximately.86 times Idc. At the end of the on state (T ) the gate drive, Vgs is removed and the current drops to zero before the voltage begins to rise. In principle, Page 3

5 there is no appreciable current flowing while drain voltage is present across the device and likewise there is no appreciable voltage across the device while drain current is flowing through the device. During switching transitions, both current and voltage have zero crossover values. With switching losses reduced in the manner just described, the only loss remaining is conduction. The ideal efficiency in a high power Class E amplifier can approach 90%. Increased efficiency not only means lower input power to the amplifier, but more importantly less heat dissipation in the active device. In Figure, the resonant load network consists of four passive elements C shunt, C series, L series, and the effective load resistance R. The values of these four elements are chosen such that the resonant frequency and Q produce the ideal waveforms shown in Figure. L RF CHOKE, shown in Figure, is essentially a high impedance. Its value should be sufficiently high so as to act as a constant current source to the resonant circuit. The value of the effective load resistance R is a function of the desired RF output power and the applied DC voltage. The Q of the resonant circuit is dependent on the following factors: ) the relative importance of the harmonic frequency delivered to the effective load resistance, and ) the transient response of the voltage and current waveforms across the active device. If the Q is too low, the voltage across the active device does not discharge to zero prior to the device turning on. Too high of a Q and the voltage across the device discharges too quickly and possibly even swings negative. Calculating Class-E Element Values Given that the desired RF output power, the frequency of operation and the DC power supply voltage are know, then assuming a value for the loaded Q of the resonant load network, we can calculate the values of the Class E resonant elements. The derivation of these equations is beyond the scope of this paper and the reader should refer to the references for more details regarding the origin of the equations. We at DEI select the value of R as follows: In order to maximize efficiency we let: R = RDS (ON) 0 The three component values in the resonant output network can now be calculated as follows. The series inductor L series, can be found from: R L series = Q π F The equation for the series capacitor C series is given as: + C Q series = π F Q R The shunt capacitor across the active switch C shunt is found as: Cshunt = Cseries Q 6.3 Page 4

6 The calculated value of C shunt is the total value of the capacitance across the active device, which means the value of C OSS of the active device should be subtracted from the value calculated for C shunt. The value of the output capacitance of the active device is a voltage dependent variable, and it is sometimes difficult to choose an exact value to subtract from C shunt without the aid of SPICE or in some cases empirically. All of the values calculated above are for ideal components, and do not take into consideration the parasitic effects, the unloaded Q of the individual components, or the network layout. These effects will either increase or decrease the calculated values depending on whether the parasitic effects are in series or parallel with the calculated element. From the two equations above for calculating the capacitors C series and C shunt, it can be seen that both equations are a function of Q. The designer may decide to vary the chosen value of Q a few tenths higher of lower to achieve standard capacitor values. Let: F=3.56Mhz Q=7 R=.5Ω Using the equations above and these three terms yields the following: L SERIES =030nH C SERIES =53pF C SHUNT =70pF For the purposes of this article, we will use the DE-375 0N0A MOSFET. The device specifications are available on the DEI web site, The power MOSFET SPICE model for the DE-375-0N0A includes all strays, capacitive and inductive, and their voltage dependencies. SPICE MODEL Figure 3 SPICE Circuit Model The Class-E SPICE model is shown in Figure 3. In the model, DC power is supplied by the voltage source V PULSE, and the DC current is monitored at I(V). R4 represents Page 5

7 the resistive loss in the conductor of L. Here L is sized so that at 3.56Mhz the X L is about 000Ω. The gate drive includes the source impedance of the FPS-4N gate driver and its loop inductance term. The current monitor I(V3) gives us the drain current. By returning the gate drive image currents as shown from V Pulse to the top of I(V3) we can eliminate their effect on the drain current waveform and reduce the busy nature of the display caused by this superimposition of the gate drive transients. First approximation values based on the preceding equations for all devices were installed in the SPICE model and prototype test circuit. The performance of both the test circuit and the SPICE model were evaluated. Each component was then optimized manually and the performance reevaluated. This process was repeated until the model included the appropriate component values and the correct strays. PROTOTYPE CIRCUIT +Vin Q DE-375 0N0A FPS-4N Gate Drive 3 C.5UF 500V L uh L 00nH C4 90pF KV.5 Ohms C6 77pF 3.5KV FERRITE CORE ea 5 Ohm Coax 5 Ohm Coax RLOAD 50 V= to Z= to 4 N=6 Figure 4 Prototype Test Circuit Schematic Figure 4 is a simplified schematic diagram of the test circuit. The.5Ω Class-E load resistance is provided by the :4 transmission line transformer shown. In all other respects the circuit of Figure 4 is identical with that of Figure. Page 6

8 FPS-4N Gate Drive The gate driver for this article is the FPS-4N ( see Figure 5). The FPS-4N is a square wave driver with extremely low output impedance (typically 0.5Ω), and series L term of about nh. The driver turn-on and off times are in the 3-5ns range with 3nF loads. In addition, the driver is capable of pulse width and frequency agility, from a minimum of about 0ns to DC. The maximum frequency is about 30MHz. This allows the gate drive to have less than 50% duty, which is key in achieving 90% efficiency. The driver and the design techniques used in its development are covered in great detail in the DEI gate driver design manual (5). Gerber files are also available at no charge for DE- SERIES users. Figure 5 FPS-4N Gate Driver Page 7

9 Figure 6 Prototype Circuit The Prototype Circuit is shown in Figure 6. All support and high voltage power has been fed through common mode chokes (CMC), to reduce ground loops and improve the quality of the data. These CMC s are the pot-cores on the left side of the figure. The FPS-4N gate driver is shown in the center of the photograph. Just below the FPS-4N is the 3.56Mhz oscillator and TTL drive circuit. Above center is the 4uH RF choke and the 0.5uF by-pass capacitors. The Tektronix P500 HV probe is in the lower center. The shunt capacitor, C4, is just to the left of the Tek probe tip. Further to the right is the series inductor and to the right of the inductor is the series capacitor. Just to the left of the BNC connector is the :4 transmission line transformer. The RF output at the right is applied to a Bird model attenuator through a Bird model 43 wattmeter via approximately meter of 50Ω coaxial cable. The oscilloscope used for this application note was the Tektronix TDS MHz digital oscilloscope. The high voltage power supply is an Electronics Measurements Inc. model EMS SMPS with voltage and current control. This type of high voltage supply is extremely well suited for power circuit development. The current control aspect allows the user to set a current limit thus offering some level of protection for the circuit under development. Page 8

10 PROTOTYPE CIRCUIT PERFORMANCE AND SPICE Performance data for the optimized test circuit follows. Figure 7 Class E Gate Drive Waveform In Figure 7, we see the gate drive waveform. The +V rise to +V fall pulse width is 30ns and the peak value is approximately 0V. The frequency is 3.56Mhz. The loop inductance and ESR of the FPS-4N gate driver are extremely low. This, combined with the low gate lead inductance of the DE-SERIES devices, allow accurate views of the gate drive signal with or without HV power. Figure 8 SPICE Model Gate Drive Comparing Figure 7 to Figure 8 above we see that, with the exception of a less pronounced mid-value notch on the falling edge and ns in width, the two waveforms are in agreement. Page 9

11 Figure 9 Class E V ds Waveform In Figure 9, we see the V ds waveform peaking at 808V. If we refer to Figure we see that the drain waveform is very close to the classical ideal in that there is not a substantial voltage across the switch at commutation. In fact the drain voltage at turn on is 0% of the HV supply and 5% of the drain peak. This near zero drain voltage allows the switch Q to operate at near the theoretical maximum efficiency. The low drain lead inductance of the DE-SERIES and high frequency, high impedance voltage probes allow the accurate capture of the drain waveform. Figure 0 SPICE Model V DS Drain Waveform Comparing the 808V drain voltage peak of Figure 9 to Figure 0, we see that the spice model drain voltage peak is 740V. This yields a 9% error. Page 0

12 Figure RF Spectrum Of The Prototype Circuit Figure illustrates the RF Spectrum of the prototype circuit into a 50Ω load. The second harmonic is 35db down from the fundamental and the third harmonic is about 53db down. The power output was measured at 000W, using a Bird model 43 wattmeter. In Table, we show several parameters for both the SPICE model and the prototype circuit. Reconciling the errors there are several factors that we must keep in mind. The preceding calculations, which describe the three key component values for Class-E operation, are based on a 50% duty factor. The duty factor in this article is 46%. In addition, RF measurements are rarely better than ±3% whereas the DC measurements when free from RF interference can be very precise. And the purpose of the SPICE model is to assist in the design and development phase, as well as to provide insight into the circuit performance, not provide exact results. Table PARAMETER SPICE MODEL PROTOTYPE CIRCUIT SPICE to PROTO ERROR V DS PK 740V 808V -9% V S 90V 90V Set point I S % P O % EFF% 88.4% 88.6% +.% For full data set see Appendix. At this point, we see a reasonable correlation between the SPICE model and the prototype test circuit. The DE-SERIES MOSFETS, (4) allow the shunt capacitors to be installed at the drain lead close enough to minimize the effect of lead inductance to the point of being inconsequential. In Figure 7, we see that there are 4 capacitors used to achieve the correct value of capacitance for C4. By placing capacitors of equal value on both sides of the drain PCB pad and attaching the other ends, on each side, to one of the two source pads, we provide two opposed and balanced current loops thus invoking Electro- Page

13 Magnetic Symmetry. This, along with the two opposed and balanced current loops of the DE-SERIES device, give us a total of four significant and distinct, opposed and balanced current loops. This action reduces the net loop inductance substantially. POWER MANAGEMENT The load in industrial applications, unlike a 50Ω application, is often extremely dynamic, ranging in impedance from a fraction of an Ohm to several hundred Ohms. The problem is further compounded by the fact that often the process requires the generator to operate with large mismatches for at least a short period while the load or the load matching circuits stabilize. There is the additional requirement of very accurate power control. A high precision of absolute RF power is far less important than an extraordinarily high accuracy of reproducibility. We can explore these conditions in the SPICE model, then verify in the prototype circuit. Power Measurement The directional coupler is the device most often used to measure RF power. When the load is purely resistive and well matched, their accuracy and reproducibility are very good. However, this is often not the case with industrial type loads. These loads can often have highly reactive time, voltage, current or power varying components, which can produce extreme variations in system load impedance. This implies that until the load is matched, the power readings from a directional coupler will be inaccurate. Construction and performance information for several designs are found in the Radio Handbook (6) and the ARRL Handbook (7). An alternate method for power measurement is the direct measurement and multiplication of the RF current and voltage waveforms. However, this requires very accurate and stable current and voltage probes. It also requires an accurate high frequency multiplier. The Analog Devices AD834 is specified at < ±.db in flatness error (7). Control Power control in a single Class-E generator is accomplished in two ways. The first method and the most common is via the DC supply. We can vary this supply from a few volts to maximum and in so doing vary the RF power in the same way. The second and less common method is to vary the pulse width of the gate drive. Again, the output power will vary accordingly. However, the efficiency of the generator can be very adversely affected. Protection Using the SPICE model we can change the R LOAD from.5ω to.5ω then to 5Ω which corresponds to 50Ω, 5Ω and 500Ω then observe which parameters change and by how much. Page

14 TABLE R LOAD Vds pk Ids pk Pout Pin Ploss Eff % REMARKS.5Ω 93V 8.7A 87W 333W 46W 56% Low.5Ω 808V 6.4A 003W 34W 3W 88.4% Normal 5Ω 636V 3A 359W 6W 757W 3% High In Table we see the effects of load dynamics for a load change of a factor of ±0. In the case of the High, the Ploss parameter is problematic, and at 757W into a device with only 340W dissipation capability, the device will be destroyed in a few milliseconds. Therefore we must provide some immediate protection. In the Low circuit condition there again is only one parameter that is a problem, the Vds peak. At 93V, the device is too close to the V DS maximum. We changed the load on the prototype circuit to approximate Table. On power up we see that for both the Low and the High cases, the test circuit operation moves in the direction of the device failures as predicted by SPICE. CONCLUSION From the preceding we see that the Class-E RF generator can produce KW from a single DE SERIES MOSFET device with very high efficiency. Several of these stages can be combined to reach the multi-kilowatt level. The close alignment between the calculated values, the SPICE model and the prototype circuit is extremely helpful for circuit design and evaluation. It is recommended that the designer simulate the chosen active device, and the calculated values along with an estimation of the circuit parasitics, in a SPICE model. This technique allows the designer to understand how varying certain element values can lead to an optimum circuit configuration. Simulation of the circuit in SPICE will also allow the designer to monitor parameters such as the currents in the active device, and each of the network components, which cannot typically be monitored on the real hardware. These parameters can also help determine the dissipation and operating margins in each of the circuit elements. In addition, this allows the designer to vary parameters, load conditions and component strays, via SPICE, in order to synthesize the optimum circuit design for the application prior to spending time at the bench. The overall impact is to enhance the quality of engineering designs while reducing the time to market. The reference list includes two patents involving Class-E operation (, ). Patent # 3,99,656 (), is the original patent. It covers the theory of operation and expired in 993. The second, patent # 5,87,580 (), focuses on a specific sub-optimum mode of operation by design. The circuit designer should become familiar with both texts prior to circuit design and implementation. Page 3

15 REFERENCES. High-Efficiency Tuned Switching Power Amplifier Patent # 3, High Power Switch-Mode Radio Frequency Amplifier Method and Apparatus Patent # 5,87, Solid State Radio Engineering, Krauss, Bostian and Raab, CH 4. Wiley An Introduction to the DE-SERIES MOSFET, George J Krausse, Directed Energy, Inc. (DEI) 5. Gate Driver Design For Switch-mode Applications and the DE-SERIES MOSFET, George J Krausse, Directed Energy, Inc. (DEI) 6. Radio Handbook, 3 Edition SAMS ISBN: , ARRL Handbook Analog Devices Application note AN- (AD834) Page 4

16 APPENDIX Spice Model *SPICE_NET *INCLUDE SMSUB.LIB.SUBCKT 0N0A * TERMINALS: D G S * 000 Volt 0 Amp.0 ohm N-Channel Power MOSFET * NEW RF DIE Low RG and Low Ciss,Crss and Coss * REV.A GJK M 3 3 DMOS L=U W=U RON DON 6 D ROF 5 7. DOF 7 D DCRS 8 D DCRS 8 D CGS 3 3.0N RD 4.5 DCOS 3 D3 RDS 3 5.0MEG LS N LD 0 4 N LG 0 5 N.MODEL DMOS NMOS (LEVEL=3 VTO=3.0 KP=3.8).MODEL D D (IS=.5F CJO=P BV=00 M=.5 VJ=.6 TT=N).MODEL D D (IS=.5F CJO=00P BV=000 M=.3 VJ=.6 TT=400N RS=0M).MODEL D3 D (IS=.5F CJO=400P BV=000 M=.3 VJ=.4 TT=400N RS=0M).ENDS.OPTION METHOD=GEAR RELTOL=0.000 VNTOL=E-3 ABSTOL=E-3.TRAN N 3U.80U 0N.VIEW TRAN V(3) VIEW TRAN V(0) VIEW TRAN V() VIEW TRAN I(V) VIEW TRAN I(V) VIEW TRAN I(V3) *INCLUDE DIODE.LIB *ALIAS V(3)=VGS *ALIAS I(V)=IVS *ALIAS V(0)=RFVOUT *ALIAS V()=VDS *ALIAS I(V)=IGS *ALIAS I(V3)=IDS.PRINT TRAN V(3) I(V) V(0) V().PRINT TRAN I(V) I(V3) R.5 L 3 NH L 6 4.0UH V 5 0 PULSE U.U.U 5U 00U C PF L NH R R R L4 9 0 NH C4 8 90PF R X0 3 0N0A V3 0 DC 0 V PULSE 0 0 0N N N 3N 74N.END Page 5

17 RF Data N0A DE-375 Class-E V DC IN I DC IN P IN P OUT Eff % V DS PK 9V.37A 4.7W 00W V 6V.76A.8W 00W V 54V.0A 33.4W 300W V 80V.44A 439.W 400W V 04V.76A 563.0W 500W V 3V 3.03A 675.7W 600W V 40V 3.6A 78.4W 700W V 57V 3.49A 896.9W 800W V 73V 3.7A 0.8W 900W V 90V 3.89A 8.W 000W V TEXT DATA +Vin Q DE-375 0N0A FPS-4N Gate Drive 3 C.5UF 500V L uh L 00nH C4 90pF KV.5 Ohms C6 77pF 3.5KV FERRITE CORE ea 5 Ohm Coax 5 Ohm Coax RLOAD 50 L =00nH D=.0in. L=.0in. N=7 W=#0ga. V= to Z= to 4 N=6 Page 6