2. INTRODUCTION. 2.1 Problem Statement


 Hannah Simmons
 2 years ago
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1 1. ABSTRACT The main purpose of this project is to design an inverter that will enable the inversion of a DC power source, supplied by Photovoltaic (PV) Cells, to an AC power source that will be either used to supply a load or connected directly to the utility grid. The system will be controlled to operate at maximum efficiency using Maximum Power Point Tracking (MPPT) algorithm. This algorithm will be installed on a controller. The controller will be implemented on an FPGA (Field Programmable Gate Arrays) and designed through a computer using a program called LABVIEW. The programmed FPGA will be able to automatically control the whole power system operation without the need of any user intervention. The benefit of this project is to give access to an everlasting and pollution free source of energy. And give the user the option to use the system in two possible operating modes; the stand alone mode which is used to satisfy his needs, and the grid connected mode which used to sell electricity to utility when in excess; thus eliminating the need of battery storage. 1
2 2. INTRODUCTION 2.1 Problem Statement The world demand for electric energy is constantly increasing, and conventional energy resources are diminishing and are even threatened to be depleted. Moreover; their prices are rising. For these reasons, the need for alternative energy sources has become indispensable, and solar energy in particular has proved to be a very promising alternative because of its availability and pollutionfree nature. Due to the increasing efficiencies and decreasing cost of photovoltaic cells and the improvement of the switching technology used for power conversion, we are interested in developing an inverter powered by PV panels and that could supply standalone AC loads and also be connected to the grid. Solar panels produce direct currents (DC), and to connect these panels to the electricity grid or use them in other industrial applications, we should have an AC output at a certain required voltage level and frequency. The conversion from DC to AC is essentially accomplished by means of a DCAC inverter, which is the major component in the system. Yet, the output of the solar panels is not continuously constant and is related to the instantaneous sunlight intensity and ambient temperature. 2
3 2.2 Objectives The main objective of our project is to design and construct a PV based system that produces electric energy and operates in dual mode, supplying stand alone AC loads or the grid, while minimizing its cost and size. The system s main properties are: Production of quality electricity from a renewable source to reduce dependence on fossil fuels and the associated emissions of pollutants. Reduce cost of energy consumption by being able connect to the grid: sell energy and remove need for storage batteries which are the most expensive system components per watt. Our goal is to design and develop an inverter that will handle the task described and implement the following: Maximum Power Point Tracking (MPPT) algorithm to achieve the maximum power possible from the PV panel. Synchronization with the utility grid. 3
4 3. REVIEW 3.1 Various Design Solutions Several circuit topologies exist that deal with PV connection to the grid or other stand alone systems. In all cases, a panel's DC voltage and current should be transformed into AC with appropriate parameters (amplitude, frequency) so that solar power may be utilized properly. In what follows is a description of the basic system topologies implemented Direct Inverter Connection Solar Panel DC/AC Filter Grid or AC Load Control Figure 1: Direct Inverter Connection As seen from the system above, a straight forward approach is adopted by changing the DC current into AC current directly. Although a simple topology, yet it contains several difficulties and limitations especially in terms of control. Considering MPPT is used, we note that the control unit will have to perform two operations simultaneously: first finding the MPP and producing an output voltage that is constant and sinusoidal. Each operation will affect the other simultaneously, and thus the MPPT system is no longer efficiently utilized, because the output voltage is the primary priority. Usually, because of isolation requirements, a transformer can be introduced if low voltage is available from the panel, but still the main control problem remains: Solar Panel DC/AC Transformer Filter Grid or AC Load Control Figure 2: Direct Inverter Connection with transformer 4
5 3.1.2 DCDC and DCAC Connection Topology In a second and more practical approach, a DCDC converter is used before the DC AC block. The system is shown in the figure below: Solar Panel DC/DC DC/AC Filter Transformer Grid or AC Load Control Figure 3: DCDC and DCAC Connection Topology The incorporation of the DC/DC converter made it possible to control the different parameters in the system: the MPP and voltage amplitude. The converter is controlled to fetch the MPP, and the inverter is controlled (PWM control) to adjust for a certain voltage amplitude and to invert the near DC input to a sinewave output. The switched signal will be filtered by a low frequency tuned filter. A low frequency transformer will then accommodate for the filtered signal and deliver power properly to the utility or an AC load. In such implementation we could benefit most from the MPPT control by splitting the control parts into MPP fetching and amplitude adjustment. Yet the system just mentioned suffers from some practical drawbacks, mainly size and weight disadvantages. The output transformer operating at low frequency will be bulky and heavy. In addition, a simple filter will also incorporate bulky capacitive and inductive elements. These drawbacks can be over passed by using high frequency switching in the DCDC converter followed directly by a high frequency transformer. The topology is presented and further discussed below: Solar Panel DC/DC HF Transformer DC/AC LPF Grid or AC Load Control Figure 4: DCDC and DCAC Connection Topology with HF Transformer 5
6 Such arrangement introduces major advantages in terms of practicality of a system and compatibility. High frequency operation will need a HF transformer which is of small size and very light weight. Thus the switching signal of the DC/DC converter enters the HF transformer, stepped to a higher voltage level, and then rectified and filtered to a DC output. The inverter later accepts the constant high voltage and transforms it with use of PWM control to be filtered by a LPF, thus producing the needed signal DCDC and Unfolding Circuit Topology Another interesting topology makes use of some characteristics of DC rectification. In the system shown below, the panel input is fed into the DCDC converter, that is later supplied to the unfolding circuit, which is actually an HBridge. The trick in this innovative topology which was tested in July 2005 at the Florida Power Electronics Center in UCF (University of Central Florida) is to use an Hbridge and a rectifier in combination with a controller that will yield a DC output that has a rectified sinusoidallike waveform. These half rectified waves are then just unfolded (by using another HBridge) to give a proper sinusoidal output to be used by the grid or load: Solar Panel DC/DC Unfolding Circuit Grid or AC Load Control Figure 5: DCDC and Unfolding Circuit Topology 3.2 Various Design Components The PV Array Fundamentals of PV Operation: The solar panel is the source of energy utilized in our project, and is responsible for transforming solar energy into electric energy. The solar panel is made up of a group of solar modules, which are in turn built up from a combination of parallel and series solar cells connected to each other. These cells are made out of a semiconductor, mostly Silicon. Solar energy is in fact delivered in photon packets that hit these solar cells [4], and are then transformed into electric energy as will be explained shortly. 6
7 If the photons are absorbed, and their energy is equal to the band gap energy, which is the amount of energy needed to free an electron from its outermost shell in a Si atom, electron hole pairs are generated. The latter phenomenon is referred to as the photovoltaic effect. These pairs will produce an electric field and thus drive an electric current to flow. If the energy of the photons is lower than the band gap energy, the photons passively go by, and if the energy is much larger electrons are produced but heating starts to occur in the panel. Electrical Models for PV Arrays: A photovoltaic array can be represented by different models some of which are more simplified equivalents. In [13] a discussion is made on a very simplified model as shown in the circuit below, and is referred to as the single diode model. The current produced by the solar cells are represented by the current source, and an ideal diode represents the intrinsic pn junction of the solar cell: The load current is: I=I ph I D, where I ph is generated by the current source, I D is given by the Schockley equation [5] to be I D = I s ( exp (qv / mkt) 1 ) Therefore, the IV relation is given by the following: Where I and V are the output current and voltage of the array respectively I ph is the photo current I D is the diode current I S is the reversed saturation diode current M is the ideally factor of the diode 7
8 V T is the thermal voltage and is equal to k.t/e (k boltzamn's constant, T is the temperature in Kelvin, and e is the electric charge) In a typical solar array, a group of solar cells are connected together to form the total panel, and thus such connectors have innate resistances. Thus a more accurate model is given by [2], [3], [4], and [13] and includes series and shunt resistances to accommodate for the cells connected in a grid type, and the diode is no more ideal. The figure below represents the model, and the IV relation is derived in a similar manner to the previous model, but taking into account the current fed into the resistors: I Finally, a more precise model is presented in [13] and is called the two diode model. In this model, the second diode accounts for the non ideal characteristics of a diode and a second variable current source is introduced that will account for currents when a high negative voltage is seen by the array, that will lead to reverse biased operation: 8
9 I A comparison is presented in [13], which provides a real IV characteristic of a solar panel, and draws on the same plot the three different characteristics of these models. The figure below shows slight differences among these models, with model 2 and 3 giving very precise representations of the IV characteristic. This indicates that even the simplest model will provide a practically acceptable representation, and produce minimal variations. Figure 6: IV curve of a multicrystalline solar cell (10 x 10 cm), irradiance E=430 W/m², temperature T=300 K. Temperature and Irradiance Dependence of the IV Characteristic: The PV's IV characteristic actually depends on the operating temperature and the level of irradiance. The latter two effects can influence the open circuit voltage (V oc ) and the short circuit current (I sc ), which are important parameters that shape the IV characteristic. The open circuit voltage is the voltage seen at the terminals of the PV panel when no load is connected, and thus zero current flow. The short circuit current is when the panel voltage is zero, and thus presents the maximum current a panel can provide. These two parameters mark the intercept values in an IV characteristic plot. 9
10 Temperature and solar irradiance are capable of changing these two parameters. A change in temperature will affect the open circuit voltage more heavily than short circuit current of the panel, and the opposite is true for insolation changes. The following figures show these effects [4]. Figure 7: VI characteristics with change in insolation and temperature Maximum Power Point Tracking Algorithms The maximum power point tracking algorithm is the basis of the control algorithm used in the project. It will control the duty cycle of the DCDC converter, thus forcing the solar panel to operate at its MPP. A number of algorithms have been developed for this issue, which present a compromise between the complexity of hardware to be used, and the complexity of the algorithm itself. In what follows is a description of some common algorithms used in such type of control. P&O Algorithm Using Adaptive Incremental Step Technique: P&O are perturb and observe algorithms. They work by perturbing the duty cycle in one direction (example: increasing it) and computing the power at that time it the power increased compared to the previous power value then, we continue to vary the duty cycle in the same direction. On the contrary, if a drop is noted in the power, the 10
11 direction of perturbation is reversed. Eventually the algorithm will reach the MPP and will be locked around it until either insolation or temperature varies. In a paper presented in [4], a P&O algorithm using an adaptive scheme to control the duty cycle of the converter is used. The conventional implementation of P&O is used, but the main distinction is found in the control of the step size of the duty cycle. The algorithm's adaptive nature is capable of detecting a monotonic increase or decrease of the duty cycle, and accordingly can modify the step size to lessen the time needed to reach the MPP, and thus perform less iterations. The following flow chart clarifies the algorithm: Start Measure V S (n) and I S (n) No P S (n) > P S (n1)? Count= 0 Δ= Δ/ 2 d(n)= d(n 1) + Δ Count= Count + 1 If Δ < 1 then Δ= 1 No Count > 4? d(n)= d(n 1) Δ Δ= Δ 2 If Δ > 32 then Δ= 32 End The algorithm initiates with a measure of the voltage and current of the panel, and thus computes the power. If the power is greater than the preceding value, the step size is increased by a value, and a counter is incremented. The process repeats, but when the counter is incremented 4 times, the step size is doubled, over and over again (until reaching the preset value =32 in this case). This maneuver will lead to faster detection of the MPP. Just when the measured power is detected to be lower than the preceding value, the counter is set to 0, the step size is halved, and the process 11
12 repeated. The overall process continues to reach a small interval where the duty cycle oscillates in. Experimental results have shown satisfactory results, even under different insolation, temperatures, and seasonal changes, and all with high efficiency conversion rates. The only issue that remains, as with most P&O algorithms, is the constant oscillation of the duty cycle when it tends to lock on the MPP. Also there is a limitation when multiple solar arrays are available and are operating at different power point because of different insolation (local clouds). In this case the total power curve will be the superposition of all the power curves corresponding to each PV array. This curve might have multiple local maxima and the P&O algorithm might get locked on a maxima that is local but non global. A MPP tracking algorithm based on I mpp =f(p mpp ): A MPP tracking algorithm based on a relation between I mpp and P mpp is discussed in [1]. This relation is the central issue in implementing the algorithm. The method suggests finding the equation relating I mpp and P mpp, either experimentally, or analytically. This relation is in fact a representation of all the points on the locus connecting the peak points in the IP plane. Thus by modeling of the PV for various irradiance levels, the peak points can be obtained. Then by curve fitting these points, the locus is determined. The suggested algorithm is tested using a specific solar panel and uses curve fitting to produce a second order polynomial to relate I mpp and P mpp, in the form of I mpp =f(p mpp ). The fit has been generated using a computer model of the PV array under simulated irradiance levels. When the locus is produced, and relation fitted to a polynomial, tracking is achieved by a continuous monitoring of average power. The value of the power is substituted into the equation, the value of I is computed. This current acts as a reference current, that will be used to force the PV current to match it. This is accomplished with a PI controller that monitors the error. Thus the control goes into a cycle of changing current and accordingly a change in power. The steady state error is reached when the operating point is optimum, and thus it denotes I mpp and P mpp. 12
13 The load used consisted of a 1 ohm resistor and a 1 mh inductor. Two different simulations were carried out in the paper to assess the effectiveness of this algorithm. A rapid insolation change test and a load change test. It proved to be robust against quick irradiance and load changes, and showed a very fast transition of the duty cycle from a level to another, and does not encounter oscillations as in the case of P&O algorithms. Yet the main drawback is its dependency on temperature effects that has not been taken into account into the relation, which is the heart of operation. Thus for a changing operating temperature, the algorithm fetches a point not along the appropriate locus of peak points, resulting in decreasing efficiency. The mentioned equation must be therefore multiplied by a correlation factor to try to accommodate for these changes. Algorithm Based on One Variable Measurement for MPP Tracking: In a paper presented by [2], another algorithm is presented that finds the MPP of a solar panel. The main premise of this algorithm is its need to measure only one variable to detect the MPP, although it operates as an ordinary P&O algorithm (description of these class of algorithm is in the next section). The paper proceeds by finding a relation between P mpp and I that can be used to identify the maximum power point. Again, reference is made to the PV model to reach such relation. Thus for the PV model including the series and shunt resistances we have: I pv =I L I o (exp(v pv +I pv R s /mv t ) 1) (V pv +I pv R s )/R sh And for a constant output voltage after the buck DC converter we have: V o = (t on /T) V pv thus Pin =V pv I pv = V o I pv /D = V o P buck * Where P buck * = I pv /D It can be shown that if P* is plotted versus D, it will have the same MPP as plotting the input power of the converter versus D. Therefore, the paper concludes with a relation between P* and I, thus making it possible to determine P mpp only from a measurement of the PV current. We now have to measure the current only, compute the power according to the relation, and then change the duty cycle and measure the 13
14 power again. This process is repeated as in usual P&O algorithms to find P mpp as shown in the flow diagram below. The duty cycle is changed in constant steps. The experimental results show a satisfactory operation of the algorithm, especially in its capability to quickly find the MPP in rapidly changing atmospheric conditions. It led to a notable increase in the efficiency of the overall system, and it is attractive because of its simple algorithm implementation and fewer hardware that are usually used in normal P&O algorithms that are also needed to measures the voltage. 14
15 A Modified Tracking Algorithm Incorporating Insolation and Temperature Loops: This algorithm is in fact a modification of P&O algorithms, by an essential incorporation of an insolation and temperature loop. The main theme, presented by C. Hua and J. Lin [3], is related to the dependency of current and voltage on changes in temperature in current and voltage respectively. This fact is true in PV arrays, since a change in temperature will affect current more heavily than voltage changes, and the opposite is true for insolation changes. The algorithm is shown in the following flow chart succeeded by an interpretation of its operation. As can be seen, a main part of the algorithm is an implementation of a normal P&O algorithm, where power is calculated, and depending on the preceding and successive values, the reference voltage controlling the duty cycle is changed. Yet, the important part of the algorithm is seen in before going into the normal P&O algorithm and by a specification of separate insolation and temperature loops. After measuring the voltage and current of the panel, the voltage is discarded temporarily, and a comparison of a previous value of I and its present value is done. Thus, by this measurement, if the current changes more than a preset small value then, we know that there has been a major temperature change that affected the current and 15
16 thus we directly change the reference voltage depending on the sign of the comparison done. Else, if this is not true, we enter into the insolation loop then we have to determine if the insolation changed or not and thus we measure the power using the saved value of V. If we encounter a change in power from two different instances we signal a change in insolation, and have to determine whether the voltage has increased or not, also based on the increase or decrease of power. The main advantage in this algorithm is that it is able to perform faster when it detects a temperature change, thus saving two steps that will be usually encountered in a P&O algorithm not taking into account the separation of insolation and temperature effects. Experimental results shown in the paper proved a satisfactory operation of the proposed algorithm, even under rapid atmospheric conditions, yet at steady state, and as with most P&O algorithms, we will have an oscillating reference voltage that moves in a small interval, and thus an oscillating duty cycle. Look Up Tables: Look up table techniques are quite fast techniques that are able to directly accommodate the duty cycle to the appropriate value [4]. In such techniques, for a known characteristic of a PV array, we have to store a duty cycle value for each insolation and temperature. Thus by measuring the latter two quantities, a duty cycle is directly associated with them. These techniques consume a lot of memory and do not serve as real searching algorithms, although the results are satisfactory. Incremental Conductance Technique: In this technique, the main theme is concerned with the PV curve of the solar panel. At the MPP, the peak of the curve, we have dp/dv =0 [5]. This technique makes use of this fact to base the changes of the duty cycle. 16
17 Since P=VI dp/dv = I +VdI/dV When the latter should equal zero we have: di/dv = I/V The derivative in this algorithm is actually accomplished by taking the difference of the two successive values of current and voltage, and the assumption is particularly true for high sampling rates. Thus whenever the derivative is equal to the negative of the ratio of the present current and voltage, we know that we are at the peak of the power curve. The overall efficiency of the algorithm is high, and operates well under varying atmospheric conditions, yet the implementation is more costly than other techniques DCDC Converters DCDC converters are devices used to change an DC input voltage from one level to another DC level at the output. These devices are also referred to as switch mode converters, because of their utilization of power switching techniques to accomplish the mentioned tasks. A buck converter is a DCDC converter that is used to decrease the input voltage, while a boost converter is intended to increase the input voltage. A buckboost converter is a combination of these and can be utilized to operate in both modes. Converters come up in different types and topologies that depend on the nature of the application. In addition, DCDC converters could be either isolated or nonisolated. In isolated converters, power is transferred from the input side to the output side by means of a transformer, whereas nonisolated types connect the input and output by a common current path [4]. Control of DCDC Converters: Powerswitching is the basis of operation of all DCDC converters. The simple circuit shown below with the ideal switch and the following output waveform clarifies this concept: 17
18 For a certain switching frequency fs, i.e. period Ts, the output voltage would be an equivalent intermittent form of the input voltage. The output voltage is given to be the average of the alternating waveform. As can be seen, if Ton tends to Ts, i.e. the switch is always on; the output voltage is equal to the input voltage. Switching control is mostly done using Pulse Width Modulation switching or PWM control [6]. In PWM switching a saw tooth waveform is compared to a reference voltage. Whenever the reference voltage is greater in amplitude than the saw tooth amplitude, the switch turns on, and it turns off when this voltage is lower. The following figure depicts PWM operation, The duty cycle, which represents the percent of time the switch is on during the whole switching period, is given by: D=t on /Ts 18
19 The Buck Converter: The Buck converter is used to change the input DC source voltage to a lower value of output DC voltage, and is also referred to as the step down converter. Its circuit is shown below: Figure 8: The Buck Converter When the switch is on, the input voltage causes the diode to be reversed biased and all input voltage appears across it. During this interval, the inductor experiences a voltage V L across it and starts to store energy, and thus the output voltage V o = V in V L, yet in the off state, V L = Vo The following waveforms are experienced across the inductor in both switching states: As can be seen, the average value of the inductor value is zero in the switching period, yet I L goes through a series of fluctuations in both states while the inductor is charging and discharging its energy. Since in the on and off states, the output current equals the 19
20 inductor current less than the capacitor current, and since the average capacitor current is also zero, then the average value of the inductor current is equal to the output current of the converter. Since the inductor waveform voltages are equal, we have: (VinVo)ton = Vo(t off ) =Vo (Tst on ) We have Vo/Vin = t on /Ts = D and since Pin=Pout, we get Io/Iin = 1/D The mentioned equation thus shows the buck operation of this converter In the explanation just outlined, the converter was operating in a continuous conduction mode, i.e. the inductor current was well above the zero level and is therefore never zero. Nevertheless, other operation modes may be implemented which are operation at the boundary of continuous and discontinuous modes and a discontinuous mode of operation. Both approaches are depicted in the figures below respectively: In both modes of operation, the same analysis as that of continuous operation follows. While operating at the boundary between these modes, the average inductor current is equal half the peak of the inductor current, and this will in turn be equal to the output current. In the discontinuous operation mode, a similar approach is made, but the time 20
21 in which the current is zero should be taken into account. Considering the voltage waveforms across the inductor in the discontinuous operation figure above, we can see that, and since the average inductor voltage is zero: (Vin Vo)DTs = (Vo)Ts 1 which implies that Vo/Vin = D/(D+ 1 ) The average output current in this case is similar to the one seen in the boundary conduction mode, but now we average this current over D+ 1 [4], and thus we have: Io=I L peak (D+ 1 )/2 Finally, an important concept that will be revisited later is concerned with the output voltage and current ripple. In this part, we consider continuous conduction operation to derive equations that enable the selection of inductors and capacitors that meet design considerations. The following figure will aid in the analysis: Because the average current of the capacitor is zero, and the output current is equal to the average inductor current, the additional charge Q accumulated in half a period covers up the voltage ripple in the same time occurring at the output. For a certain capacitance C at the output we thus have: Vo = Q/C and since Q= I. T then Vo= I/2. T/2. 1/C 21
22 Here the division by 2 accounts for the half period change in current and time Also, during the off state in the converter, and from the basic equation of V L =Ldi/dt we have: I= Vo(1D)Ts/L Therefore, with these equations, it is now possible to select the components of the output filter while considering the desired voltage and current ripples. The Boost Converter: The boost converter is used to raise the input DC voltage to another higher output DC voltage. The approach to analyzing the boost circuit is quite similar to the methods mentioned earlier. The boost converter circuit is shown in the following schematic: Figure 9: The Boost Converter The key difference in this circuit is the utilization of the inductor at the input of the converter. During the on state, the DC input and the inductor are isolated from the remaining circuit since the diode becomes reverse biased. In this state, the inductor stores energy, so that at the off state, the DC input along with the stored energy of the inductor are applied to the load. The following presents the voltage and current waveforms for continuous conduction in both switching states: 22
23 Again the average inductor voltage and capacitor current are zero. This leaves us with the following equations that describe the boost operation of this converter: Vo/Vin=1/(1D) and Io/Iin =(1D) The Buck Boost Converter: As its name implies, the buck boost converter is capable of decreasing and increasing the input DC voltage applied. It is actually a cascade connection of a buck and a boost converter as seen in the schematic below along with the relevant waveforms: Figure 10: The Buck Boost Converter 23
24 From the above inductor waveforms, the voltages are constrained to Vin and Vo, and thus Vin.t on + Vo.t off =0. Therefore buckboost converter is characterized by the following voltage equation: Vo/Vin = D/(1D) Therefore, from the above equation, it can be seen that the duty cycle dictates the mode of operation of the converter as a buck or a boost converter. For values above 0.5, the converter operates as a boost and for D<0.5, the converter operates as a buck. The Flyback Converter: The flyback converter is an isolated converter, i.e. a high frequency transformer is utilized to transfer power from the input to the output. The circuit schematic is shown: Figure 11: The Flyback Converter As the switch is turned on and off with a specific duty cycle, the pulse wave voltage generated is fed into the high frequency transformer. Depending on the turn ratio of the transformer, the voltage is either boosted or lowered. The following waveforms show the primary voltage variations: 24
25 As seen from the waveforms, the operation resembles that of the buck boost converter but the turns ration is incorporated. Thus the input output voltage relation, knowing that the average voltage of the primary inductor is zero, is given by: Vin t on = Vot off (N1/N2 ) and Vo/Vin = N2/N1 D/(1D) The Forward Converter: The forward converter adopts the basic principle of operation of the buckforward converter, but now incorporates an additional diode and inductor as shown below: Figure 12: The Forward Converter When the switch is on, D2 becomes reversed biased and D1 is forward biased, and therefore the inductor voltage is given by: V L =Vin.N2/N1 Vo When the switch is off, the energy stored in the second turn is carried through D2 which is now forward biased to the output which leaves: V L = Vo Therefore, the input output relation is given by: Vo/Vin = N2/N1.D The operation of the forward converter is actually adopted from the buck converter but the topology is now isolated. The difference in turns ration can also play a role in decreasing the voltage. 25
26 The HBridge Converter: The Hbridge converter is shown in the following schematic: Figure 13: The HBridge Converter Switches that make vertically opposite pairs (T1 and T2) and (T3 and T4) operate simultaneously in an Hbridge converter. An interesting advantage of this type of converter is that the type of control that can be implemented. Although the switches can be driven by direct PWM control, another approach can be used that has some advantages. Using a constant duty cycle of 0.5, the input to the high frequency transformer is a square wave. The duty cycle of this wave can be controlled by phase shifting the operation of switches T3 and T4. In other words, if we delay the switching of T3 and T4, then v1= V T1,T2 V T3,T4 will have varying widths depending the magnitude of this delay: V1V2 will be: 26
27 This waveform is supplied to the transformer that is followed by a full wave rectifier. The transformer's ratio plays a role in regulating this voltage. The input output voltage equation is given by: Vo/Vin =N2/N1 This converter will be revisited and a more detailed discussion will be carried out Switch Mode DCAC inverters DCAC inverters are devices utilized to convert a DC input voltage to an alternating voltage by means of switching techniques. In this section, we will outline some basic properties of DCAC inverters and the type of control used. PWM Control of Switch Mode Inverters: Pulse width modulation is the mode of control used in switch mode inverters, and some special cases arise from this central scheme as will be shortly discussed. The following figure depicts the PWM scheme using a sinusoidal control waveform: As seen, whenever the sinusoidal control signal is higher than the triangular signal, the output is positive, and the opposite is true when the control signal is lower. This type of control is usually referred to as bipolar PWM switching. 27
28 A few terms will be defined before proceeding: The amplitude modulation ratio m a and frequency modulation m f ratio are given by: m a = V control,peak /V tri, peak and m f = f s /f 1 where fs is the frequency of the triangular waveform, and f1 is the frequency of the control sinusoidal signal, which will also be equal to the frequency of the final output voltage. As can be noticed, m a can range from 0 to 1. When m a is greater than 1, over modulation occurs, and switching is turned into ordinary square wave switching, because the sinusoidal control is constrained to two states: either totally greater or totally lower than the triangular waveform. V Ao which is the fundamental frequency of the PWM output is to be extracted by use of appropriate filter. Thus we expect that several harmonics are to be associated with the final desired output voltage. This is illustrated in the following for a certain value of m a and m f Thus harmonics are located at multiples of mf and its neighborhood as shown. Also, since ma =0.8, we see that the fundamental frequency is mavin/2. In addition, if mf is chosen to be an odd integer, only odd harmonics will be present as also seen above for mf=15, this will turn to be very beneficial when implementing the final output filter of the inverter. 28
29 A very common switchmode inverter used is the full bridge inverter, and its circuit is shown below: Figure 14: The switchmode inverter used is the full bridge inverter With a PWM output then fed into a low pass filter, the sinusoidal output should have a frequency of f1 and the amplitude is given by Vo =m a Vin. A modified version of the PWM control discussed earlier is the unipolar PWM control. In this method, the switches are not switched simultaneously as before, but the switches of each leg is controlled separately by the use of two sinusoidal control signals as seen below: 29
30 The waveforms correspond to the full bridge circuit shown earlier. As can be seen from the waveforms, each sinusoidal signal will take care of one leg of the single phase inverter. This way, each switch will experience a positive voltage based on the sinusoidal and triangular wave comparisons. The final output obtained by V AN V BN will constitute the last waveform shown in (d). The main and crucial advantage of unipolar switching is its effect in adjusting the harmonic spectrum. The advantages are seen in the following illustration: The harmonics are located at even multiples of mf with amplitudes varying faster. Still the fundamental frequency is attenuated by ma as in the case of bipolar switching. 30
31 3.3 The Initial Proposed Solution Among the different topologies examined, in this project we require the topology to be isolated. We also reject the single stage DC/AC inverter because of its control problems. The two stage inverter implemented by UCF is an innovative design however it is very complex to realize given the special DC output (rectified sinewave) of the first stage. We therefore choose the two stage inverter (DC/DC + DC/AC). We still have to decide where to place the isolating transformer. We could put it just before the output or between the incorporate it in the DC/DC stage. The latter solution is better since the transformer would be a high frequency one, thus smaller and lighter than the output low frequency one. The DC/AC inverter will be implemented as a single phase HBridge controlled by bipolar switching. The isolated DC/DC can either be a flyback/forward or a rectified HBridge. Since we wish to operate at a high frequency (50KHz) we prefer to go for the HBridge followed by a rectifier and switching through a phase shift mode because it would allow us to have switching pulse trains always having a 0.5 duty cycle. In this case, only the phase shift between the different pulse trains will control the output voltage. If we use a flyback/forward converter, we would need to vary the duty cycle to control the output voltage. However, at high frequency if the duty cycle happens to be too small or too big, the transistors might not have enough time to switch on and off and we would have a drop in efficiency. For these reasons, a full wave rectifier preceded by an HBridge is adopted as our isolated DC/DC converter. As for the MPPT algorithm, we use the adaptive P&O program for its simplicity and high efficiency. 3.4 Problems Faced with Initially Proposed Solution As stated in the initial design, we needed to use a high frequency transformer in the design of the Hbridge in the DcDc converter; but unfortunately we were not able to locate the required transformer or even the right core that can handle saturation problems due to high frequency operation. 31
32 Due to the limitations of the RIO card concerning division and floating point operations, we were not able to completely implement the adaptive P&O algorithm. For the inverter stage, in the fall semester we had reached the conclusion that the best topology was that of the HBridge because of its high power handling capability as well as its good operation at high switching frequency. However while building our first HBridge prototype we noticed that the main problem resided in the drivers. The HBridge has four switches and the two switches that lie on the high voltage side require a floating reference. The only driver that is available in Lebanon for this application is the IRF2110. However this driver isn t robust at all and we discovered that it caused many problems when used to drive a high side switch. It is possible to make it drive a resistive load however we need to force a dead time through a processor. However, once we introduce a reactive load, some problems persist and the high side output of the driver often blows up. For reliability purposes, this driver was abandoned. More robust drivers exist however such built in chips aren t available in Lebanon. 3.5 The Implemented solution Due to the circumstances we faced, we had to look over the HBridge design for the DCDC converter, and decided to replace it with a simple singleswitch boost converter. This will offer ease of control and reliable operation. Yet as a consequence we had to bear with the bulky low frequency transformer placed at the output of the inverter. Regarding the MPPT algorithm, we used a regular P&O algorithm that will offer high efficiency. In order to proceed with the implementation of the inverter, we focused on finding a reliable driver, able to drive high side switches. We finally selected the IGBT driver developed by the Lebanese company SACCAL Systems. We tested the half bridge before going to the HBridge and we found good results. The half bridge also operates well at high frequency and has the advantage of having 2 switches instead of four. However one of its major drawbacks is that its output is half the input DC voltage. Another major problem was found to be the reference point. In the half bridge we split 32
33 Vcc into +Vcc/2 and Vcc/2 however to do so we use capacitors or two separate boosts. The problem resides in the discrepancy that may be present in either the capacitors or boost converters. The slightest difference causes a DC component at the output. This was highly problematic since it caused the isolating transformer at the output to saturate. The final solution that we adopted and that solved this DC saturation problem was reached by using a modified center tapped half bridge. This system was inspired of the center tapped full wave rectifier. Its advantages are that we only use one booster and that the output amplitude is the input one. It also doesn t require highside drivers. 33
34 4. PROJECT DESIGN Now that we have all the different required components from hardware to software, the complete project design is available. The system has four basic components which include the PV Panel, the DCDC converter, the DCAC inverter, and the controller. The project design schematic is shown in figure 4.1 below. PV Panel DCDC Converter DCAC Inverter Filter Power Switches Controller RIO (Reconfigurable Input Output) PCI7831 R + FPGA Transformer Transformer Utility Grid Stand Alone AC Load Figure 15: Complete design schematic of the system 34
35 Schematic of the Initial Design DCDC Converter Block DCAC Inverter Block HBridge High Frequency Transformer Rectifier & Filter HBridge Driver Circuit Driver Circuit Control Control Figure 16: DCDC Converter Block for Initial Design Figure 17: DCAC Inverter block for Initial Design Schematic of Implemented Project Design DCDC Converter Block DCAC Inverter Block Boost Converter Half Bridge Inverter Center Tapped Transformer Driver Circuit Driver Circuit Control Control Figure 18: DCDC Converter Block for Implemented Design Figure 19: DCAC Inverter block for Implemented Design 35
36 4.1 Solar Cell Modeling The Solar Panel Used (characteristic): The PV panel to be used in this project is the PV120 panel Golden Genesis Company. The panel is actually available at AUB, and has the following ratings at 1000W/m 2 solar irradiance, and 25 o C cell temperature:  Rated Power:120W  Open Circuit Voltage: 21V  Short Circuit Current: 7.7A  Rated Voltage: 16.9V  Rated Current: 7.1A  Maximum System Open Circuit Voltage: 600V  Bypass Diode: 8A Three panels connected in series are actually available thus tripling the rated voltage of the panel, and increasing the total panel power to 360Watts. The PV Model: As was previously mentioned in the literature review section, three different models were presented for a PV panel. Because the single diode model with series and shunt resistances provides a simple mathematical model to deal with, as well as a very realistic IV curve, we shall be adopting this model for our project. We include again the IV equation given by: 36
37 4.2 DCDC Converter DCDC Converter Design: The DCDC converter being implemented in this project is the simple singleswitch boost converter. Although using an HBridge will introduce the advantage of having a smaller transformer and an overall practical product, the simple singleswitch design introduces reliability in the system because of the little number of components used on one side, and the availability of the hardwearing and reliable components on another. Furthermore, this DCDC converter design will prove later excellent functioning and will deliver any required output voltage promptly. This boost converter includes one MOSFET that will be controlled through a square wave control signal. The following simple schematic devises the latter scheme: Figure 20: DCDC Converter circuit The major parts involved in designing the boost converter are related to the choice of the inductor and capacitor. This choice is primarily governed by the overall power rating this converter should attain, switching frequency, and quality of output DC voltage in terms of ripple in the DC output. Based on the discussion presented earlier in section on the operation of the boost converter, we note that the average current in the inductor at boundary conditions is half that of the peak current passing in the inductor; with the latter quantity equaling 0.5V in D/L. In addition using the fact that the input current and the inductor current are the same, and using the boost inputoutput voltage equations we can finally reach the following equation: I out = V out D(1D) 2 /(2F s L) 37
38 The choice of the capacitance as mentioned is directly linked to the ripple in the output voltage. The figure shown below will further clarify the issue at hand: The shaded area in the above figure represents the total charge Q accumulated in the capacitor as the diode current flows through it and the resistor. This charge is exactly equal to V o C, where V o is the value of the offset ripple away from our required DC value. In percentile notion, the percent ripple the boost should be able to provide is given by: %V ripple = Q/C =I o.d/(c.f s.v o ) Therefore, to proceed with the design of the boost, several parameters have to be defined and set, in order to choose the correct values of the inductance and capacitance. And as previously mentioned, these values depend on the power rating, switching frequency, ripple percentage. Our system will consist of three photovoltaic panels for a total power of 360W. Each panel has the following characteristics: P max = 120 W V oc = 21 V V (at Pmax)= 16.9 V I (at Pmax)= 7.1 V 38
39 Our input will be the voltage resulting from the series combination of three panels (i.e. P max =360W). We assume that the three panels operate at the same voltage given the fact that they are adjacent to each other and are exposed to similar insolation and temperature. The boost should be able to handle all power delivered and is thus chosen to be of a rating of 400W. For an output of 100V, this means that is should handle a 4A current from the PV panels without difficulty. In addition, our boost should deliver the required output voltage for a range of input values, and this has been decided to be between 30V to 60V input and an output of 100V. Using the inputoutput voltage equations of the boost, we have D ranging between: [ ] As for the switching frequency, we have chosen it to be 25 khz. The equations show that as the frequency of switching increases, both the inductance and capacitance decrease in value, and thus in volume, leading to an increase in the overall product's practicality. The determination of the inductance now is straightforward, by considering the parameters just mentioned and defined. The values of D yield in two inductance values, and we choose the larger values which turned out to be around 63μH. As for the determination of the capacitance, the boost should deliver an output with a maximum of 1% ripple, and therefore C will equal 112μF. 39
40 4.3 DCAC Inverter and control Design of Inverter Power Stage: For the reasons exposed in the Selected Typology section, we adopted the center tapped half bridge inverter. Figure 21: DCAC Inverter circuit Our first task was to choose the switches we would use. Our choice went on the IGBT. These switches are a combination of a MOSFET and a BJT and as such offer a good trade off taking the advantages of both switches. They have high power capability, low onstate resistance, as well as relatively high switching speeds capabilities. Their high voltage capabilities make them suitable for our application. The input to the inverter stage coming from the boost converter will be typically about 100V so in theory this will be approximately the voltage across each switch. However in the simulations on PSPICE and in practice we noticed that especially when having a reactive load (which is our case since we are using a transformer and connecting it to the grid) we have some ringing and overshoot at across the switch and this could reach as high as 400V when using a 100V input. In order to solve this problem we selected the BUP307 IGBT. The origin of the overshoot comes from the inductive kick since we have a high di/dt, the presence of an inductance will give rise to a high voltage given by Ldi/dt. In order to reduce the overshoot and ringing on the switches, we used RCD snubbers. 40
41 When the switch turns off, the energy flows in its snubber and charges the capacitor through a diode (charge reaches about 200V). Once the switch turns off again, the capacitor discharges into the switch through the resistor and the diode is reversed biased. To select the snubbers components we first used fast switching diodes able to operate at the 33KHz switching frequency we are using. Next we estimated the RC time constant by approximating it with the inverse of the frequency. RC=1/33KHz With this initial guess, we tried to slightly increase it or decrease it and to observe the ringing and overshoot behavior. Because we didn t have a broad choice of capacitances, we fixed the capacitor and modified the RC constant by modifying the resistors. However, we kept in mind that we shouldn t use a resistor that was too small since when the capacitor discharges its maximum current is given by the ratio of the stored voltage over the discharge resistance. A small resistance might lead to a high current and damage the switch. We finally reached an acceptable overshoot of about 50% when using the following Snubber: Capacitor: 4.92 uf, 400V Resistor: 10 kohms Diode: Fast recovery RUR30100 When using a duty cycle inferior to 0.9 we reached a voltage at the primary of the transformer around 54V rms, knowing that to connect it to the grid we need a voltage of 220V rms, we calculated the turn ratio to be around The exact turn ratio isn t important since we could increase or decrease the output voltage by playing with the duty cycle. A higher duty cycle would yield a higher output voltage. We then ordered a low frequency transformer center tapped at the primary with the following specifications: 110 turns for each half primary and 450 turns at the secondary for a turn ratio of The number of turns was calculated by the manufacturer and a big cross section of 72cm^2 was chosen in order to compensate for the high magnetizing current that appeared because of the high switching frequency on the primary side. At 33KHz, steel cores saturate easily and a better core would have been a ferrite one. However, they are unavailable in Lebanon. 41
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