U S DESCRIPTIO. LT1110 Micropower DC-DC Converter Adjustable and Fixed 5V, 12V

Transcription

1 Micropower DC-DC Converter Adjustable and Fixed 5V, 12V FEATRE perates at upply Voltages From 1.0V to 30V Works in tep-p or tep-down Mode nly Three External ff-the-helf Components Required Low-Battery Detector Comparator n-chip ser-adjustable Current Limit Internal 1A Power witch Fixed or Adjustable utput Voltage Versions pace-aving 8-Pin MiniDIP or 8 Package Pagers Cameras ingle-cell to 5V Converters Battery Backup upplies Laptop and Palmtop Computers Cellular Telephones Portable Instruments Laser Diode Drivers Hand-Held Inventory Computers DECRIPTI The is a versatile micropower DC-DC converter. The device requires only three external components to deliver a fixed output of 5V or 12V. The very low minimum supply voltage of 1.0V allows the use of the in applications where the primary power source is a single cell. An on-chip auxiliary gain block can function as a low battery detector or linear post regulator. The 70kHz oscillator allows the use of surface mount inductors and capacitors in many applications. Quiescent current is just 300µA, making the device ideal in remote or battery powered applications where current consumption must be kept to a minimum. The device can easily be configured as a step-up or step-down converter, although for most step-down applications or input sources greater than 3V, the LT1111 is recommended. witch current limiting is user-adjustable by adding a single external resistor. nique reverse battery protection circuitry limits reverse current to safe, nondestructive levels at reverse supply voltages up to 1.6V. TYPICAL All urface Mount ingle Cell to 5V Converter Efficiency 1.5V AA CELL* MIDA CD54-470K 47µH ENE 8 MBR120T3 5V 15µF TANTALM EFFICIENCY (%) V IN = 1.50V V IN = 1.25V V IN = 1.00V PERATE WITH CELL VLTAGE 1.0V *ADD 10 µ F DECPLING CAPACITR IF BATTERY I MRE THAN 2" AWAY FRM. TA01 LAD CRRENT (ma) TA02 1

2 ABLTE AXI RATI G W W W upply Voltage, tep-p Mode... 15V upply Voltage, tep-down Mode... 36V Pin Voltage... 50V Pin Voltage V to V IN Feedback Pin Voltage () V witch Current A Maximum Power Dissipation mW perating Temperature Range... 0 C to 70 C torage Temperature Range C to 150 C Lead Temperature (oldering, 10 sec.) C PACKAGE/RDER I FR I LIM 1 V IN TP VIEW N8 PACKAGE 8-LEAD PLATIC DIP *FIXED VERIN T JMAX = 90 C, θ JA = 130 C/W I LIM 1 V IN TP VIEW 8 PACKAGE 8-LEAD PLATIC IC *FIXED VERIN T JMAX = 90 C, θ JA = 150 C/W 8 (ENE)* 7 ET 6 A (ENE)* ET A0 W Consult factory for Industrial and Military grade parts. ATI RDER PART NMBER CN8 CN8-5 CN C8 C8-5 C PART MARKING ELECTRICAL CHARA CTERITIC T A = 25 C, V IN = 1.5V, unless otherwise noted. YMBL PARAMETER CNDITIN MIN TYP MAX NIT I Q Quiescent Current witch ff 300 µa V IN Input Voltage tep-p Mode V V tep-down Mode 30 V Comparator Trip Point Voltage (Note 1) mv V T utput ense Voltage -5 (Note 2) V -12 (Note 2) V Comparator Hysteresis 4 8 mv utput Hysteresis mv mv f C scillator Frequency khz DC Duty Cycle Full Load (V < V REF ) % t N witch N Time µs I Feedback Pin Bias Current, V = 0V na I ET et Pin Bias Current V ET = V REF na V A A utput Low I A = 300µA, V ET = 150mV V Reference Line Regulation 1.0V V IN 1.5V %/V 1.5V V IN 12V %/V 2

3 ELECTRICAL CHARA CTERITIC T A = 25 C, V IN = 1.5V, unless otherwise noted. YMBL PARAMETER CNDITIN MIN TYP MAX NIT V CEAT witch aturation Voltage V IN = 1.5V, I W = 400mA mv tep-p Mode 600 mv V IN = 1.5V, I W = 500mA mv 750 mv V IN = 5V, I W = 1A mv A V A2 Error Amp Gain R L = 100kΩ (Note 3) V/V I REV Reverse Battery Current (Note 4) 750 ma I LIM Current Limit 220Ω Between I LIM and V IN 400 ma Current Limit Temperature 0.3 %/ C Coefficient I LEAK witch FF Leakage Current Measured at Pin 1 10 µa V Maximum Excursion Below I 10µA, witch ff mv The denotes the specifications which apply over the full operating temperature range. Note 1: This specification guarantees that both the high and low trip point of the comparator fall within the 210mV to 230mV range. Note 2: This specification guarantees that the output voltage of the fixed versions will always fall within the specified range. The waveform at the sense pin will exhibit a sawtooth shape due to the comparator hysteresis. Note 3: 100kΩ resistor connected between a 5V source and the A pin. Note 4: The is guaranteed to withstand continuous application of 1.6V applied to the and pins while V IN, I LIM, and pins are grounded. TYPICAL PERFR A W CE CHARA CTERITIC CILLATR FREQENCY (KHz) scillator Frequency scillator Frequency witch n Time FREQENCY (KHz) N TIME (µs) TEMPERATRE ( C) TPC01 INPT VLTAGE (V) TPC02 TEMPERATRE ( C) TPC03 3

4 TYPICAL PERFR A W CE CHARA CTERITIC DTY CYCLE (%) aturation Voltage Duty Cycle witch aturation Voltage tep-p Mode TEMPERATRE ( C) TPC04 V CEAT (mv) V IN = 1.5V I W = 500mA TEMPERATRE ( C) TPC05 V CEAT (V) V IN = 1.0V V IN = 1.5V V = 1.2V IN I WITCH (A) V = 2.0V IN V IN = 3.0V V IN = 5.0V TPC06 N VLTAGE (V) witch n Voltage Minimum/Maximum Frequency vs tep-down Mode n Time Quiescent Current 1.4 V IN = 12V I WITCH (A) CILLATR FREQENCY (KHz) C T A 70 C WITCH N TIME (µs) INPT VLTAGE (V) TPC07 TPC08 TPC09 QIECENT CRRENT (µa) Maximum witch Current vs Maximum witch Current vs Quiescent Current R LIM tep-p R LIM tep-down QIECENT CRRENT (µa) WITCH CRRENT (A) TEP-P MDE V IN 5V WITCH CRRENT (A) TEP-DWN MDE V IN = 12V TEMPERATRE ( C) TPC10 R LIM (Ω) TPC11 R LIM (Ω) TPC12 4

5 TYPICAL PERFR A W CE CHARA CTERITIC BIA CRRENT (na) 160 et Pin Bias Current Pin Bias Current Reference Voltage BIA CRRENT (na) V REF (mv) TEMPERATRE ( C) TPC13 TEMPERATRE ( C) TPC14 TEMPERATRE ( C) TPC15 PI F CTI I LIM (Pin 1): Connect this pin to V IN for normal use. Where lower current limit is desired, connect a resistor between I LIM and V IN. A 220Ω resistor will limit the switch current to approximately 400mA. V IN (Pin 2): Input supply voltage. (Pin 3): Collector of power transistor. For step-up mode connect to inductor/diode. For step-down mode connect to V IN. (Pin 4): Emitter of power transistor. For step-up mode connect to ground. For step-down mode connect to inductor/diode. This pin must never be allowed to go more than a chottky diode drop below ground. (Pin 5): Ground. A (Pin 6): Auxiliary Gain Block (GB) output. pen collector, can sink 300µA. ET (Pin 7): GB input. GB is an op amp with positive input connected to ET pin and negative input connected to 220mV reference. /ENE (Pin 8): n the (adjustable) this pin goes to the comparator input. n the -5 and -12, this pin goes to the internal application resistor that sets output voltage. BLCKDAGRA I W ET V IN A2 A GAIN BLCK/ERRR AMP I LIM 220mV REFERENCE A1 CILLATR Q1 CMPARATR DRIVER BD01 5

6 PERATI The is a gated oscillator switcher. This type architecture has very low supply current because the switch is cycled only when the feedback pin voltage drops below the reference voltage. Circuit operation can best be understood by referring to the block diagram above. Comparator A1 compares the pin voltage with the 220mV reference signal. When drops below 220mV, A1 switches on the 70kHz oscillator. The driver amplifier boosts the signal level to drive the output NPN power switch Q1. An adaptive base drive circuit senses switch current and provides just enough base drive to ensure switch saturation without overdriving the switch, resulting in higher efficiency. The switch cycling action raises the output voltage and pin voltage. When the voltage is sufficient to trip A1, the oscillator is gated off. A small amount of hysteresis built into A1 ensures loop stability without external frequency compensation. When the comparator is low the oscillator and all high current circuitry is turned off, lowering device quiescent current to just 300µA for the reference, A1 and A2. The oscillator is set internally for 10µs N time and 5µs FF time, optimizing the device for step-up circuits where V T 3V IN, e.g., 1.5V to 5V. ther step-up ratios as well as step-down (buck) converters are possible at slight losses in maximum achievable power output. A2 is a versatile gain block that can serve as a low battery detector, a linear post regulator, or drive an under voltage lockout circuit. The negative input of A2 is internally connected to the 220mV reference. An external resistor divider from V IN to provides the trip point for A2. The A output can sink 300µA (use a 47k resistor pull up to 5V). This line can signal a microcontroller that the battery voltage has dropped below the preset level. To prevent the gain block from operating in its linear region, a 2MΩ resistor can be connected from A to ET. This provides positive feedback. A resistor connected between the I LIM pin and V IN adjusts maximum switch current. When the switch current exceeds the set value, the switch is turned off. This feature is especially useful when small inductance values are used with high input voltages. If the internal current limit of 1.5A is desired, I LIM should be tied directly to V IN. Propagation delay through the current limit circuitry is about 700ns. In step-up mode, is connected to ground and drives the inductor. In step-down mode, is connected to V IN and drives the inductor. utput voltage is set by the following equation in either step-up or stepdown modes where is connected from to and is connected from V T to. V mv R T = ( ) 1. ( 01) -5, -12 BLCK DIAGRA 6 V IN 220mV REF ET 300kΩ A2 A1 ENE A GAIN BLCK/ERRR AMP CMPARATR CILLATR I LIM DRIVER -5: = 13.8kΩ -12: = 5.6kΩ W BD02 Q1-5, -12 PERATI The -5 and -12 fixed output voltage versions have the gain setting resistors on-chip. nly three external components are required to construct a 5V or 12V output converter. 16µA flows through and in the -5, and 39µA flows in the -12. This current represents a load and the converter must cycle from time to time to maintain the proper output voltage. utput ripple, inherently present in gated oscillator designs, will typically run around 90mV for the -5 and 200mV for the -12 with the proper inductor/capacitor selection. This output ripple can be reduced considerably by using the gain block amp as a pre-amplifier in front of the pin. ee the Applications section for details.

7 Inductor election General A DC-DC converter operates by storing energy as magnetic flux in an inductor core, and then switching this energy into the load. ince it is flux, not charge, that is stored, the output voltage can be higher, lower, or opposite in polarity to the input voltage by choosing an appropriate switching topology. To operate as an efficient energy transfer element, the inductor must fulfill three requirements. First, the inductance must be low enough for the inductor to store adequate energy under the worst case condition of minimum input voltage and switch N time. The inductance must also be high enough so maximum current ratings of the and inductor are not exceeded at the other worst case condition of maximum input voltage and N time. Additionally, the inductor core must be able to store the required flux; i.e., it must not saturate. At power levels generally encountered with based designs, small surface mount ferrite core units with saturation current ratings in the 300mA to 1A range and DCR less than 0.4Ω (depending on application) are adequate. Lastly, the inductor must have sufficiently low DC resistance so excessive power is not lost as heat in the windings. An additional consideration is Electro- Magnetic Interference (EMI). Toroid and pot core type inductors are recommended in applications where EMI must be kept to a minimum; for example, where there are sensitive analog circuitry or transducers nearby. Rod core types are a less expensive choice where EMI is not a problem. Minimum and maximum input voltage, output voltage and output current must be established before an inductor can be selected. Inductor election tep-p Converter I FR W In a step-up, or boost converter (Figure 4), power generated by the inductor makes up the difference between input and output. Power required from the inductor is determined by ( )( ) P = V V V I ( 01) L T D IN MIN T ATI where V D is the diode drop (0.5V for a 1N5818 chottky). Energy required by the inductor per cycle must be equal or greater than PL ( 02) fc in order for the converter to regulate the output. When the switch is closed, current in the inductor builds according to I L V () t = R' IN Rt ' 1 e L ( 03) where R' is the sum of the switch equivalent resistance (0.8Ω typical at 25 C) and the inductor DC resistance. When the drop across the switch is small compared to V IN, the simple lossless equation IL ()= t L t ( 04) can be used. These equations assume that at t = 0, inductor current is zero. This situation is called discontinuous mode operation in switching regulator parlance. etting t to the switch N time from the specification table (typically 10µs) will yield I PEAK for a specific L and V IN. nce I PEAK is known, energy in the inductor at the end of the switch N time can be calculated as EL= 1 2 LI 2 PEAK ( 05) E L must be greater than P L /f C for the converter to deliver the required power. For best efficiency I PEAK should be kept to 1A or less. Higher switch currents will cause excessive drop across the switch resulting in reduced efficiency. In general, switch current should be held to as low a value as possible in order to keep switch, diode and inductor losses at a minimum. As an example, suppose 12V at 120mA is to be generated from a 4.5V to 8V input. Recalling equation (01), ( )( )= P L = 12V 0. 5V 4. 5V 120mA 960mW. ( 06) Energy required from the inductor is PL 960mW = = 13. 7µ J. ( 07) fc 70kHz 7

8 I FR W ATI Picking an inductor value of 47µH with 0.2Ω DCR results in a peak switch current of 45. V 10. W 10ms IPEAK = 1 e 47mH = 862mA. ( 08) 10. W ubstituting I PEAK into Equation 05 results in 1 2 EL = ( 47µ H)( A) = 17. 5µ J. ( 09) 2 ince 17.5µJ > 13.7µJ, the 47µH inductor will work. This trial-and-error approach can be used to select the optimum inductor. Keep in mind the switch current maximum rating of 1.5A. If the calculated peak current exceeds this, an external power transistor can be used. A resistor can be added in series with the I LIM pin to invoke switch current limit. The resistor should be picked such that the calculated I PEAK at minimum V IN is equal to the Maximum witch Current (from Typical Performance Characteristic curves). Then, as V IN increases, switch current is held constant, resulting in increasing efficiency. Inductor election tep-down Converter The step-down case (Figure 5) differs from the step-up in that the inductor current flows through the load during both the charge and discharge periods of the inductor. Current through the switch should be limited to ~800mA in this mode. Higher current can be obtained by using an external switch (see Figure 6). The I LIM pin is the key to successful operation over varying inputs. After establishing output voltage, output current and input voltage range, peak switch current can be calculated by the formula 2I V V I T T D PEAK = DC VW V D where DC = duty cycle (0.69) V W = switch drop in step-down mode V D = diode drop (0.5V for a 1N5818) I T = output current ( 10) V T = output voltage V IN = minimum input voltage V W is actually a function of switch current which is in turn a function of V IN, L, time and V T. To simplify, 1.5V can be used for V W as a very conservative value. nce I PEAK is known, inductor value can be derived from MIN VW VT L = tn IPEAK where t N = switch N time (10µs). ( 11) Next, the current limit resistor R LIM is selected to give I PEAK from the R LIM tep-down Mode curve. The addition of this resistor keeps maximum switch current constant as the input voltage is increased. As an example, suppose 5V at 250mA is to be generated from a 9V to 18V input. Recalling Equation (10), I 2 250mA 069. PEAK = ( ) = 498mA. ( 12) Next, inductor value is calculated using Equation (11) L = 10µ s = 50µ H. ( 13) 498mA se the next lowest standard value (47µH). Then pick R LIM from the curve. For I PEAK = 500mA, R LIM = 82Ω. Inductor election Positive-to-Negative Converter Figure 7 shows hookup for positive-to-negative conversion. All of the output power must come from the inductor. In this case, ( )( ) PL = VT VD IT. ( 14) In this mode the switch is arranged in common collector or step-down mode. The switch drop can be modeled as a 0.75V source in series with a 0.65Ω resistor. When the 8

9 switch closes, current in the inductor builds according to I L ( )= V L R' where R' = 0.65Ω DCR L V L = V IN 0.75V Rt ' 1 e L ( 15) As an example, suppose 5V at 75mA is to be generated from a 4.5V to 5.5V input. Recalling Equation (14), ( )( )= PL = 5V 0. 5V 75mA 413mW. ( 16) Energy required from the inductor is PL fc 413mW = = 59. µ J. ( 17) 70kHz Picking an inductor value of 56µH with 0.2Ω DCR results in a peak switch current of IPEAK ( 45. V 075. V ). 085Ω 10µ s = 1 e 56µ H ma = 621. ( 18). Ω. Ω ( ) I FR W ATI ubstituting I PEAK into Equation (04) results in capacitors provide still better performance at more expense. We recommend -CN capacitors from anyo Corporation (an Diego, CA). These units are physically quite small and have extremely low ER. To illustrate, Figures 1, 2 and 3 show the output voltage of an based converter with three 100µF capacitors. The peak switch current is 500mA in all cases. Figure 1 shows a prague 501D, 25V aluminum capacitor. V T jumps by over 120mV when the switch turns off, followed by a drop in voltage as the inductor dumps into the capacitor. This works out to be an ER of over 240mΩ. Figure 2 shows the same circuit, but with a prague 150D, 20V tantalum capacitor replacing the aluminum unit. utput jump is now about 35mV, corresponding to an ER of 70mΩ. Figure 3 shows the circuit with a 16V -CN unit. ER is now only 20mΩ. 50mV/DIV 1 2 EL = ( 56µ H)( A) = 10. 8µ J. ( 19) 2 ince 10.8µJ > 5.9µJ, the 56µH inductor will work. 5 µ s/div Figure 1. Aluminum TA19 With this relatively small input range, R LIM is not usually necessary and the I LIM pin can be tied directly to V IN. As in the step-down case, peak switch current should be limited to ~800mA. 50mV/DIV Capacitor election 5 µ s/div TA20 electing the right output capacitor is almost as important as selecting the right inductor. A poor choice for a filter capacitor can result in poor efficiency and/or high output ripple. rdinary aluminum electrolytics, while inexpensive and readily available, may have unacceptably poor Equivalent eries Resistance (ER) and EL (inductance). There are low ER aluminum capacitors on the market specifically designed for switch mode DC-DC converters which work much better than general-purpose units. Tantalum 50mV/DIV Figure 2. Tantalum 5 µ s/div Figure 3. -CN TA21 9

10 Diode election peed, forward drop, and leakage current are the three main considerations in selecting a catch diode for converters. General purpose rectifiers such as the 1N4001 are unsuitable for use in any switching regulator application. Although they are rated at 1A, the switching time of a 1N4001 is in the 10µs-50µs range. At best, efficiency will be severely compromised when these diodes are used; at worst, the circuit may not work at all. Most circuits will be well served by a 1N5818 chottky diode, or its surface mount equivalent, the MBR130T3. The combination of 500mV forward drop at 1A current, fast turn N and turn FF time, and 4µA to 10µA leakage current fit nicely with requirements. At peak switch currents of 100mA or less, a 1N4148 signal diode may be used. This diode has leakage current in the 1nA-5nA range at 25 C and lower cost than a 1N5818. (You can also use them to get your circuit up and running, but beware of destroying the diode at 1A switch currents.) tep-p (Boost Mode) peration A step-up DC-DC converter delivers an output voltage higher than the input voltage. tep-up converters are not short circuit protected since there is a DC path from input to output. The usual step-up configuration for the is shown in Figure 4. The first pulls low causing V IN V CEAT to appear across L1. A current then builds up in L1. At the end of the switch N time the current in L1 is 1 : IPEAK = L t N ( 20) V IN * = PTINAL 10 R3* I FR W L1 D1 Figure 4. tep-p Mode Hookup. ATI C1 V T TA14 Immediately after switch turn off, the voltage pin starts to rise because current cannot instantaneously stop flowing in L1. When the voltage reaches V T V D, the inductor current flows through D1 into C1, increasing V T. This action is repeated as needed by the to keep V at the internal reference voltage of 220mV. and set the output voltage according to the formula R VT = 2 1 ( 220mV). ( 21) tep-down (Buck Mode) peration A step-down DC-DC converter converts a higher voltage to a lower voltage. The usual hookup for an based step-down converter is shown in Figure 5. V IN C2 R3 220Ω L1 D1 1N5818 Figure 5. tep-down Mode Hookup C1 V T TA15 When the switch turns on, pulls up to V IN V W. This puts a voltage across L1 equal to V IN V W V T, causing a current to build up in L1. At the end of the switch N time, the current in L1 is equal to VW VT IPEAK = t N. ( 22) L When the switch turns off, the pin falls rapidly and actually goes below ground. D1 turns on when reaches 0.4V below ground. D1 MT BE A CHTTKY DIDE. The voltage at must never be allowed to go below 0.5V. A silicon diode such as the 1N4933 will allow to go to 0.8V, causing potentially destructive power Note 1: This simple expression neglects the effects of switch and coil resistance. This is taken into account in the Inductor election section.

11 dissipation inside the. utput voltage is determined by R VT = 2 1 ( 220mV). ( 23) R3 programs switch current limit. This is especially important in applications where the input varies over a wide range. Without R3, the switch stays on for a fixed time each cycle. nder certain conditions the current in L1 can build up to excessive levels, exceeding the switch rating and/or saturating the inductor. The 220Ω resistor programs the switch to turn off when the current reaches approximately 800mA. When using the in stepdown mode, output voltage should be limited to 6.2V or less. Higher output voltages can be accommodated by inserting a 1N5818 diode in series with the pin (anode connected to ). Higher Current tep-down peration utput current can be increased by using a discrete PNP pass transistor as shown in Figure 6. serves as a current limit sense. When the voltage drop across equals a V BE, the switch turns off. For temperature compensation a chottky diode can be inserted in series with the I LIM pin. This also lowers the maximum drop across to V BE V D, increasing efficiency. As shown, switch current is limited to 2A. Inductor value can be calculated based on formulas in the Inductor election tep-down V IN 25V MAX C2 V IN 0.3Ω I L I FR 220 W Q1 MJE210 R ZETEX ZTX789A R3 330 ATI Figure 6. Q1 Permits Higher-Current witching. Functions as Controller. R5 R4 L1 D1 1N5821 V T C1 R4 V T = 220mV (1 R5) TA16 Converter section with the following conservative expression for V W : VW = V VAT 09. V. ( 24) provides a current path to turn off Q1. R3 provides base drive to Q1. R4 and R5 set output voltage. Inverting Configurations The can be configured as a positive-to-negative converter (Figure 7), or a negative-to-positive converter (Figure 8). In Figure 7, the arrangement is very similar to a step-down, except that the high side of the feedback is referred to ground. This level shifts the output negative. As in the step-down mode, D1 must be a chottky diode, and V T should be less than 6.2V. More negative output voltages can be accommodated as in the prior section. V IN C2 R3 L1 D1 1N5818 C1 V T TA03 Figure 7. Positive-to-Negative Converter In Figure 8, the input is negative while the output is positive. In this configuration, the magnitude of the input voltage can be higher or lower than the output voltage. A level shift, provided by the PNP transistor, supplies proper polarity feedback information to the regulator. V IN C2 A L1 Figure 8. Negative-to-Positive Converter D1 C1 ( ) 2N3906 V T V T = 220mV 0.6V TA04 11

12 12 sing the I LIM Pin I FR W ATI The switch can be programmed to turn off at a set switch current, a feature not found on competing devices. This enables the input to vary over a wide range without exceeding the maximum switch rating or saturating the inductor. Consider the case where analysis shows the must operate at an 800mA peak switch current with a 2.0V input. If V IN rises to 4V, peak current will rise to 1.6A, exceeding the maximum switch current rating. With the proper resistor selected (see the Maximum witch Current vs R LIM characteristic), the switch current will be limited to 800mA, even if the input voltage increases. Another situation where the I LIM feature is useful occurs when the device goes into continuous mode operation. This occurs in step-up mode when VT VDIDE 1 <. ( 25) V V 1 DC IN W When the input and output voltages satisfy this relationship, inductor current does not go to zero during the switch FF time. When the switch turns on again, the current ramp starts from the non-zero current level in the inductor just prior to switch turn on. As shown in Figure 9, the inductor current increases to a high level before the comparator turns off the oscillator. This high current can cause excessive output ripple and requires oversizing the output capacitor and inductor. With the I LIM feature, however, the switch current turns off at a programmed level as shown in Figure 10, keeping output ripple to a minimum. Figure 11 details current limit circuitry. ense transistor Q1, whose base and emitter are paralleled with power switch Q2, is ratioed such that approximately 0.5% of Q2 s collector current flows in Q1 s collector. This current is passed through internal 80Ω resistor and out through the I LIM pin. The value of the external resistor connected between I LIM and V IN set the current limit. When sufficient switch current flows to develop a V BE across R LIM, Q3 turns on and injects current into the oscillator, turning off the switch. Delay through this circuitry is approximately 800ns. The current trip point becomes less accurate for switch N times less than 3µs. Resistor values programming switch N time for 800ns or less will cause spurious response in the switch circuitry although the device will still maintain output regulation. IL N WITCH FF Figure 9. No Current Limit Causes Large Inductor Current Build-p IL N WITCH FF PRGRAMMED CRRENT LIMIT TA05 TA06 Figure 10. Current Limit Keeps Inductor Current nder Control V IN Q3 CILLATR R LIM (EXTERNAL) DRIVER I LIM 80Ω (INTERNAL) Q1 Q2 TA17 Figure 11. Current Limit Circuitry sing the Gain Block The gain block (GB) on the can be used as an error amplifier, low battery detector or linear post regulator. The gain block itself is a very simple PNP input op amp with an open collector NPN output. The negative input of the gain block is tied internally to the 220mV reference. The positive input comes out on the ET pin.

13 I FR W ATI Arrangement of the gain block as a low battery detector is straightforward. Figure 12 shows hookup. and need only be low enough in value so that the bias current of the ET input does not cause large errors. 33kΩ for is adequate. R3 can be added to introduce a small amount of hysteresis. This will cause the gain block to snap when the trip point is reached. Values in the 1M-10M range are optimal. The addition of R3 will change the trip point, however. 5V utput ripple of the, normally 90mV at 5V T can be reduced significantly by placing the gain block in front of the input as shown in Figure 13. This effectively reduces the comparator hysteresis by the gain of the gain block. utput ripple can be reduced to just a few millivolts using this technique. Ripple reduction works with stepdown or inverting modes as well. For this technique to be effective, output capacitor C1 must be large, so that each switching cycle increases V T by only a few millivolts. 1000µF is a good starting value. V IN 47k L1 D1 V T V BAT 220mV REF ET A T PRCER V BAT R3 270k A C1 R3 ET = V LB ( 220mV ) 4.33µA V LB = BATTERY TRIP PINT = 33kΩ R3 = 2MΩ TA07 ( )( ) V = T 1 220mV TA08 Figure 12. etting Low Battery Detector Trip Point Table 1. Inductor Manufacturers MANFACTRER PART NMBER Coiltronics International CTX100-4 eries 984.W. 13th Court urface Mount Pompano Beach, FL umida Electric Co. A CD CDR74 CD05 urface Mount Figure 13. utput Ripple Reduction sing Gain Block Table 2. Capacitor Manufacturers MANFACTRER PART NMBER anyo Video Components -CN eries 2001 anyo Avenue an Diego, CA Nichicon America Corporation PL eries 927 East tate Parkway chaumberg, IL prague Electric Company 150D olid Tantalums Lower Main treet 550D Tantalex anford, ME Matsuo 267 eries urface Mount Table 3. Transistor Manufacturers MANFACTRER PART NMBER Zetex ZTX eries Commack, NY FZT eries urface Mount 13

14 TYPICAL All urface Mount Flash Memory V PP Generator L1* 47µH MBR12T3 5V ±10% MMBT k 22µF 1= PRGRAM 0 = HTDWN 1k MMBF170 C8-12 ENE 47µF 20V V PP 12V 120MA *L1= MIDA CD M TA18 1.5V Powered Laser Diode Driver THIBA TLD nF 1N k 6 A 2N Ω 10Ω MJE210 C1 100 µ F -CN 0.22 µ F CERAMIC 2Ω 1.5V 8 5 ET 4 7 1k* 1N5818 L1 2.2 µ H * ADJT FR CHANGE IN LAER TPT PWER TK 262LYF-0076M LAER DIDE CAE CMMN T BATTERY TERMINAL 170mA CRRENT DRAIN FRM 1.5V CELL (50mA DIDE) N VERHT TA13 1.5V Powered Laser Diode Driver 14

15 TYPICAL All urface Mount 3V to 5V tep-p Converter All urface Mount 9V to 5V tep-down Converter L1* 47µH 220 3V 2x AA CELL ENE MBRL120 5V 40mA 10µF 9V -5 ENE L1* 47µH MBRL120 10µF 5V 40mA *L1 = CILCRAFT 1812L-473 TA09 *L1 = CILCRAFT 1812L-473 TA10 All urface Mount 1.5V to 10V, 5V Dual utput tep-p Converter All urface Mount 1.5V to ±5V Dual utput tep-p Converter L1* 82µH 4.7µF 10V 3mA 490k 1.5V AA R AAA CELL 4.7µF 11k L1* 82µH 4.7µF 5V 3mA 4.7µF 1.5V AA R AAA CELL ENE 4.7µF 5V 4mA 5V 4mA 4.7µF = MBRL120 *L1 = CILCRAFT 1812L-823 TA11 = MBRL120 *L1 = CILCRAFT 1812L-823 TA12 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15

16 PACKAGE DECRIPTI Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead Plastic DIP ( ) ( ) ± (3.302 ± 0.127) (10.160) MAX ( ) ( ) (1.651) TYP ± (1.143 ± 0.381) ± (2.540 ± 0.254) (3.175) MIN ± (0.457 ± 0.076) (0.508) MIN ± (6.350 ± 0.254) 8 Package 8-Lead Plastic IC * ( ) ( ) ( ) TYP ( ) ( ) ( ) *THEE DIMENIN D NT INCLDE MLD FLAH R PRTRIN. MLD FLAH R PRTRIN HALL NT EXCEED INCH (0.15mm) (1.270) BC ( ) * ( ) 16 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA (408) FAX: (408) TELEX: LT/GP K REV B PRINTED IN A LINEAR TECHNLGY CRPRATIN 1994