# S-Band Low Noise Amplifier Using the ATF Application Note G004

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1 S-Band Low Noise Amplifier Using the ATF Application Note G004 Introduction This application note documents the results of using the ATF in low noise amplifier applications at S band. The ATF device is capable of a 0.5 db noise figure at 2.3 GHz. The ATF is packaged in a low cost 100 mil micro-x package. PT.A Γ O Zo PT.B Design Procedure To achieve the lowest possible noise figure that the device is capable of producing, the input matching structure must transform the system impedance, usually 50 ohms, to an impedance represented by Γ O. Γ O is the source reflection coefficient that must be presented to the device for it to yield the rated noise figure. This is in contrast to presenting to the device the complex conjugate of S11 which will match the device for maximum gain and minimum input VSWR. Figure 1. Smith Chart Shows Input Match Plotting Γ O versus frequency for the ATF and interpolating the data for a frequency of 2.3 GHz reveals a reflection coefficient of 59 degrees. This is plotted on the Smith Chart shown in Figure 1 as Point B. One matching solution is to use a series quarterwave transmission line to match the real part and a reactive component to match the imaginary part of the reflection coefficient. A quarterwave transmission line of the appropriate characteristic impedance Zo will match two impedances over a narrow bandwidth. The following is the formula for calculating the required characteristic impedance: Zo = Z1 x Z2 A transmission line of characteristic impedance of 44 ohms transforms the 50 ohm source impedance to 39 ohms. This is plotted as Point A on Figure 1. At this point, it is still the resistive axis. A series inductor with an inductive reactance of 74 ohms then transforms the impedance at Point A to Γ O at Point B. 74 ohms of inductive reactance is equivalent to a series inductance of 5 nh at 2.3 GHz (Z = jwl). An inductor of this size is often difficult to build; and more difficult to tune during the initial breadboard stage of design. Quite conveniently,

2 2 a series inductive reactance can be simulated by an equivalent shunt capacitive reactance when it is physically spaced a quarter wavelength away on the transmission line. Using the Smith Chart, it can be shown that the equivalent shunt component can be calculated from the following formula: Xc = -Zo 2 XI For a 5 nh inductor, Xc calculates to be 26.2 ohms or 2.6 pf at 2.3 GHz. λ/4 Zo = 44 Ω 5 nh Shown in Figure 2 are the resultant input matching networks. The LNA design will be based on the network with the shunt capacitive element. Since the input is terminated in Γ O and not 50 ohms, the output of the FET must be matched to Γ L = S22'. This accounts for the imperfect isolation through the FET. Γ L is given by: Γ L = S 22 + S 12 x S 21 x Γ O 1 - S 11 x Γ O Substituting in the parameters for the ATF at 2.3 GHz: SERIES INDUCTOR APPROACH 2.6 pf λ/4 Zo = 44 Ω SHUNT CAPACITOR APPROACH Figure 2. Input Matching Networks S11 = -74 degrees S22 = -34 degrees S12 = 55 degrees S21 = 105 degrees Γ O = 59 degrees 1.7 nh λ/4 Zo = 37 Ω Yields Γ L = 115 degrees Note: The output reflection coefficient has been significantly transformed due to the change in generator impedance presented to the device. A Smith Chart exercise similar to that used to generate the input network yields the circuits shown in Figure 3 for the output network. Computer Optimization The initial design can be performed quite easily with a basic understanding of the Smith Chart. It provides a basis with which to begin optimization. With the help of a computer, modeling and optimizing the basic matching networks for a desired performance is attainable. One of the main advantages of a computer optimization program is the ability to model the electrical effects associated with the actual circuit realization of the LNA. Examples include the discontinuity associated with soldering the inch wide lead down to a piece of etch that is significantly wider; the effect of several lines intersecting at one point; the effect of adding plated through-holes to ground the source leads. These small yet often significant discontinuities are often tedious to model on the Smith Chart, but are a trivial matter for the computer. The simulation software is especially useful when modeling source inductance on the GaAs FET and making SERIES INDUCTOR APPROACH λ/4 Zo = 37 Ω 1.2 pf SHUNT CAPACITOR APPROACH Figure 3. Output Matching Networks

3 3 tradeoffs between noise figure, gain, input VSWR and stability. Although an LNA design may be unconditionally stable at the operating frequency, there will be no guarantee that the LNA will not oscillate at some other frequency unless a stability analysis is performed. The analysis should be performed at all frequencies over which the device has gain. An out-of-band oscillation could possibly be at a large enough amplitude to cause a reduction in in-band gain or an increase in noise figure. Although it is not necessary that an LNA be unconditionally stable at all frequencies, stability should be analyzed so that the areas of potential instability can be evaluated and problem source and load impedances can be avoided. The revised circuit after optimization is shown in Figure 4. The input matching network consists of microstriplines Z1 and Z2. It was later found empirically and confirmed on the computer that adding an additional stub Z3 reduces the noise figure slightly. The blocking capacitors were chosen to be 10 pf to offer some low frequency rejection. Quarterwave lines are utilized for bias decoupling. 50 ohm chip resistors are used for low frequency resistive loading and ensure stability at low frequencies. Both 1000 pf and 0.1 µf capacitors provide a low impedance path to ground for the 50 ohm chip resistors. It was also determined with the help of the computer that using a quarterwave open circuited low impedance line in the gate bias decoupling circuit as opposed to using a bypass capacitor also improved stability near the band of operation. Z3 Z2 Z5 INPUT 50 Ω 10 Z1 Z Ω OUTPUT 1000 LL K 10 K 10 µf 2N2907 R1 R2 REGULATOR 5 V V CC 10 µf + DC - DC INVERTER R3 Figure GHz LNA Schematic

5 5 The negative voltage can be supplied from a DC-DC converter driven by the same voltage regulator that feeds the active bias network. A DC- DC converter circuit can be built around a standard NE555 timer device. A suitable circuit is described in detail in Hewlett-Packard Application Note AN-A002. A simpler approach is to use one of the newer integrated DC-DC converters. The Teledyne Semiconductor TSC7662A used in the active bias circuit shown in Figure 4 has been used successfully in many circuits. The drain voltage is determined by resistors R2 and R3 shown in Figure 4. Resistor R1 sets the drain current. The typical bias point for low noise operation for the ATF is shown in Table 1. Suggested resistor values for the active bias networks are also shown. Table 1. Device V ds I d R 1 R 2 R 3 ATF V 25 ma Performance Typical LNA performance for the ATF LNA is shown in Table 2. Measured performance of several LNAs as compared to the computer simulation is within a tenth of a db on noise figure and within a db on gain. Table 2. ATF Noise Figure Gain Actual 0.60 db 13.5 db Simulation 0.54 db 14.4 db

6 6 Figure 5. 1X Artwork The artwork shown in Figure 5 is designed for material with a dielectric constant of 2.2 and a thickness of inch. Suggested material includes Duroid 5880 or Taconics TLY-5. The Touchstone computer simulation for the ATF LNA is contained in Tables 3 and 4. Reference: 1. A. Ward, Low-Noise VHF and L-Band GaAsFET Amplifiers, RF Design, Feb All registered trademarks and trademarks remain the property of their respective companies.

7 Figure 6. TouchstoneTM Circuit Configuration 7

8 Table 3. Touchstone Simulation for ATF LNA 8

9 For technical assistance or the location of your nearest Hewlett-Packard sales office, distributor or representative call: Americas/Canada: or (408) Far East/Australasia: Call your local HP sales office. Japan: (81 3) Europe: Call your local HP sales office. Data Subject to Change Copyright 1993 Hewlett-Packard Co. Obsoletes E (10/93) Printed in U.S.A E (8/98)

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